FIELD
The present disclosure relates to the field of radar sensors, in particular to a novel concept of phase calibration with crosstalk cancellation.
BACKGROUND
Radar sensors can be found in numerous sensing applications in which distances and velocities of objects are to be measured. In the automotive sector, there is an increasing demand for radar sensors that may be used in so-called advanced driver-assistance systems (ADAS). Examples of an advanced driver assistive system are “adaptive cruise control” (ACC) and “radar cruise control” systems. Such systems may be used to automatically adjust the speed of an automobile so as to maintain a safe distance from other automobiles driving ahead. Other examples of advanced driver assistive system are blind-spot monitors, which may employ radar sensors to detect other vehicles in the blind spot of a vehicle. Particularly autonomous cars may use numerous sensors, such as radar sensors, to detect and locate various objects in their surroundings. Information about the position and velocity of objects in the area of an autonomous car is used to help navigate safely.
Modern radar systems make use of highly integrated RF circuits which may incorporate all core functions of an RF font-end of a radar transceiver in one single package (single chip transceiver). Such RF front-ends usually include, inter alia, a local RF oscillator (LO), power amplifiers (PA), low-noise amplifiers (LNA), and mixers. Frequency-modulated continuous-wave (FMCW) radar systems use radar signals whose frequency is modulated by ramping the signal frequency up and down. Such radar signals are often referred to as “chirp signals” or simply as “chirps”. A radar sensor usually radiates sequences of chirps using one or more antennas, and the radiated signal is backscattered by one or more objects (referred to as radar targets) located in the “field of view” of a radar sensor. The backscattered signals (radar echoes) are received and processed by the radar sensor. The detection of the radar targets is usually accomplished using digital signal processing.
Modern FMCW radar systems may include multiple input and multiple output channels and are thus referred to as multiple input/multiple output (MIMO) systems. However, in simple systems one input channel and one output channel may be sufficient. The RF front-ends of the radar systems may be distributed across a plurality of semiconductor chips, which are referred to as monolithic microwave integrated circuits (MMICs). Such radar systems are not only capable of measuring distances but also the respective velocities and azimuth angles (also referred to as Direction of Arrival, DoA, of the radar echoes). However, the RF frontend of a radar system may also be integrated in a single MMIC (single chip radar).
Particularly the angle measurement needs a calibration of the phases of the transmitted radar signal in order to obtain the desired accuracy. In known systems crosstalk may negatively affect the quality of the calibration.
SUMMARY
A method for the use in a radar system is described herein. In accordance with one embodiment, the method includes providing a local oscillator signal to an RF output channel of a radar system. The RF output channel is configured to generate, in an enabled state, an RF output signal based on the local oscillator signal. The method further includes determining a first measurement signal based on the local oscillator signal and a representation of the RF output signal, while the RF output channel is disabled, and thus the first measurement signal represents crosstalk. Further, the method includes determining a second measurement signal based on the local oscillator signal and the representation of the RF output signal while the RF output channel is enabled. A phase value associated with the RF output channel is determined based on the first measurement signal and the second measurement signal.
Furthermore, a radar system is described herein. In accordance with one embodiment the radar system includes a local oscillator providing a local oscillator signal to an RF output channel of the radar system. The RF output channel is configured to generate, when in an enabled state, an RF output signal based on the local oscillator signal. Further, the system includes a circuit configured to determine a first measurement signal based on the local oscillator signal and a representation of the RF output signal, while the RF output channel is in a disabled state, whereby the first measurement signal represents crosstalk; the circuit is further configured to determine a second measurement signal based on the local oscillator signal and the a representation of the RF output signal, while the RF output channel is in the enabled state. Further, the system includes a processing unit configured to determining a phase value based on the first measurement signal and the second measurement signal.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention can be better understood with reference to the following drawings and descriptions. The components in the figures are not necessarily to scale; instead emphasis is placed upon illustrating the principles of the invention. In the figures, like reference numerals designate corresponding parts. In the drawings:
FIG. 1 is a sketch illustrating the operating principle of an FMCW radar system for distance and/or velocity measurement.
FIG. 2 includes two timing diagrams illustrating the frequency modulation of the RF signal used in FMCW radar systems.
FIG. 3 is a block diagram illustrating the basic structure of an FMCW radar device.
FIG. 4 is a circuit diagram illustrating one example of an analog RF frontend, and analog base-band signal processing.
FIG. 5 is a circuit diagram illustrating a radar system with a plurality of RF output channels and a monitoring circuit for measuring the phases of the RF output signals of the RF output channels.
FIG. 6 illustrates the superposition of crosstalk and a representation of the RF output signal of an RF output channel as an addition of complex phasors.
FIG. 7 illustrates one example of a measurement signal.
FIG. 8 illustrates the discrete spectrum of the measurement signal.
FIG. 9 is a flow chart illustrating one exemplary embodiment of the phase measurement described herein.
DETAILED DESCRIPTION
FIG. 1 illustrates a conventional FMCW radar sensor 1. In the present example, separate transmission (TX) and reception (RX) antennas 5 and 6, respectively, are used (bistatic or pseudo-monostatic radar configuration). However, it is noted that a single antenna can be used, so that the reception antenna and the transmission antenna will be physically the same (monostatic radar configuration). The transmission antenna 5 (quasi-) continuously radiates an RF signal sRF(t), which is frequency-modulated, for example, by a saw-tooth-shaped signal. When the radiated signal sRF(t) is back-scattered at an object T, which is located in the field of view of the radar system, the back-scattered RF signal yRF(t) is received by the reception antenna 6. The object T is usually referred to as “radar target”. In a more general example, several targets may be in the field of view of a radar sensor. Further, an antenna array may be used instead of a single RX antenna. Similarly, an antenna array may be used instead of a single TX antenna. Using multiple RX and TX antennas in a multi-channel radar system allows for the measurement of the angle of incidence of a radar echo (azimuth angle), usually referred to as direction of arrival (DoA). Measurement of the direction of arrival is important for many applications, and thus most radar sensors will make use of antenna arrays. To keep the drawings simple, only one TX antenna and one RX antenna are shown in FIGS. 1 and 3. It is understood that the concepts described with reference to these figures are readily applicable to radar sensors with multiple input and output channels and respective antenna arrays.
FIG. 2 illustrates the mentioned conventional frequency-modulation of the signal sRF(t). As shown in the top diagram of FIG. 2, the signal sRF(t) is composed of a sequence of “chirps”, i.e. sinusoidal waveforms with increasing (up-chirp) or decreasing (down-chirp) frequency. In the present example, the instantaneous frequency fLO(t) of a chirp increases linearly from a start frequency fSTART to a stop frequency fSTOP within a defined time span TRAMP (see bottom diagram of FIG. 2). Such a chirp is also referred to as linear frequency ramp. A frequency-modulated signal with a sequence of three identical linear frequency ramps is illustrated in FIG. 2. It is noted, however, that the parameters fSTART, fSTOP, TCHIRP as well as the pause between the individual frequency ramps may vary dependent on the actual implementation of the radar device 1 and may also vary during operation of the radar device. For one radar measurement a ramp sequence of, e.g., 256 or 512 frequency ramps may be used.
FIG. 3 is a block diagram that illustrates an exemplary structure of radar sensor 1. Accordingly, at least one transmission antenna 5 (TX antenna(s)) and at least one reception antenna 6 (RX antenna(s)) are connected to an RF frontend 10, which may be integrated in a single semiconductor chip, usually referred to as monolithic microwave integrated circuit (MMIC). As mentioned, the RF circuitry may also be distributed across more than one chip. The RF frontend 10 may include all the circuit components needed for RF signal processing. Such circuit components may include, for example, a local oscillator (LO), RF power amplifiers, low noise amplifiers (LNAs), directional couplers such as rat-race-couplers and circulators, and mixers for the down-conversion of RF signals (e.g. the received signal yRF(t), see FIG. 1) into the base-band or IF-band. As mentioned, antenna-arrays may be used instead of single antennas. The depicted example shows a bistatic (or pseudo-monostatic) radar system, which has separate RX and TX antennas. In case of a monostatic radar system, a single antenna or a single antenna array may be used for both, receiving and transmitting electromagnetic (radar) signals. In this case a directional coupler (e.g. a circulator) may be used to separate RF signals to be transmitted to the radar channel from RF signals received from the radar channel.
In the case of an FMCW radar sensor, the RF signals radiated by the TX antenna 5 may be in the SHF (Super High Frequency) or the EHF (Extremely High Frequency) band, e.g. in the 24 GHz ISM band or in the range of e.g. about 76-81 GHz in automotive applications. As mentioned, the RF signal received by the RX antenna 6 includes the radar echoes, i.e. the signals that have been back-scattered at the radar target(s). The received RF signal yRF(t) is down-converted into the base-band and further processed in the base-band using analog signal processing (see FIG. 3, base-band signal processing chain 20), which basically includes filtering and amplification of the base-band signal and thus determines the bandwidth of the received signal. The base-band signal is finally digitized using one or more analog-to-digital converters 30 and further processed in the digital domain (see FIG. 3, digital signal processing chain implemented, e.g., in digital signal processor 40). The overall system is controlled by a system controller 50, which may be at least partly implemented using a processor executing appropriate software/firmware. The processor may be included, e.g. in a microcontroller, a digital signal processor, or the like. The digital signal processor 40 (DSP) may be part of the system controller 50 or separate therefrom. The RF frontend 10 and the analog base-band signal processing chain 20 and optionally the also the ADC 30 as well as part of the digital signal processing may be integrated in a single MMIC. However, the components may be distributed among two or more integrated circuits.
FIG. 4 illustrates one exemplary implementation of the RF frontend 10, which may be included in the radar system shown in FIG. 3. It is noted that FIG. 4 is a simplified circuit diagram illustrating the basic structure of an RF frontend. Actual implementations, which may heavily depend on the application, are of course more complex and may include several RX channels (input channels) and/or TX channels (output channels) in a single MMIC. The RF frontend 10 includes a local oscillator 101 (LO) that generates a RF signal sLO(t), which may be frequency-modulated as explained above with reference to FIG. 2. The signal sLO(t) is also referred to as LO signal. Usually, the local oscillator 101 includes a phase-locked loop which is clocked by a clock signal sCLK(t).
The LO signal sLO(t) is processed in the transmission signal path as well as in the reception signal path. The transmission signal sRF(t) (outgoing radar signal), which is radiated by the TX antenna 5, is generated by amplifying the LO signal sLO(t), e.g., using an RF power amplifier 102. The output of the amplifier 102 is coupled to the TX antenna 5. The RF signal processing chain coupled between the local oscillator 101 and a particular TX antenna is referred to as TX channel or RF output channel, which is labelled TX1 in the example of FIG. 4. In order to adjust the phase of the outgoing radar signal sRF(t) the respective output channel TX1 includes a phase shifter 103, which may be coupled between an input circuit node of the output channel TX1 (at which the LO signal sLO(t) is received) and the RF amplifier 102. The phase shifter 103 may also be placed after RF amplifier 102 or may be a part of RF amplifier 102. An RF output channel may be enabled or disabled, e.g. by enabling or disabling the RF amplifier 102.
The received signal yRF(t) (incoming radar signal), which is provided by the RX antenna 6, is directed to a mixer 104. In the present example, the received signal yRF(t) (i.e. the antenna signal) is pre-amplified by RF amplifier 105 (gain g), so that the mixer receives the amplified signal g·yRF(t) at its RF input port. The mixer 104 further receives the LO signal sLO(t) at its reference input port and is configured to down-convert the amplified signal g·yRF(t) into the base band. The resulting base-band signal at the mixer output is denoted as yBB(t). The base-band signal yBB(t) is further processed by the analog base band signal processing chain 20 (see also FIG. 3), which basically includes one or more filters (e.g. a band-pass 21 or a combination of high pass and low pass filters) to remove undesired side bands and image frequencies as well as one or more amplifiers such as amplifier 22. The analog output signal, which may be supplied to an analog-to-digital converter (cf. FIG. 3), is denoted as y(t). Various techniques for the digital post-processing of the digitized output signals (digital radar signal) are as such known (e.g. Range Doppler Analysis) and thus not further explained herein. The RF signal processing chain coupled between a particular RX antenna and the ADC that provides the respective digital base band signal is referred to as RX channel or RF input channel, which is labelled RX1 in the example of FIG. 4.
In the present example, the mixer 104 down-converts the RF signal g·yRF(t) (amplified antenna signal) into the base band. The respective base band signal (mixer output signal) is denoted by yBB(t). The down-conversion may be accomplished in a single stage (i.e. from the RF band into the base band) or via one or more intermediate stages (from the RF band into an intermediate frequency band and subsequently into the base band).
FIG. 5 illustrates one example of a radar system with more than one RF output channel, wherein the RF input channels, which are not necessary for the further discussion have been omitted to simplify the illustrations. It is understood, that the RX channels of the radar systems described below may be implemented as shown, for example, in FIG. 4. According to FIG. 5, the radar system includes a local oscillator 101 and four RF output channels TX1, TX2, TX3, and TX4. Each one of the RF output channels TX1, TX2, TX3, and TX4 receives the LO signal sLO(t) and is configured to generate—in an enabled state—an RF output signal sRF,1(t), sRF2(t), sRF,3(t), and, respectively, sRF,4(t) based on the local oscillator signal sLO(t)). The RF output channels TX1, TX2, TX3, and TX4 may be implemented similar to the output channel TX1 in the example of FIG. 4.
The RF output channels TX1, TX2, TX3, and TX4 are configured to be enabled or disabled, which may be done by enabling and disabling the power amplifier 102 and/or the phase shifter 103 included in each RF output channel. The LO signal sLO(t) is distributed to the RF output channels TX1, TX2, TX3, and TX4 by the signal distribution circuit 100, which may be implemented, for example, using one or more RF power divider circuits, strip lines and other RF circuit components. Although not shown in the example of FIG. 6, the signal distribution circuit 100 may also be configured to provide the LO signal sLO(t) to the RF input channels (RX channels, not shown in FIG. 5). Furthermore, each RF output channels TX1, TX2, TX3, and TX4 is configured to provide a representation of the respective RF output signals sRF,1(t), sRF,2(t), sRF,3(t), sRF,4(t). For this purpose, each RF output channel TX1, TX2, TX3, and TX4 includes a coupler 106 (e.g. a directional coupler such as a rat race coupler, a coupled-line directional coupler, or the like) which is arranged between the amplifier output 102 and the RF output port, which forms basically an interface to the antenna. The coupler 106 directs the RF output signal sRF,1(t), sRF,2(t), sRF,3(t), sRF,4(t) to the respective output port and also provides a representation of the respective RF output signal, which is herein also referred to as feedback signal sFB,1(t), sFB,2(t), sFB,3(t), and sFB,4(t). The RF output signals sRF,k(t) and the respective feedback signals sFB,k(t) (with k=1, . . . , 4) have a fixed phase relation and differ in amplitude. That is, the feedback signals are scaled and phase shifted (with a fixed offset) versions of the respective RF output signals, and, when the phase of the RF output signals sRF,k(t) change, then the phases of the respective feedback signals will change accordingly. In other words, the phase of a feedback signal is indicative of the phase of the respective RF output signal. The antennas connect to the respective RF output ports at chip contacts which may be solder balls or the like dependent on the chip package used.
Particularly in radar systems with more RF output channels the phases of the RF output signals al sRF,1(t), sRF,2(t), sRF,3(t), sRF,4(t ) need to be calibrated in order to improve the quality and accuracy of measurements (particularly DoA measurements). In this regard it should be noted that the phase may vary, for example, due to temperature variations, aging effects, etc. Therefore, a monitoring of the phases and a regular calibration may be desirable.
To determine a phase a reference phase is usually needed, and, in the present example, the reference phase is the phase ϕLO of the LO signal sLO(t). In the example of FIG. 5, to obtain a measurement signal which is indicative of the phase ϕTXk of the RF output signal sRF,k(t) of channel TXk (k=1, . . . , 4), the representation of the RF output signal sRF,k(t) (i.e. the feedback signal sFB,k(t)) is mixed with the LO signal sLO(t) using a mixer 302, wherein an additional phase shift is imposed on the LO signal sLO(t). This additional phase shift is referred to as test phase ϕTSG, which can be set by phase shifter 301, which receives a control input indicative of the desired test phase ϕTSG. It is noted that, when the feedback signal sFB,k(t)) is a sine signal with a specific frequency fLO and phase ϕTXk, and the LO signal sLO(t) is a sine signal with the same frequency fLO and phase ϕLO, then the mixer output signal will be a DC signal with a signal level indicative of the phase ϕTXk (relative to the phase ϕLO). In essence, the mixer output signal will be indicative of the phase difference ϕTXk−ϕLO, wherein ϕLO may be defined as zero for the current discussion without loss of generality. The mixer output signal is referred to as measurement signal mk(ϕTSG); it depends on the setting of the test phase ϕTSG and the index k corresponds to the output channel TXk which is enabled during the measurement. k=0 means that none of the RF output channels is enabled during a phase measurement.
Only one selected RF output channel, e.g. TX1, is enabled for a measurement of the respective phase, e.g. ϕTX1, wherein the other RF output channels (TX2-TX4 in the present example) are disabled. In the following the phase measurement is explained in more detail for RF output channel TX1. It is understood that the same procedure can be done for the other RF output channels TX2-TX4. In practice, the feedback signal sFB,1(t)) will be distorted by crosstalk when arriving at the RF port of the mixer 302. The distorted feedback signal is denoted as h1(t) and generally as hk(t) with k=1, . . . , 4 for the channels TX1-TX4. Accordingly, the feedback signal including crosstalk can be written as:
h
1(t)=sRF,1(t)+c(t), (1)
wherein c(t) denotes the crosstalk which is also a sine signal with the same frequency as the LO signal sLO(t) and therefore may disturb the measurement of phase ϕTX1. When the second channel TX2 is enabled (and all other channels disabled), the feedback signal including crosstalk can be written as:
h
2(t)=sRF,2(t)+c(t), (2)
and, similarly, for the third and the fourth channel TX3 and TX4, respectively.
h
3(t)=sRF,3(t)+c(t), (3)
h
4(t)=sRF,4(t)+c(t), (4)
It is clear from equation (1) that the crosstalk c(t) contributes to the measured phase ϕTX1 and thus causes a measurement error. The concept developed herein provides a crosstalk cancellation before the actual calculation of the sought phase value ϕTX1. The concept is illustrated by the phasor diagrams of FIG. 6, which illustrate the signals sRF,1(t) and c(t) and h1(t) as complex-valued phasors sRF,1*(t) and, respectively c*(t). The angle between the real axis and the phasor represents the phase of the respective signal. It is noted that the representation of sinusoidal signals by phasors is well-known and thus not further explained here. When disabling all RF output signals TX1-TX4 the mixer 302 only receives the crosstalk which is also denoted as
h
0(t)=c(t). (5)
Generally, the feedback signal arriving at the mixer 302 is denoted as hk(t). As mentioned, an index k=0 refers to a situation in which all RF output channels are disabled, whereas k=1, 2, 3, 4 refers to the channels TX1, TX2, TX3 or TX4 enabled, respectively. The corresponding measurement signal is denoted as mk(ϕTSG). As mentioned, for a single specific test phase value ϕTSG, mk(ϕTSG) represents a DC signal. However, when acquiring several samples of the measurement signal mk(ϕTSG) for different test phase values ϕTSG, the resulting discrete signal (sequence of samples) represents amplitude and phase of the corresponding feedback signal hk(t).
The crosstalk signal h0(t)=c(t) is illustrated in the left diagram of FIG. 6. The corresponding measurement signal m0(ϕTSG) provided by the mixer 302 is indicative of the crosstalk (amplitude and phase). In a second measurement, e.g., channel TX1 is enabled and the mixer 302 receives the superposition h1(t)=c(t)+sFB,1(t) of crosstalk c(t) and feedback signal sFB,1(t) this superposition is illustrated in the right diagram of FIG. 6. The respective measurement signal m1(ϕTSG) provided by the mixer 302 is indicative of the superposition ‘feedback signal plus crosstalk’ (i.e amplitude and phase of the distorted feedback signal h1(t)). The same can be done for the remaining RF output channels and the respective feedback signals.
In the right diagram of FIG. 6 one can see that a (theoretically) correct crosstalk cancellation can be achieved by subtracting the crosstalk phasor c*(t) from the phasor representing the distorted feedback signal h1*(t). At this point it should be noted that it is not enough to simply subtract the phase of the crosstalk (see FIG. 6, left diagram, phasor c*) from the phase of the distorted feedback signal (see FIG. 6, left diagram, phasor h1*). To obtain a correct cancellation, the complex phasors must be considered.
Referring again to FIG. 5 it is noted that the subtraction of the crosstalk can be done in the base band. As mentioned, the measurement signals/mixer output signals mk(ϕTSG) are DC signals which can easily be sampled, digitized and stored for different values of ϕTSG. Accordingly, the measurement signal m0(ϕTSG) that represents the crosstalk (and is obtained when all output channels are disabled) can be stored for a sequence of different test phases ϕTSG and used later in the calculation of the sought phase ϕTX1.
The process of the acquisition of a measurement signal mk(ϕTSG) is further explained below with reference to FIG. 7. In the present example the test phase ϕTSG is varied in steps of 45° from 0° to 315° which corresponds to a full rotation of the phase; a further step of 45° would result in a phase of 360° which would be the start of the next rotation. That is, the test phase ϕTSG can be expressed as
ϕTSG[n]=n·360/N, for n=0, . . . , N−1, (6)
wherein N describes the length of the sequence and thus the number of samples obtained for the measurement signal mk(ϕTSG), which can also be written as discrete signal mk[n·360/N]. In the example of FIG. 7, the number of samples is N=8. The phase of the discrete signals mk[n·360/N] represents the phase of the signal hk(t).
In the example of FIG. 7, the phase ϕTSG is varied in equidistant phase steps Δϕ=360/N until a full phase rotation has been reached. In other words, the N samples of mk[n·360/N] may be regarded as exact one period of a periodic sinusoidal discrete signal and, therefore, the Discrete Fourier Transform of mk[n·360/N] yields a sequence Mk[m] (for m=0, . . . N−1), in which all samples are zero except the spectral value Mk[1]. One example of such a sequence of spectral values (also referred to as spectral lines) is shown in FIG. 8.
It is understood that the spectral value Mk[1] of the sequence
M
k[m]=DFT{mk[n·360/N]} (7)
is a complex value having a magnitude |Mk[1]| and an argument arg{Mk[1]}. If the phase of the ϕTX1 of the RF output signal sRF,1(t) of the RF output channel TX1 is calculated based on the difference m1[n·360/N]−m0[n·360/N], then the effect of the crosstalk is cancelled. As the Discrete Fourier Transform is a linear operation, this difference may be calculated before or after the Fourier transform. That is,
ϕTX1=arg{DFT{m1[n·360/N]−m0[n·360/N]}|m=1}, or (8)
ϕTX1=arg{M1[1]−M0[1]}, (9)
wherein |m=1 means that only the sample with index m=1 is considered when calculating the argument. The same can be done for the other RF output channels TX2, TX3, TX4.
The Discrete Fourier Transform may be implemented using the well-known Fast Fourier Transform (FFT) algorithm. However, other algorithms are known to calculate the spectral values Mk[1]. It is noted that, according to the examples described herein, the crosstalk can be represented by a single complex number, namely M0[1], which can be stored in a memory and later used to cancel the crosstalk component in the calculations of the phases ϕTX1, e.g. according to equation 9, which may be generalized to
ϕTXk=arg{Mk[1]−M0[1]}, for k=1, 2, etc. (10)
The complex-valued difference Mk[1]−M0[1] corresponds to the subtraction of the phasors hk*(t)−c*(t) illustrated in the right diagram of FIG. 6. However, the difference Mk[1]−M0[1] is determined using base-band signals instead of RF signals. If the crosstalk cancellation is done before the Discrete Fourier Transform in accordance with equation 8, then the crosstalk is, alternatively, represented by N real values, namely m0[n·360/N] for n=0, . . . N−1, (instead of by a single complex number) which can also be stored in a memory and later used for crosstalk cancellation.
The concept for phase measurement of the phases ϕTX1, ϕTX2, ϕTX3, ϕTX4 of the RF output signals sRF,1(t), sRF,2(t), SRF,3(t), sRF,4(t) of the RF output channels TX1, TX2, TX3, and TX4, respectively, are further described below with reference to the flow chart of FIG. 9. According to the embodiment illustrated by FIG. 9, the method includes providing a local oscillator signal sLO(t) to an RF output channel TX1 of a radar system (see FIG. 9, step S1). The RF output channel is configured to generate, in an enabled state, an RF output signal sRF,1(t) based on the local oscillator signal sLO(t).
The method illustrated by FIG. 9 further includes determining a first measurement signal m0(ϕTSG) (see also FIG. 7) based on the local oscillator signal sLO(t) and a representation (denoted SFB,1(t)) of the RF output signal sRF,1(t), while the RF output channel TX1 is disabled (see FIG. 9, step S2). In this situation, as the RF output channel TX1 is disabled (as well as all other RF output channels if present), the first measurement signal m0(ϕTSG) represents merely crosstalk c(t). The method further includes determining a second measurement signal m1(ϕTSG) based on the local oscillator signal sLO(t) and the representation (sFB,1(t)) of the RF output signal sRF,1(t), while the RF output channel TX1 is enabled (see FIG. 9, step S2). In this situation, the second measurement signal m1(ϕTSG) represents the RF output signal sRF,1(t) and crosstalk c(t).
The method further includes determining a phase value ϕTX1 associated with the RF output channel TX1 based on the first measurement signal m0(ϕTSG) and the second measurement signal m1(ϕTSG) (see FIG. 9, step S4). Determining the phase value ϕTX1 may include a crosstalk cancellation as explained in detail above (see, for example, equations 7 and 10). Accordingly, a test signal sTSG(t) is generated based on the local oscillator signal sLO(t), wherein either the test signal sTSG(t) or the representation sFB,1(t) of the RF output signal sRF,1(t) is phase shifted by a by a test phase ϕTSG (see FIG. 5, phase shifter 301). In other words, an additional phase shift equal to the test phase ϕTSG is generated between the test signal sTSG(t) and the representation sFB,1(t) of the RF output signal sRF,1(t) and, subsequently, the test signal sTSG(t) and the representation sFB,1(t) of the RF output signal (sRF,1(t)) are mixed (see FIG. 5, mixer 302), while the RF output channel TX1 is disabled (as well as any other RF output channels if present). Accordingly, the mixer 302 receives only the test signal sTSG(t) and crosstalk signal h0(t)=c(t), wherein the test phase ϕTSG is imposed either on the test signal sTSG(t) or the crosstalk signal h0(t). The first measurement signal m0(ϕTSG) is determined by sampling the mixer output signal for various test phases ϕTSG, while the RF output channel TX1 (and all other RF output channels if present) are disabled. The second measurement signal m1(ϕTSG) is determined in the same way wherein, however, the RF output channel TX1 is enabled (while all other RF output channels, if present, remain disabled). Similarly, a third measurement signal m2(ϕTSG) may be determined, if (at least) a second RF output channel TX2 is present. In this case the RF output channel TX2 will be enabled while all other RF output channels are disabled. This procedure can be repeated for an arbitrary number of RF output channels.
As explained in detail above, the crosstalk compensation may be achieved by subtracting the measurement signal m0(ϕTSG), which represents the crosstalk, from the other measurement signals mk(ϕTSG) (k=1, 2, . . . , for channels TXk). As discussed above, this difference may also be calculated in the frequency domain. The sought phase ϕTXk associated with the RF output channel TXk is then calculated based on a spectral line associated with the difference signals mk(ϕTSG)−m0(ϕTSG).
In the examples described above, the measurement signals m0(ϕTSG), m1(ϕTSG), m2(ϕTSG), etc. have been recorded by sampling the mixer output signal for different test phases ϕTSG, wherein the test phase is increased, starting at 0° by a constant increment of, e.g. 360°/N, wherein N is the number of samples recorded for each measurement signal m0(ϕTSG), m1(ϕTSG), etc. Accordingly, the test phases, at which the mixer output signal is samples, form a sequence that can be written as
ϕTSG=n·360°/N, for n=0, . . . , N−1,
and that covers exactly one full rotation of the phase ϕTSG, i.e. the interval [0°, 360°). In this case, the second bin with index m=1 of the resulting discrete spectrum Mk[m] is used for calculating the phase ϕTXk (see equation 10). Alternatively, the phase ϕTSG may be varied in constant steps over two or more full rotations; for two full rotations the sequence can be written as
ϕTSG=n·720°/N, for n=0, . . . , N−1,
wherein, in this case, the third bin with index m=2 of the resulting discrete spectrum Mk[m] is used for calculating the phase ϕTXk (analogous to equation 10 but for m=2). Generally, the phase ϕTSG may be varied in constant steps R full rotations resulting in a sequence that can be written as
ϕTSG=n·R·360°/N, for n=0, . . . , N−1,
wherein R is an integer number equal to or greater than 1. In this case, the (R+1)th bin with index m=R of the resulting discrete spectrum Mk[m] is used for calculating the phase ϕTXk (analogous to equation 10 but for m=R).
It is understood that the calculations needed for crosstalk cancellation and all related calculations (such as, e.g. for calculating the Fourier Transform), may be performed by a processing unit, which may be included in, for example, the signal processor 40 or the system controller 50 (see FIG. 3). However, this is not necessarily the case. Generally, the term processing unit may be any entity that is capable of performing the herein-described functions and calculations. Accordingly, the processing unit may include a processor that is capable of executing software/firmware instructions that cause to processor to perform the calculations described herein for crosstalk cancellation, phase measurement and related functions. Additionally, or alternatively, the processing unit may include hard-wired circuitry which is configured to perform calculations without the need for software/firmware instructions. For example, the radar system may include hard-wired circuits that are capable of performing a Fast Fourier Transform.
Once the phase values ϕTX1, ϕTX2, ϕTX3, and ϕTX4 have been determined using the crosstalk compensation technique as explained in detail above, the phase shifts provided by the respective RF output channels TX1, TX2, TX3, and TX4 may be adjusted dependent on the determined phase values ϕTX1, ϕTX2, ϕTX3, and ϕTX4. The phase shifts provided by the respective RF output channels TX1, TX2, TX3, and TX4 can be adjusted by adjusting the phase shift ϕ1, ϕ2, ϕ3, and ϕ4 effected by the phase shifters 103 included in the respective RF output channels (see FIG. 5, phase shift ϕ1 effected by phase shifter 103 of channel TX1). It is understood that the phase shifters 103 as well as the phase shifter 301 may be controlled by the system controller 50 (see FIG. 3). Dependent on the application, one or more of the phase shifts ϕ1, ϕ2, ϕ3, and ϕ4 effected by the phase shifters 103 (of channels TX1-TX4) may be adjusted in way that the differences ϕTX2−ϕTX1, ϕTX3−ϕTX2, and ϕTX4−ϕTX3 match predetermined desired phase differences. It is understood that the concepts described herein can be directly generalized to more than four output channels.
Although the concept described herein has been illustrated and explained with respect to one or more implementations, alterations and/or modifications may be made to the illustrated examples without departing from the spirit and scope of the appended claims. In particular regard to the various functions performed by the above described components or structures (units, assemblies, devices, circuits, systems, etc.), the terms (including a reference to a “means”) used to describe such components are intended to correspond—unless otherwise indicated—to any component or structure, which performs the specified function of the described component (e.g., that is functionally equivalent), even though not structurally equivalent to the disclosed structure, which performs the function in the herein illustrated exemplary implementations of the invention.
Patent Application
20210025973
