Patent Application
20060252396
The present invention relates generally to RF integrated circuits, and particularly to methods and devices for generating multiple-phase signals using polyphase architecture.
Polyphase signals, i.e., groups of signals having multiple phases at a common frequency, are used for a variety of RF (Radio Frequency) and communications applications.
Polyphase signals are often used as local-oscillator (LO) signals for frequency up-conversion and down-conversion. For example, Sheng et al., in a paper entitled “A Wide-Bandwidth Si/SiGe HBT Direct Conversion Sub-Harmonic Mixer/Downconverter,” IEEE Journal of Solid-State Circuits (35:9), September 2000, pages 1329-1337, which is incorporated herein by reference, describe a wideband sub-harmonic mixer/direct-conversion down-converter implemented in a Si/SiGe HBT technology. The circuit utilizes a combination of phase shifters operating at 45° and 90°.
The use of RC (resistive-capacitive) networks for generation of polyphase LO signals are described by Galal et al., in a paper entitled “RC Sequence Asymmetric Polyphase Networks for RF Integrated Transceivers,” IEEE Transactions on Circuits and Systems-II: Analog and Digital Signal Processing (47:1), January 2000, pages 18-27, which is incorporated herein by reference.
Koh et al., in a paper entitled “Subharmonically Pumped CMOS Frequency Conversion (Up and Down) Circuits for 2-GHz WCDMA Direct-Conversion Transceiver,” IEEE Journal of Solid-State Circuits (39:6), June 2004, pages 871-884, which is incorporated herein by reference, describe quadrature sub-harmonic mixers that operate in quadrature double-balanced mode. The mixers require local oscillator (LO) signals with octet-phases (0°, 45°, 90°, 135°, 180°, 225°, 270° and 315°) , which are produced by an active multiphase LO generator composed of a polyphase filter and active 45° phase shifting circuits.
Polyphase networks comprise multiple inputs and multiple outputs and are designed to manipulate the input signals to provide desired polyphase output signals. Networks of this sort are sometimes used for generating polyphase signals. For example, asymmetric polyphase networks are described in the paper by Galal et al., cited above. The general behavior of asymmetric polyphase networks is discussed and analyzed, as well as their distinct property of discriminating between positive and negative sequences. A design methodology is proposed and a sensitivity analysis based on Monte Carlo simulation is presented. The effect of the absolute variation of resistance and capacitance values on the phase and gain matching, as well as on the image signal suppression, is also investigated for one-, two-, and three-stage networks.
An eight-phase phase splitter is described in U.S. Pat. No. 6,388,543, whose disclosure is incorporated herein by reference. The inventors describe an eight-phase 45° polyphase splitter with amplitude matching. A 45° split is achieved by tying together the inputs of two offset four-phase 90° phase splitters. The inventors assert that amplitude matching may be achieved by obtaining those inputs from an additional single four-phase 90° splitter.
Embodiments of the present invention provide improved methods and devices for generating polyphase signals. In some embodiments, these signals are used as local oscillator (LO) signals for driving a pair of sub-harmonic mixers in a quadrature configuration.
Currently-available phase generators, such as the polyphase splitter described in U.S. Pat. No. 6,388,543 cited above, suffer from several limitations. For example, the phase-shifting networks used in such phase generators typically change their phase and amplitude response as a function of the load impedances applied to their outputs. Conventional phase generator configurations are also typically sensitive to component value variations that occur in IC (Integrated Circuit) fabrication processes. As a result, currently-available phase generator configurations are limited to narrow-band applications and process-corrected applications.
In embodiments of the present invention, the polyphase signal is produced by a novel polyphase network comprising phase-shifting elements arranged in a circularly-symmetrical configuration, as will be described below. The polyphase network converts one or more sinusoidal reference signals into a polyphase sequence of signals, having any phase increment of the form 360/n. Several phase generator stages may be cascaded to provide wide operation bandwidth and robustness to RFIC process variations.
Several circuit implementations of the phase-shifting elements are provided, including the use of an inverted reference signal to replace a 180° phase shift. This configuration offers lower insertion loss in comparison with alternative designs.
Embodiments of the present invention thus provide a balanced, equal-amplitude phase generator design that is robust in the presence of RFIC fabrication process variations and changing load conditions. Such a design is useful in a variety of applications, such as sub-harmonic quadrature mixers.
There is therefore provided, in accordance with an embodiment of the present invention, a phase generator including:
at least one input port for accepting a reference signal having a frequency;
a plurality of n output ports, n>4; and
a network of phase-shifting elements coupled in a circularly-symmetrical configuration having a symmetry increment of 360/n degrees between the at least one input port and the output ports, so as to generate at the output ports respective polyphase signals at the frequency, the polyphase signals having respective phases that are consecutive integer multiples of 360/n degrees.
In a disclosed embodiment, n=8.
In a disclosed embodiment, the network of phase-shifting elements includes resistor-capacitor (RC) circuits. Additionally or alternatively, the network of phase-shifting elements is implemented in an integrated circuit.
In another embodiment, the network of phase-shifting elements includes a first set of the elements providing a first phase shift, and a second set of the elements providing a second phase shift, the second phase shift differing by (360/n)-180 degrees from the first phase shift at the frequency. Additionally or alternatively, the elements in the first set include an inverter followed by a 360/n degree phase shifter, providing a combined phase shift of (360/n)-180 degrees at the frequency.
In yet another embodiment, the network of phase-shifting elements includes a cascaded connection of two or more networks of the phase-shifting elements.
In still another embodiment, the phase generator includes one or more pluralities of n buffer amplifiers coupled between the two or more networks of the phase-shifting elements. In another embodiment, each of the two or more networks has a zero at a different, respective frequency.
In another embodiment, the at least one input port includes multiple input ports, which are coupled to receive a set of real reference signals.
In still another embodiment, the phase generator includes one or more voltage-dividers for deriving the real reference signals from an external input signal.
There is also provided, in accordance with an embodiment of the present invention, a sub-harmonic quadrature mixer including:
a phase generator including:
at least one input port for accepting a reference signal having a frequency;
a set of eight output ports; and
a network of phase-shifting elements coupled between the at least one input port and the output ports, so as to generate at the output ports respective polyphase signals at the frequency, the polyphase signals having respective phases that are consecutive integer multiples of 45 degrees; and
a first mixer for converting a first RF (radio frequency) signal at an input frequency to a second RF signal at an output frequency, the first mixer accepting a first set of LO signals, including two or more of the eight polyphase signals having respective phases that are integer multiples of 90 degrees; and
a second mixer for converting the first RF signal at the input frequency to the second RF signal at the output frequency, the second mixer accepting a second set of LO signals, including two or more of the eight polyphase signals having respective phases that are integer multiples of 90 degrees, the respective phases of the second set differing by 45 degrees from the respective phases of the first set.
There is additionally provided, in accordance with an embodiment of the present invention, a method for signal generation including:
accepting one or more reference signals having a frequency; and
processing the reference signals using a network of phase-shifting elements arranged in a circularly-symmetrical configuration having a symmetry increment of 360/n degrees so as to generate a plurality of n output ports, n>4, respective polyphase signals at the frequency, the polyphase signals having respective phases that are consecutive integer multiples of 360/n degrees.
There is further provided, in accordance with an embodiment of the present invention, a method for sub-harmonic quadrature mixing, the method including:
accepting one or more reference signals having a frequency;
processing the reference signals using a network of phase-shifting elements so as to generate eight polyphase signals at the frequency, the polyphase signals having respective phases that are consecutive integer multiples of 45 degrees; and
converting a first RF (radio frequency) signal at an input frequency to a second RF signal at an output frequency, by:
mixing the first RF signal in a first mixer with a first set of two or more of the eight polyphase signals having respective phases that are integer multiples of 90 degrees; and
mixing the first RF signal in a second mixer with a second set of two or more of the eight polyphase signals having respective phases that are integer multiples of 90 degrees, the respective phases of the second set differing by 45 degrees from the respective phases of the first set.
The present invention will be more fully understood from the following detailed description of the embodiments thereof, taken together with the drawings in which:
The mixer shown in
Quadrature mixer 10 comprises two second-order sub-harmonic mixers 12 and 14 for down-converting the In-phase (I) and Quadrature (Q) components of the signal, respectively. Mixers 12 and 14 convert the RF input signal (marked as RF IN in
Mixer 10 further comprises a frequency generation circuit 16 and a phase generator 20 for generating the required eight LO signals. In some embodiments, circuit 16 comprises a single oscillator providing a sinusoidal reference signal at the LO frequency. In other embodiments, circuit 16 comprises several phase-synchronized oscillators, generating sinusoidal reference signals having different amplitudes, as will be explained in the description of
Phase generator 20 accepts the reference signals provided by circuit 16 and manipulates them to provide a polyphase sequence of eight LO signals having a 45° phase increment, at generator outputs 24. Four LO signals (having phases 0°, 90°, 180° and 270°) are used to drive mixer 12. The remaining four signals (having phases 45°, 135°, 225° and 315°) are used to drive mixer 14. It can be seen that the two conditions mentioned above are fulfilled: each mixer is driven by a polyphase sequence having a 90° phase increment, and the two mixers are driven by sequences having a relative 45° phase offset.
In an alternative implementation of sub-harmonic quadrature mixer 10, mixers 12 and 14 are each driven by two LO signals in anti-phase. The 45° phase offset between the LO signals driving mixers 12 and 14 is maintained. Thus, two LO signals having phases 0° and 180° drive mixer 12, while two LO signals having phases 45° and 225° drive mixer 14.
Before describing the inner structure of phase generator 20, the theory of polyphase sequence generation will now be explained in detail.
Phase generator 20 can be viewed as a polyphase network having eight inputs and eight outputs. The polyphase network is driven by a polyphase sequence comprising eight signals. A polyphase sequence is defined as a “clockwise sequence” if its phase increment is positive. Similarly, a polyphase sequence whose phase increment is negative is defined as an “counter-clockwise sequence,” referring to the direction of phase rotation in the complex, or polar, representation of the signals.
For example, the sequence {1,e−jπ/4,e−jπ/2,e−j3π/4,e−jπ, . . . } is a clockwise sequence. The sequence {1,ejπ/4,ejπ/2,ej3π/4,ejπ, . . . } is a counter-clockwise sequence. The direction of phase rotation is referred to as the polarity of the sequence. As described in the paper by Galal et al., cited above, a polyphase network may exhibit asymmetric response to sequences of opposite polarities. This sequence-discrimination property is used to generate polyphase sequences from a real reference signal, as will be explained below.
Phase generator 20 comprises a complex filter whose spectral response has a zero at a negative frequency (−f0), wherein f0 is equal to the LO frequency of the reference signals provided by circuit 16. In the explanation that follows, only the complex envelope of the signals is considered. The following table describes input and output signals of phase generator 20:
Inputs 22 of phase generator 20 are excited by the eight reference signals given in the second column of the table. Note that all eight reference signals are real, and their complex envelopes are therefore represented by real numbers. Each signal in the second column of the table may be mathematically represented as a superposition of two complex signals, as shown in the third column. Note that this representation comprises a superposition of two polyphase sequences—a clockwise sequence (shown separately as “sequence 1” in the fourth column) and a counter-clockwise sequence (shown separately as “sequence 2” in the fifth column).
Since phase generator 20 is a linear system, the response of the phase generator to the excitation signal (second column, or equivalently the third column) is equal to the sum of its responses to the two polyphase sequences (fourth and fifth columns) . Since the spectral response of the phase generator has a zero at frequency −f0, its response to the counter-clockwise polyphase sequence (fifth column of the table) is zero. Therefore, the response of the phase generator comprises only the clockwise sequence given in the fourth column.
To summarize, phase generator 20 is a complex filter that translates the eight real excitation signals {2, √2, 0, −√{square root over (2)}, −2, −√{square root over (2)}, 0, √{square root over (2)}} into the polyphase sequence eight sinusoidal LO signals having a π/4 (45°) phase increment. The eight input values {2, √{square root over (2)}, 0, −√{square root over (2)}, −2, −√{square root over (2)}, 0, √{square root over (2)}} give the relative amplitudes of eight phase-synchronized sinusoidal reference signals that should be provided by frequency generation circuit 16 in order to produce the desired eight LO signals at outputs 24 of phase generator 20. Several exemplary configurations of circuit 16 providing the required excitation signals are given in the description of
The 135° phase difference between elements 26 and 28 causes the phase generator to attenuate the counter-clockwise polyphase sequence, as described above. Assume that a counter-clockwise sequence at the LO frequency having a 45° phase increment is applied to inputs 22 of the phase generator. Consider the signal at one of outputs 24 of the phase generator, for example OUT 3 in
The 135° phase-shifting difference between elements 26 and 28 is closely-related to the 45° phase increment of the polyphase sequence. The structure of phase generator 20 described herein may be generalized to produce polyphase sequences having any phase increment of the form 360°/n, wherein n is an arbitrary integer. In the general case, the phase-shifting difference between elements 26 and 28 is (180-360/n)° at the LO frequency. The example described herein assumes n=8. The generalization to other values of n may be used to produce quadrature sub-harmonic mixers of higher harmonic orders.
A circular-symmetric configuration is beneficial, inter alia, since it produces equal amplitude output signals, independent of the loading of the respective outputs, as long as the load impedances are equal (in other words—as long as the load is also circularly-symmetric).
In
Note that the two alternative circuit implementations of elements 26 and 28 in
Phase generator 56 is implemented in a similar fashion to phase generator 20. The configuration of phase generator 55 varies according to the implementation of frequency generation circuit 16. In some embodiments, circuit 16 is configured to produce all eight reference signals, having the amplitudes and phases described above, and to provide these reference signals to the phase generator. Such an implementation of circuit 16 is shown in
In other embodiments, circuit 16 produces only one external reference signal. In this case, phase generator 55 is similar to phase generator 20 (and 56), but also comprises voltage-dividers for generating the eight reference signals from the external reference signal. This configuration is explained in the description of
In some applications it is beneficial to connect two or more phase generators in cascade in this manner. For example, each of the cascaded phase generators may have a zero at a different frequency. This configuration provides phase generation (in other words—attenuation of the counter-clockwise polyphase sequence) over a larger bandwidth, and also allows for process errors in the RFIC fabrication. In a typical CMOS process, for example, the actual values of the resistance-capacitance product (RC) may vary by as much as ±30%. This variation is translated into a similar variation in the spectral zero of the phase generator (frequency −f0). Having several cascaded phase generators causes the design shown in
Optionally, the outputs of phase generator 55 are amplified using a set of buffer amplifiers 57, before driving the inputs of phase generator 56. Positioning the buffer amplifiers between the phase generator stages reduces the DC current consumption of the amplifiers, in comparison to a configuration in which the amplifiers are located at the output of the last stage in the cascade. Positioning amplifiers 57 between the stages also reduces the output noise level of the cascade. A low noise level is particularly desirable in some applications of polyphase phase generators, such as receiving antenna beam-forming networks.
While
In an alternative design of circuit 16, the eight reference signals having amplitudes {1, 1/√{square root over (2)}, 0, −1/√{square root over (2)}, −1, −1/√{square root over (2)}0, 1/√{square root over (2)}} Volts may be produced using a single oscillator 58 that provides a single external reference signal and its inverse (180° phase-shifted replica) to phase generator 20. In this configuration circuit 16 comprises a single oscillator 58 having an amplitude of 1 Volt. The ±1/√{square root over (2)} amplitudes are produced by using voltage dividers (shown in detail in
Although phase generators are described in this patent application as part of an LO source driving a quadrature sub-harmonic mixer, the principles of this phase generation method may also be used in other applications requiring polyphase signal generation. Such applications include phase shifting circuits for antenna beam-forming and MPSK (M-ary Phase Shift Keying) signal generation circuits, for example.
It will thus be appreciated that the embodiments described above are cited by way of example, and that the present invention is not limited to what has been particularly shown and described hereinabove. Rather, the scope of the present invention includes both combinations and sub-combinations of the various features described hereinabove, as well as variations and modifications thereof which would occur to persons skilled in the art upon reading the foregoing description and which are not disclosed in the prior art.
