Piezoelectric cross-sectional Lamé mode transformer

Abstract

A transformer including a piezoelectric plate and interdigital electrodes is provided. The interdigitated electrodes includes a plurality of conductive strips disposed over the piezoelectric plate. A cross-sectional Lamé mode resonance is excited in a cross sectional plane of the piezoelectric plate in response to input voltage applied through the interdigitated electrode, producing a voltage gain. A device including the aforementioned transformer is also provided.

Description

BACKGROUND

Micro Electro Mechanical (MEM) resonators are key enablers for the development of miniaturized and low-power multi-band radio-frequency (RF) systems capable of operating in the crowded modern commercial and military spectral environment. The use of micro- and nano-electromechanical systems to form low-loss passive filters presents challenges as a result of their still low Figure of Merit. Because of this reason, pre-existing resonator technologies, such as Surface Acoustic Wave (SAW), are still preferred despite the impossibility of monolithically integrating them on the same chip than the rest of the electronic components.

SUMMARY

In view of the foregoing, the present inventors have recognized and appreciated the advantages of a transformer with the features described herein to be employed in future RF components including micro- and nano-electromechanical systems.

In one aspect, the instant invention provides a piezoelectric transformer comprising:

a piezoelectric layer having first and second opposed surfaces extending in a length direction and a width direction, and having a thickness, T, between the opposed surfaces;

an input portion comprising:

• • an input interdigital electrode comprising a plurality of input conductive strips disposed on the surfaces of the piezoelectric layer, each of the plurality of input conductive strips extending in the length direction and having a width, W_in, extending in the width direction, and
  • an input port configured to receive an input signal and in electrical communication with the input interdigital electrode; and

an output portion comprising:

• • an output interdigital electrode comprising a plurality of output conductive strips disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips, each of the plurality of output conductive strips extending in the length direction and having a width, W_out, extending in the width direction, and
  • an output port configured to transmit an output signal and in electrical communication with the output interdigital electrode;

wherein the input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port.

In another aspect, the instant invention provides a method of making the above-described piezoelectric transformer, the method comprising lithographically patterning the input interdigital electrode and the output interdigital electrode in contact with the surfaces of the piezoelectric layer to provide a desired resonant frequency range.

Yet another aspect of the invention is a method of producing a voltage gain comprising: (a) providing the transformer of claim 1; and (b) operating the transformer to provide the voltage gain.

Another aspect of the invention is a method of using a cross-sectional Lamé mode resonator as a voltage-transformer.

Yet another aspect of the invention is a micro-electromechanical device including the above-described transformer.

The invention can also be summarized with the following list of embodiments.

1. A micro-electromechanical piezoelectric transformer comprising:

a piezoelectric layer having first and second opposed surfaces extending in a length direction and a width direction, and having a thickness, T, between the opposed surfaces;

an input portion comprising:

• • an input interdigital electrode comprising a plurality of input conductive strips disposed on the surfaces of the piezoelectric layer, each of the plurality of input conductive strips extending in the length direction and having a width, W_in, extending in the width direction, and
  • an input port configured to receive an input signal and in electrical communication with the input interdigital electrode; and

an output portion comprising:

• • an output interdigital electrode comprising a plurality of output conductive strips disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips, each of the plurality of output conductive strips extending in the length direction and having a width, W_out, extending in the width direction, and
  • an output port configured to transmit an output signal and in electrical communication with the output interdigital electrode;

wherein the input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port.

2. The transformer of embodiment 1, wherein the piezoelectric layer comprises at least one of aluminum nitride, lithium niobate, lithium tantalate, zinc oxide, gallium nitride, scandium nitride, and quartz.

3. The transformer of embodiment 1, wherein the piezoelectric layer comprises aluminum nitride.

4. The transformer of embodiment 1, wherein the input portion and the output portion are asymmetrical, the asymmetry configured to produce the voltage gain.

5. The transformer of embodiment 1, wherein the plurality of input conductive strips are interdigitated with the plurality of output conductive strips with a pitch, P, in the width direction, and the pitch is so configured to maximize an electromechanical coupling coefficient for the resonator

6. The transformer of embodiment 5, wherein the pitch is in a range of about 50 nm to about 100 μm.

7. The transformer of embodiment 5, wherein the thickness of the piezoelectric layer is about equal to the pitch.

8. The transformer of embodiment 5, wherein the thickness of the piezoelectric layer is equal to the pitch within about 0.5%.

9. The transformer of embodiment 1, wherein the resonant frequency of the transformer is lithographically defined.

10. The transformer of embodiment 1, wherein input and output portions are electrically asymmetrical.

11. The transformer of embodiment 10, wherein an effective capacitance at the input port differs from an effective capacitance at the output port.

12. The transformer of embodiment 1, wherein the input port and the output port are mechanically asymmetrical.

13. The transformer of embodiment 12, wherein the input conductive strips and the output conductive strips differ in one or more of strip size, width, length, and area.

14. The transformer of embodiment 1, wherein an acoustic wavelength in the width direction is about equal to the thickness of the peizoelectric layer.

15. The transformer of embodiment 1, wherein the voltage gain is equal to or larger than about 100 for quality factors greater than about 2000.

16. The transformer of embodiment 1, wherein the thickness of the piezoelectric layer is from about 50 nanometers to about 100 micrometers.

17. The transformer of embodiment 1, wherein e31 and e33 piezoelectric coefficients of the piezoelectric layer are coherently combined to excite the two-dimension al Lamé mode of vibration.

18. The transformer of embodiment 1, wherein a frequency of the Lamé mode of vibration ranges from about 1 MHz to about 100 GHz.

19. A method of making the piezoelectric transformer of embodiment 1, the method comprising, lithographically patterning the input interdigital electrode and the output interdigital electrode in contact with the surfaces of the piezoelectric layer to provide a desired resonant frequency range.

20. A method of producing a voltage gain comprising:

(a) providing the transformer of embodiment 1;

(b) operating the transformer to provide the voltage gain.

21. Use of a cross-sectional Lamé mode resonator as a voltage-transformer.

22. A device comprising the transformer of embodiment 1.

23. The device of embodiment 22, wherein the device is an AC-DC converter or a DC-DC converter.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a schematic representation of thickness field excitation (TFE) for cross-sectional Lamé mode resonators (CLMRs) and degenerate cross-sectional Lamé mode resonators (dCLMRs), according to one embodiment.

FIG. 1B is a schematic representation of lateral field excitation (LFE) for CLMRs and dCLMRs, according to one embodiment.

FIG. 2 depicts the distribution of the resonance frequency (f_r) of a CLMR as a function of the acoustic wavelength along the lateral direction (λ_x), according to one embodiment.

FIG. 3 depicts distributions of displacements and simulated through finite element analysis (FEA) along cut lines (BB′ and BB″) in the cross-section of a CLMR.

FIG. 4 depicts piezoelectric coupling coefficient k_t^2FEM simulated through two-dimensional FEA, according to one embodiment.

FIG. 5 depicts k_t^2FEM for TFE CLMRs of different T_AlN/λ_x and mode number n values, according to one embodiment.

FIG. 6 depicts k_t^2FEM for LFE CLMRs of different T_AlN/λ_x and mode number n values, according to one embodiment.

FIG. 7 compares resonance frequencies f_r simulated through FEA with resonance frequencies obtained analytically for TFE dCLMRs, as the acoustic wavelength varies, according to one embodiment.

FIG. 8 compares resonance frequencies f_r simulated through FEA with resonance frequencies obtained analytically for LFE dCLMRs, as the acoustic wavelength varies, according to one embodiment.

FIG. 9 depicts the resonance frequency f_r simulated through FEA as the coverage of the interdigitated electrode(s) a varies, for TFE CLMRs and LFE CLMRs, according to one embodiment.

FIG. 10 depicts the he distribution of the k_t^2FEM is shown as the coverage of the interdigitated

FIG. 11 depicts the steps of fabricating a TFE CLMR by using a four-mask microfabrication process, according to one embodiment.

FIG. 12 depicts an SEM image of a fabricated AlN TFE CLMR, according to one embodiment.

FIG. 13 depicts the distribution of admittance amplitude as a function of frequency for the AlN TFE CLMR of FIG. 12.

FIG. 14 depicts the simulated k_t², the simulated admittance Y, and the simulated mode-shape of total displacement for a TFE CLMR, according to one example.

FIG. 15 depicts the simulated k_t², the simulated admittance Y, and the simulated mode-shape of total displacement for a LFE CLMR, according to one embodiment.

FIG. 16 depicts the steps of fabricating a LFE CLMR by using two-mask microfabrication process, according to one embodiment.

FIG. 17 depicts a scanning electron microscope (SEM) image of a fabricated AlN LFE CLMR, according to an embodiment.

FIG. 18 depicts the distribution of admittance amplitude as a function of frequency for the AlN LFE CLMR of FIG. 17.

FIG. 19 depicts a schematic representation of a 2-port cross-sectional Lamé mode transformer (CLMT).

FIG. 20 depicts the simulated open-circuit voltage gain of CLMTs assuming a quality factor of 5000, a pitch of the IDTs equal to 5 μm and when varying the thickness of the AlN plate between 1 μm to 7 μm.

FIG. 21A depicts an equivalent circuit model of a CLMT.

FIG. 21B depicts the Thevenin's equivalent used to compute the voltage _out at the input of the electrical transformer in FIG. 22A.

FIG. 21C depicts a 2-port Y-matrix representation adopted to compute _out/V_in and G_v from the circuit shown in FIG. 22B.

FIG. 22A depicts simulated G_v^(max) as a function of W and assuming a Q_load-value equal to 1000.

FIG. 22B depicts simulated frequency of operation (f_res) as a function of W and assuming a Q_load-value equal to 1000.

FIG. 23 shows 2D-FEA simulated Y_in- and G_v- as functions of the frequency (f) of the input signal, V_in, and assuming a Q_load-value equal to 1000.

FIG. 24 shows a scanned-Electron-Microscope picture of a fabricated CLMT.

FIG. 25 shows measured Y_in relative of the fabricated CLMT (red); analytically predicted Y_in using the Q_load, C_in, C_out, k_t² and η values measured experimentally (black). Measured G_v of the fabricated CLMT (orange); and analytically predicted G_v (blue).

DETAILED DESCRIPTION OF THE INVENTION

Following below are more detailed descriptions of various concepts related to, and embodiments of, micro-electromechanical resonators and methods of producing the same. It should be appreciated that various concepts introduced above and discussed in greater detail below may be implemented in any of numerous ways, as the disclosed concepts are not limited to any particular manner of implementation. Examples of specific implementations and applications are provided primarily for illustrative purposes.

MEM resonators have been researched for their ability to attain high quality factors (Q) and large electromechanical coupling coefficients (k_t²) in small volumes, thereby achieving desirable Figure of Merit. As used herein, k_t² is a measure of the conversion efficiency between electrical and acoustic energy in piezoelectric materials; Figure of merit is the product of the quality factor (Q) and k_t². The Figure of Merit may directly determine the motional resistance in a resonator, impact oscillator design by setting the required gain (i.e., power consumption) and phase noise of oscillator, and impact filter design by setting insertion loss in properly terminated filters and device bandwidth.

Aluminum Nitride (AlN) Film-Bulk-Acoustic resonators (FBAR) can replace off-chip surface acoustic wave (SAW) devices in commercial products, hence significantly reducing the form-factor of the RF front-ends. The AlN FBAR technology relies on the e₃₃ piezoelectric coefficient of AlN to transduce resonant vibration along the thickness of an AlN plate. Since the device resonance frequency (f_r) is set by the thickness of the AlN plate (T_AlN), the resonance frequency cannot be tuned lithographically. Thus, integration of multi-frequency FBAR based filters on a same chip may increase the fabrication complexity (i.e., by mass-loading or trimming).

AlN contour-mode resonator (CMR) technology allows lithographical tuning of the resonance frequency. The AlN CMR relies on the e₃₁ piezoelectric coefficient of AlN to transduce resonant vibration along an in-plane direction of an AlN plate (e.g., width extensional or length extensional motion). The lithographically set lateral dimension of the device determines the resonance frequency of the device. Therefore, CMRs operating in the Ultra-High- (UHF) and Very-High- (VHF) frequency ranges can be fabricated on a same chip. However, the electromechanical coupling coefficient (k_t²) of AlN CMRs is lower than that of FBARs due to the intrinsically lower amplitude of the e₃₁ piezoelectric coefficient compared to the e₃₃. Thus, FBAR-based filters are still preferred to the CMR-based ones for the implementation of low insertion loss and wideband passive filtering networks.

Cross-Sectional Lamé Mode Resonators

Resonators disclosed herein are a new class of AlN MEM resonators based on the piezoelectric transduction of a Lamé mode or a degenerate Lamé mode in the cross-section of an AlN plate, called cross-sectional Lamé mode resonators (CLMRs) or degenerate cross-sectional Lamé mode resonators (dCLMRs). The CLMRs and dCLMRs rely on a coherent combination of the e₃₁ and e₃₃ piezoelectric coefficients of AlN to transduce a two dimensional (2D) mechanical mode of vibration, which is characterized by longitudinal vibrations along both the width and the thickness of the AlN plate. For the CLMRs, the peak-to-peak displacement along the width direction is the same as the peak-to-peak displacement along the thickness directions. For the dCLMRs, the peak-to-peak displacements along the width direction and the thickness direction are different. CLMRs and dCLMRs can achieve high values of electromechanical coupling coefficient, for example, as high as 7%. In addition, since such a 2D mode of vibration depends on the lateral dimensions of the plate, CLMRs/dCLMRs operating at different resonance frequencies can be lithographically defined on a same substrate without requiring additional fabrication steps. Thus, both high electromechanical coupling coefficient and integration of multi-frequency and low insertion-loss filters on a same chip can be achieved without additional costs and fabrication complexity.

Referring to FIG. 1A, a schematic representation of thickness field excitation (TFE) for CLMR/dCLMR is shown, according to one embodiment. The CLMR/dCLMR may be part of nano-electromechanical or micro-electromechanical based RF systems to be used for applications such as sensing and communication. The TFE CLMR/dCLMR in FIG. 1A includes a piezoelectric plate 111, a top interdigitated electrode 112 disposed on a top surface of the plate 111, and a bottom interdigitated electrode 116 disposed on a bottom surface of the plate 111. The plate 111 may have any suitable dimensions. Depending on the geometry of the resonator, the term “dimension” may refer to any dimension of interest in the resonator. For example, the dimension may refer to thickness, width, height, length, diameter, radius, etc. According to one embodiment, the plate may have a thickness of less than or equal to about 10 microns e.g., less than about 9 microns, about 8 microns, about 7 microns, about 6 microns, about 5 microns, about 4 microns, about 3 microns, about 2 microns, about 1 micron, about 900 nm, about 800 nm, about 700 nm, about 600 nm, about 500 nm, about 400 nm, about 300 nm, about 200 nm, about 100 nm, or less. According to another embodiment, the plate is a nano-plate, referring to a plate whose largest dimension is in the nanometer range, such as less than or equal to about 1 micron, or any of the aforedescribed values with respect to dimension. According to another embodiment, nano-plate may refer to a plate with at least one dimension in the nanometer range, such as less than or equal to about 1 micron, or any of the aforedescribed values with respect to dimension. According to another embodiment, the plate is a micro-plate, referring to a plate whose largest dimension is in the micrometer range, such as less than or equal to about 1 mm. According to another embodiment, micro-plate may refer to a plate with at least one dimension in the micrometer range, such as less than or equal to about 1 mm.

The plate 111 may be made of any suitable piezoelectric material. According to one embodiment, the plate 111 may include a compound, such as a nitride, such as an aluminum nitride (AlN). According to another embodiment, the plate may include at least one of aluminum nitride, lithium niobate, lithium tantalate, zinc oxide, gallium nitride, scandium nitride, and quartz.

The top electrode 112 and the bottom electrode 116 of the resonator may be made of any suitable material. According to one embodiment, the electrodes 112 and 116 may include a metal, such as a noble metal, such as platinum or gold. In some embodiments, the top electrode 112 and the bottom electrode 116 are in direct contact with the piezoelectric plate 111. In some embodiments, at least one of the top electrode 112 and the bottom electrode 116 is not in directed contact with the piezoelectric plate 111. The top electrode 112 may include multiple conductive strips, each strip having a width W_e and arranged with a pitch p from adjacent strips (i.e., p is the combination of the width of a strip and the width of the space between two adjacent strips). The bottom electrode 116 has an identical pattern as the top electrode 112. As shown, two electric field components can be excited simultaneously, E₁ along the lateral direction and E₂ along the thickness direction. E₁ has field lines connecting adjacent conductive strips forming the top interdigitated electrode 112. E₂ has field lines connecting two conductive strips facing each other, one conductive strip from the top interdigitated electrode 112 and the other from the bottom interdigitated electrode 116.

Referring to FIG. 1B, a schematic representation of lateral field excitation (LFE) for CLMR/dCLMR is shown, according to one embodiment. The LFE CLMR/dCLMR in FIG. 1B includes a piezoelectric plate 121 and a top interdigitated electrode 124 disposed on a top surface of the plate 121. The plate 121 is similar to the pate 111 of FIG. 1A. The top electrode 124 is similar to the top electrode 112 of FIG. 1A. Different than the TFE CLMR/dCLMR, the LFE CLMR/dCLMR does not include a bottom electrode. Therefore, electrical field component E₁ along the lateral direction can be excited but not electrical field component along the thickness direction. As will be discussed in greater detail below, LFE CLMR/dCLMR may simplify the fabrication of the device at the expense of a lower k_t², comparing to TFE CLMR/dCLMR. Experiments demonstrated a k_t² of about 2.5% and a quality factor Q of about 1900 for a LFE CLMR/dCLMR operating at 2.8 GHz, resulting in a k_t²Q product (i.e. Figure of Merit (FoM)) close to 46.

Principle of Operation of Cross-Sectional Lamé Mode Resonators (CLMRs)

When an alternating voltage is applied to the interdigitated electrode(s), a cross-sectional Lamé mode of vibration may be excited in the resonators shown in FIGS. 1A-B. Starting from the equations of motion in solids, the Lamb wave theory can identify all modes of vibration that can be described by a one dimensional (1D) displacement vector in a plate. Thus, the modes of vibration can be characterized by a predominant displacement along one geometrical dimension of the plate, whose size determines the frequency of operation. Different from 1D modes of vibration, CLMRs/dCLMR shows comparable displacements along two dimensions, for example, displacement along the lateral direction () and displacement along the thickness direction () of the plate 111 or 121 of FIG. 1. For the CLMRs, the peak-to-peak displacement along the lateral direction is the same as the peak-to-peak displacement along the thickness directions. For the dCLMRs, the peak-to-peak displacements along the lateral direction and the thickness direction are different. Equation (1) is the general expression of displacements along the lateral () and thickness () directions for the CLMR:

$\begin{array}{cc}\left[\begin{array}{c}\\ \end{array}\right]=\left[\begin{array}{c}A\left(x\right)B\left(z\right)\\ C\left(x\right)D\left(z\right)\end{array}\right]=\left[\begin{array}{c}\cos \left({\beta }_{x}x\right)\sin \left({\beta }_{z}z\right)\\ -\sin \left({\beta }_{x}x\right)\cos \left({\beta }_{z}z\right)\end{array}\right],& \left(1\right)\end{array}$ wherein β_x and β_z are the wave-vectors relative to the motion along the x- and z-directions.

The resonance frequency (f_r) of the CLMR can be obtained by solving the equations of motion (Equations (2) and (3)), which describe the distribution of the x- (μ_x) and z-displacements (μ_z) in an AlN plate, with proper boundary conditions.

$\begin{array}{cc}{C}_{11}\frac{d^{″}\left({\mu }_x\right)}{{\mathrm{dx}}^2}+C_{55}\frac{d^{″}\left({\mu }_x\right)}{{\mathrm{dz}}^2}+2C_{15}\frac{d^{″}\left({\mu }_x\right)}{\mathrm{dxdz}}+\left(C_{55}+C_{13}\right)\frac{d^{″}\left({\mu }_z\right)}{\mathrm{dxdz}}=\rho ·{\left(2\pi f_r\right)}^2·{\mu }_x,& \left(2\right)\\ C_{15}\frac{d^{″}\left({\mu }_x\right)}{{\mathrm{dx}}^2}+C_{55}\frac{d^{″}\left({\mu }_z\right)}{{\mathrm{dx}}^2}+C_{33}\frac{d^{″}\left({\mu }_z\right)}{{\mathrm{dz}}^2}+\left(C_{55}+C_{13}\right)\frac{d^{″}\left({\mu }_x\right)}{\mathrm{dxdz}}=\rho ·{\left(2\pi f_r\right)}^2·u_z,& \left(3\right)\end{array}$ wherein C_ij are components of the stiffness matrix of the piezoelectric material of which the plate is made (e.g., AlN) and ρ is the mass density of the piezoelectric material.

By setting μ_x and μ_z in Equations (2) and (3) to and in Equation (1), respectively, Equations (2) and (3) are simplified as:C₁₁β_x²−(C₁₃+C₅₅)β_xβ_z+C₅₅β_z²=ρ·(2πf_r)²  (4)C₃₃β_z²−(C₁₃+C₅₅)β_xβ_z+C₅₅β_x²=ρ·(2πf_r)²  (5).

Equations (4) and (5) have four sets of (,) solution, but only one set corresponds to positive wave-vectors along both vibrational directions, hence enabling the excitation of the cross-sectional Lamé mode (CLM) in the resonator. The set of (,) is:

$\begin{array}{cc}=\sqrt{\frac{B{\rho \left(2\pi f_r\right)}^2}{C_{11}{C}_{33}-C_{55}^2+\sqrt{\mathrm{AB}}}},& \left(6\right)\\ =\sqrt{\frac{\left(C_{11}-C_{55}\right)}{\left(C_{33}-C_{55}\right)}},& \left(7\right)\end{array}$ wherein A=(C₁₁−C₅₅)(C₁₃+C₅₅)² and B=(C₃₃−C₅₅).

All sides of AlN CLMRs behave as stress-free boundaries. Thus, and satisfy the following boundary conditions:

$\begin{array}{cc}=\frac{n·\pi }{W_{\mathrm{AlN}}},& \left(8\right)\\ =\frac{m·\pi }{T_{\mathrm{AlN}}},& \left(9\right)\end{array}$ wherein n is the mode number along the x-direction, m is the mode number along the z-directions, W_AlN is the width of the AlN plate, which is equivalent to n·p (See FIGS. 1A-B), and T_AlN is the thickness of the AlN plate. Analysis below discusses the vibration mode with the minimum vibrational order allowed by both TFE and LFE CLMRs (i.e. n equal to 2 and m equal to 1). In this scenario, W_AlN coincides with the acoustic wavelength along the x-direction (λ_x), and λ_x=2p.

By comparing Equation (8) and Equation (6), the resonance frequency, f_r, of the CLMR is:

$\begin{array}{cc}{f}_r=\frac{1}{{\lambda }_x}\sqrt{\frac{C_{11}{C}_{33}-C_{55}^2-\left(C_{13}+C_{55}\right)\sqrt{\left(C_{11}-C_{55}\right)\left(C_{33}-C_{55}\right)}}{\left(C_{33}-C_{55}\right)\rho }}.& \left(10\right)\end{array}$

By combining Equations (7) through (9), the following λ_x/T_AlN ratio should be satisfied in order to enable excitation of the CLM in the AlN plate:

$\begin{array}{cc}\frac{{\lambda }_x}{T_{\mathrm{AlN}}}=2\sqrt{\frac{\left(C_{11}-C_{55}\right)}{\left(C_{33}-C_{55}\right)}}.& \left(11\right)\end{array}$

FIG. 2 illustrates the distribution of the resonance frequency f_r as a function of the acoustic wavelength λ_x. The curve in FIG. 2 represents the analytically found distribution; the dots represent the simulated f_r value through finite element analysis (FEA) as λ_x varies between 1 and 10 μm. T_ALN has been adjusted, for each λ_x value, to satisfy Equation (11). FIG. 2 also shows the geometry used to compute f_r and the two λ_x values experimentally explored (i.e., λ_x=3 μm for the “LFE experiment” and λ_x=8 μm for the “TFE experiment”). The analytically found distribution of f_r with respect to λ_x shows excellent agreement with the simulated trend through FEA.

FIG. 3 depicts distributions of and simulated through FEA along cut lines (BB′ and BB″) in the cross-section of a CLMR. In the simulation, a W_AlN of 1.5 μm (mode number n=1) is used and a resonator quality factor Q of 2500 is assumed. The displacement reaches the maximum at (x,z)=(0.75 μm, 0.75 μm). As expected, for CLMR, the simulated distributions of and show the same peak-to-peak value (See Equation (1)) along cut lines BB′ and BB″.

The resonator may be configured to resonate at any appropriate frequency. According to one embodiment, the resonator resonates at a frequency of at least about 10 MHz, e.g., at least about 50 MHz, about 100 MHz, about 200 MHz, about 300 MHz, about 400 MHz, about 500 MHz, about 600 MHz, about 700 MHz, about 800 MHz, about 900 MHz, or higher. According to another embodiment, the resonator resonates at a frequency of about 10 MHz to about 100 GHz, e.g., about 50 MHz to about 90 GHz, about 100 MHz to about 80 GHz, about 200 MHz to about 70 GHz, about 300 MHz to about 60 GHz, about 400 MHz to about 50 GHz, etc.

For MEM piezoelectric resonators, piezoelectric coupling constant (K²) identifies the maximum electromechanical coupling coefficient (k_t²) for a specific mode of vibration. k_t² is a measure of the conversion efficiency between electrical and acoustic energy in piezoelectric resonators. In particular, k_t² represents the portion of the electrical energy, stored in the resonator's static capacitance (C₀), which is converted into mechanical motion (i.e., proportional to the ratio between motional and static capacitances, C_m/C₀). k_t² of a resonator can be directly extracted from its electrical response as:

$\begin{array}{cc}{k}_t^2=\frac{{\pi }^2}{8}\frac{f_p^2-f_s^2}{f_s^2}=\frac{{\pi }^2}{8}\frac{C_m}{C_0},& \left(12\right)\end{array}$ wherein f_p is the parallel resonance frequency, and f_s is the series resonance frequency of the resonator.

The K² of a first order CLMR (i.e., fundamental Cross-Sectional Lamé mode of vibration, n and m both equal to 1) can be analytically derived as discussed below. Since CLMRs displace along both lateral and thickness directions, their piezoelectric coupling coefficient is the sum of two components. One component (K_x²) is produced by the lateral motion (i.e., ) whereas the second component (K_z²) is originated from the thickness vibration (i.e., ). Since the electric field is exclusively distributed across the thickness of the AlN plate (z-oriented), the K² associated with each of the two displacement components of the CLM can be computed by solving its corresponding Christoffel equation. The so found K² value (K_Chr²) is an effective measure of the actual piezoelectric coupling coefficient if the mode-shape is uniform along the directions orthogonal to the vibrational one. In this scenario, K_eff² can be approximated as:K_eff²=K_Chr²∫_−T/2^T/2B(z)dz  (13),

If the resonator were to displace along the x-direction, with displacement distribution, μ_x, not uniform along the z-direction (i.e. μ_x=cos(β_xx)·B(z)), its effective K² (K_eff²) would be lower than K_Chr². In the CLM case where and are not uniform along thickness and width of the AlN plate, K_x² and K_z² are computed as:

$\begin{array}{cc}{K}_x^2=\frac{K_{\mathrm{Chr}\left[x\right]}^2}{T}{\int }_0^T\sin \left({\beta }_{z}z\right)\mathrm{dz}=\frac{2}{\pi }{K}_{\mathrm{Chr}\left[x\right]}^2,& \left(14\right)\\ K_z^2=\frac{K_{\mathrm{Chr}\left[z\right]}^2}{p}{\int }_0^p\sin \left({\beta }_{x}x\right)\mathrm{dx}=\frac{2}{\pi }{K}_{\mathrm{Chr}\left[z\right]}^2,& \left(15\right)\end{array}$ wherein K_Chr[x]² and K_Chr[z]² are the piezoelectric coupling coefficients derived from Christoffel equation, respectively, for the lateral and vertical longitudinal vibration components of the CLM.

K_Chr[x]² and K_Chr[z]² are computed as:

$\begin{array}{cc}{K}_{\mathrm{Chr}\left[x\right]}^2=\frac{{e_{31}}^2}{\varepsilon ·C_{\mathrm{lat}}},& \left(16\right)\\ K_{\mathrm{Chr}\left[z\right]}^2=\frac{{e_{33}}^2}{\varepsilon ·C_{\mathrm{thic}}}=\frac{{e_{33}}^2}{\varepsilon ·C_{\mathrm{lat}}}\sqrt{\frac{\left(C_{11}-C_{55}\right)}{\left(C_{33}-C_{55}\right)}},& \left(17\right)\end{array}$ wherein C_lat is the effective stiffness coefficients of the CLMR along the x-direction, C_thic is the effective stiffness coefficient of CLMRs along the z-directions, e₃₁ is an AlN piezoelectric coefficient equal to −0.58 C/m², e₃₃ is another AlN piezoelectric coefficient equal to 1.55 C/m², and ε is the AlN dielectric permittivity.

Applying Equations (6) and (7), C_lat and C_thic are:

$\begin{array}{cc}{C}_{\mathrm{lat}}=\frac{C_{11}{C}_{33}-{C_{55}}^2}{\left(C_{33}-C_{55}\right)}+\frac{\left(C_{13}+C_{55}\right)\sqrt{\left(C_{11}-C_{55}\right)\left(C_{33}-C_{55}\right)}}{\left(C_{33}-C_{55}\right)}=320\mathrm{GPa},& \left(18\right)\\ C_{\mathrm{thic}}=C_{\mathrm{lat}}\frac{\left(C_{33}-C_{55}\right)}{\left(C_{11}-C_{55}\right)}=300\mathrm{GPa}.& \left(19\right)\end{array}$

Therefore, for CLMs excited in AlN plates, K² (K²_CLM) is:

$\begin{array}{cc}{K}_{\mathrm{CLM}}^2=\frac{2}{\pi }\left(K_{\mathrm{Chr}\left[x\right]}^2+K_{\mathrm{Chr}\left[z\right]}^2\right)=7.1\%.& \left(20\right)\end{array}$

The K²_CLM value obtained from Equation (20) can be verified by the piezoelectric coupling coefficient k_t², which is extracted from its simulated admittance (See Equation (12)), through 2D-Finite Element Analysis (FEA), as shown in FIG. 4. A schematic representation of the specific CLMR geometry used in the simulation is also shown in FIG. 4. In particular, a CLMR characterized by a W_AlN of 1.5 μm and a T_ALN of 1.45 μm (See Equation (11)) is used. A resonator length (L) equal to 50 μm and no damping is assumed. In addition, the presence of the IDTs is neglected by exciting the devices through infinitesimal excitation boundaries. As evident from A schematic representation of the specific CLMR geometry used in the simulation is also shown in FIG. 4, the FEM computed electromechanical coupling coefficient k_t^2FEM, closely matches the value derived through the simplified analytical model (See Equation (20)). The validity of Equation (20) is hence confirmed.

A comparison between k_t^2FEM and the previously reported K² values for AlN FBARs and AlN CMRs is shown in Table 1.

TABLE 1
Comparison of piezoelectric coupling coefficient of different types.
Technology K2
AlN FBARs  7%
AlN CMRs   2%
AlN CLMRs  6.4%

A. Higher-order CLMs in AlN Thin Plates

The performance of a first order CLMR (i.e., fundamental Cross-Sectional Lamé mode of vibration, n and m both equal to 1) is discussed above. In order to reach small values of input impedance, piezoelectric resonators often recur to higher-order vibrational modes (i.e. n>1) excited through the use of conductive interdigitated structures (IDTs). For a conventional S₀ mode Lamb-wave, the pitch of the IDTs used to excite the S₀ mode Lamb-wave resonators may define the acoustic wavelength and, consequently, the frequency of operation. However, when higher order CLMs are excited in AlN plates, the resonance frequency and electromechanical performance depend on both λ_x and T_AlN (See Equation (11)). In particular, when the resonance frequency of the Lamb-wave S₀ mode propagating along the lateral direction (i.e. x) matches the resonance frequency of the longitudinal mode propagating along the thickness direction (i.e. z), a large electromechanical coupling coefficient can be expected through the excitation of a CLM in the AlN plate. This phenomenon can be studied, through Finite Element Analysis, by extracting k_t^2FEM of the S₀ mode for different T_AlN/λ_x values and for a representative set of n-values. This can be done through the identification, for each investigated case, of the metallized (f_m) and the non-metallized (f_o) frequencies. Once f_m and f_o are found, k_t^2FEM can be obtained through Equation (12) by replacing f_s with f_m and f_p with f_o.

Referring to FIG. 5, k_t^2FEM according to Equation (12) for TFE CLMRs of different T_AlN/λA_x and n values is shown. In calculating k_t^2FEM, the presence of the metallic IDTs is neglected. Curve 501 corresponds to k_t^2FEM changing with T_AlN/λ_x for mode number n=2; curve 502 for n=5; curve 503 for n=15, curve 504 for n=40; and curve 505 for n→∞. In simulating the case related to n→∞, periodic boundary conditions are applied to both displacement and electric fields, at the lateral edges of a second order CLMR (i.e., W_AlN=λ_x).

Referring to FIG. 6, k_t^2FEM according to Equation (12) for LFE CLMRs of different T_AlN/λ_x and n values is shown. In calculating k_t^2FEM, the presence of the metallic IDTs is neglected. Curve 601 corresponds to k_t^2FEM changing with T_AlN/λ_x for mode number n=2; curve 602 for n=5; curve 603 for n=15, curve 604 for n=40; and curve 605 for n→∞. In simulating the case related to n→∞, periodic boundary conditions are applied to both displacement and electric fields, at the lateral edges of a second order CLMR (i.e., W_AlN=λ_x).

As shown, a maximum k_t^2FEM value (k_t^2max) is found at a T_AlN/λ_x that approaches the optimum value (˜0.5) enabling the excitation of a CLM in the AlN plate (See Equation (11)), for both the TFE and the LFE.

For the case of TFE shown in FIG. 5, k_t^2max reduces as n increases. This reduction originates from the increase of the parasitic capacitance produced by the lateral component of the modal distribution of the electric field. In particular, k_t² for a higher-order vibrational mode (N), k_t^2n=N, can be related to k_t² for n equal to 1,k_t²ⁿ⁼¹. The relationship between k_t^2n=N and k_t²ⁿ⁼¹ is as follows.

$\begin{array}{cc}{k}_t^{2^{n=N}}\propto \frac{N·C_m^{n=1}}{{C_0}^{n=N}}=\hspace{1em}\frac{N·C_m^{n=1}}{N·{C_0}^v+\left(N-1\right)·{C_0}^l}=k_t^{2^{n=1}}·\frac{1}{1+\frac{C_0^{\mathrm{Lat}}·\left(N-1\right)}{N·{C_0}^v}},& \left(21\right)\end{array}$ wherein C_mⁿ⁼¹ is the resonator motional capacitance when n is equal to 1, C₀^n=N is the resonator static capacitance when n is set to N, C₀^v and C₀^Lat are the capacitances associated with the electric field modal distributions along the thickness and lateral directions, respectively. According to Equation (21), the negative impact of C₀^Lat on the device kt2 is minimum when N is equal to 1 and reaches maximum when N approaches infinity. In particular, when N→∞, k_t^2n=N can be rewritten as:

$\begin{array}{cc}{k}_t^{2^{N\to \infty }}\propto k_t^{2^{n=1}}·\frac{{C_0}^v}{{C_0}^v+C_0^{\mathrm{Lat}}}.& \left(22\right)\end{array}$

It is worth noting that the value of C₀^Lat depends on T_AlN/λ_x. In particular, C₀^Lat is minimum for T_AlN/λ_x→0 while increases for larger T_AlN/λ_x values.

Nevertheless, when T_AlN/λ_x→0.5, C₀^Lat is still smaller than C₀^v. Therefore, boosting of the k_t² associated with the excitation of the CLM in the structure is not significantly compromised. In addition, the T_AlN/λ_x value at which the k_t² value reaches the maximum slightly lowers when n is increased. This may be due to the different sensitivity of C_lat and C_thic to C₀^lat, which originates, in piezoelectric plates, from the different piezoelectric coefficients involved in the motion along the lateral and thickness directions.

For the case of LFE shown in FIG. 6, a much lower sensitivity of k_t^2FEM with respect to n is found. This is mostly due to the fact that C₀^Lat remains significantly larger than C₀^v for all T_AlN/λ_x values and is independent of n.

Degenerate Cross-Sectional Lamé Mode Resonators (dCLMRs)

As discussed above, CLMRs/dCLMRs have displacements along two dimensions, for example, displacement along the lateral direction () and displacement along the thickness direction () of the piezoelectric plate. For CLMRs, the peak-to-peak displacement along the lateral direction is the same as the peak-to-peak displacement along the thickness directions. For dCLMRs, the peak-to-peak displacements along the lateral direction and the thickness direction are different.

CLM excitation can be enabled when the ratio of acoustic wavelength to the thickness of the piezoelectric plate (λ_x/T_AlN) satisfies Equation (11). For the minimum vibrational order, the acoustic wavelength is two times the pitch of the interdigitated electrode(s) (i.e., λ_x=2p). For dCLMRs, the ratio of the acoustic wavelength (i.e., the pitch of the interdigitated electrode(s)) to the thickness of the piezoelectric plate no longer satisfies Equation (11). dCLMRs can be formed by stationary motion along the thickness and lateral directions of the plate and are possible for a much wider range of T_AlN/λ_x values. The mode degeneracy is enabled by the special dispersive properties of AlN that allow the lateral coupling of each conductive strip of the IDTs that vibrates along the thickness direction. When the coupling happens, stationary waves occur along both directions of the plate and consequently, the electromechanical performance and resonance frequency depend on the motion along both directions.

Equation (10) shows that as the value of acoustic wavelength (i.e., the pitch) changes, the resonance frequency would change accordingly. Thus, the dCLMRs may achieve various resonance frequencies by varying the pitch of the interdigitated electrode(s) in lithographic processes without the need to change the thickness of the piezoelectric plate. As discussed in greater detail below, it is possible for dCLMRs to achieve k_t² values that are comparable to the best k_t² values attained in the non-degenerate CLMRs in a broad range of lithographically defined operating frequencies (i.e., different λ_x-values for a given T_AlN). Thus, it is possible to use dCLMRs to achieve, simultaneously, high k_t² and a lithographic tunability of f_r.

As discussed above, The mode degeneracy is enabled by the special dispersive properties of AlN that allow the lateral coupling of each conductive strip of the IDTs that vibrates along the thickness direction. The dispersion equation for two-dimensional modes of vibration relates the wave-vectors relative to both directions (See Equations (8) and (9)) to the resonance frequency as follows:

$\begin{array}{cc}\frac{f_r}{v_{\mathrm{eq}}}=\sqrt{+},& \left(23\right)\end{array}$ wherein v_eq is an equivalent sound velocity for the two-dimensional motion. Equation (23) can be used to describe the resonance frequency, f_r, of degenerate CLMRs as a function of λ_x. To do so, v_eq is calculated. v_eq can be estimated as

$v_{\mathrm{eq}}\sim \sqrt{\frac{2C_{\mathrm{thic}}}{4\rho }}$ if the small difference of sound velocities relative to the motion along the two directions is neglected. f_r in Equation (23) may be set equal to the resonance frequency of non-degenerate CLMRs in Equation (10). Although Equation (10) is obtained for CLMRs in which Equation (11) is strictly satisfied. The validity of such approximation is valid for dCLMRs because the difference between C_lat and C_thic is small (See Equations (18) and (19)). Assuming that v_eq is almost independent of

$\frac{T_{\mathrm{AlN}}}{{\lambda }_x}$ for a limited range of

$\frac{T_{\mathrm{AlN}}}{{\lambda }_x}$ values around the optimum (˜0.5), f_r can be calculated as:

$\begin{array}{cc}{f}_r\approx \frac{1}{2T_{\mathrm{AlN}}}·\sqrt{1+{\left(\frac{2T_{\mathrm{AlN}}}{{\lambda }_x}\right)}^2}·\sqrt{\frac{2C_{\mathrm{thic}}}{4\rho }}.& \left(24\right)\end{array}$

Referring to FIG. 7, resonance frequencies f_r simulated through FEA is compared with resonance frequencies obtained analytically for TFE dCLMRs, as the acoustic wavelength λ_x varies. Curve 701 represents the simulated f_r, and curve 702 represents the analytical f_r with the presence of interdigitated electrode(s) neglected in both approaches. The thickness of the AlN plate (T_AlN) is 4 μm, which is the same value as used in the TFE experiment in FIG. 2. λ_x varies between about 7 μm and about 11 μm. It can be seen from FIG. 7 that the analytical f_r closely matches the simulated f_r. FIG. 7 also shows that degenerate TFE CLMRs achieve a lithographic frequency shift of ˜350 MHz around 1.2 GHz by varying λ_x from about 7 μm to about 11 μm. Such variation does not lead to substantial decrease of k_t².

Referring to FIG. 8, resonance frequencies f_r simulated through FEA is compared with resonance frequencies obtained analytically for LFE dCLMRs, as the acoustic wavelength λ_x varies. Curve 801 represents the simulated f_r, and curve 802 represents the analytical f_r with the presence of interdigitated electrode(s) neglected in both approaches. The thickness of the AlN plate (T_AlN) is 1.5 μm, which is the same value as used in the LFE experiment in FIG. 2. λ_x varies between about 2.4 μm and about 3.6 μm. It can be seen from FIG. 8 that the analytical f_r closely matches the simulated f_r. FIG. 8 also shows that degenerate LFE CLMRs achieve a lithographic frequency shift of ˜670 MHz around 3.2 GHz by varying λ_x from about 2.4 μm to about 3.6 μm. Such variation does not lead to substantial decrease of k_t².

Frequency Tuning Through Metal Coverage Variation

The resonance frequency of the CLMRs may be tuned by changing the coverage (i.e., α=W_e/p) of the interdigitated electrode(s). As used herein, the coverage of the interdigital electrode(s) refers to a ratio of the width of each conductive strip (W_e) to the pitch of adjacent conductive strips (p). As both and change along the x-direction, the equivalent mass density (ρ^(eq)) of CLMRs is a function of the coverage α. As α varies, the effective sound velocity of the CLMRs would change and, consequently, the operating frequency of the device would change. In other words, the effective sound velocity and resonance frequency of the resonators can be lithographically changed by varying W_e for a given p.

Referring to FIG. 9, resonance frequency f_r simulated through FEA is shown for two-finger (i.e., n=2) TFE CLMRs and LFE CLMRs, as the coverage of the interdigitated electrode(s) a varies. Curve 901 is for the TFE CLMRs, and curve 902 is for the LFE CLMRS. CLMRs having two fingers may not achieve a device input impedance of 50Ω. However, the input impedance of 50Ω may be achieved through the adoption of resonator arrays. In computing f_r, it is assumed that the top and bottom IDTs of the TFE CLMRs are made of platinum and each have a thickness of 0.07·T_AlN. The top IDTs of the LFE CLMRs are made of aluminum and have a thickness of 0.07·T_AlN. The simulation results show that a lithographic frequency variation of 30% can be attained on the TFE CLMRs, while the resonance frequency of the LFE is less sensitive to α. Larger frequency tuning can be attained by using thicker platinum IDTs on the TFE CLMRs.

Referring to FIG. 10, the distribution of the k_t^2FEM is shown as the coverage of the interdigitated electrode(s) a varies for TFE CLMRs and LFE CLMRs of FIG. 9. Curve 1001 is for the TFE CLMRs, and curve 1002 is for the LFE CLMRs. As discussed above, a lithographic frequency variation of ˜30% can be attained on TFE CLMRs using platinum IDTs (as in the experimental demonstration), and larger frequency tuning can be achieved if thicker platinum IDTs are used. At the same time, a k_t^2FEM larger than 5% can be achieved on the TFE CLMRs for such high frequency tuning. In contrast, for the LFE CLMRs using aluminum IDTs, both the resonance frequency and k_t^2FEM are less sensitive to α (as in our experimental demonstration). This may be due to the larger impact that platinum electrodes have on the effective sound velocity (v_eq) of the device comparing to aluminum electrodes.

Fabrication of CLMRs/dCLMRs and Experimental Results

The CLMR/dCLMRs described herein may be fabricated by appropriate microfabrication process. According to another embodiment, the resonators may be fabricated by a four-mask microfabrication process. The four-mask fabrication process may include: disposing a first electrode layer over a substrate, patterning the first electrode layer to form a bottom interdigitated electrode, disposing a piezoelectric layer over the substrate, disposing a second electrode layer over the piezoelectric layer, patterning the second electrode layer to form a top interdigitated electrode, etching the piezoelectric layer to form a piezoelectric micro-plate, and releasing the micro-plate from the substrate. The substrate may comprise, or be, any suitable material, such as silicon. According to one embodiment, the disposing of the electrode layers may include any suitable process, such as a sputter deposition process. According to one embodiment, the patterning of the electrode layers to form interdigitated electrodes may include any suitable process, such as a lift-off process. According to one embodiment, the disposing of the piezoelectric layer may include any suitable process, such as a sputter deposition process. The piezoelectric layer may include any suitable material, such as the aforedescribed piezoelectric materials. According to one embodiment, the etching of the piezoelectric layer to form a piezoelectric micro-plate may include any suitable process, such as an ion conductive plasma (ICP) process. The forming of the micro-plate may include forming a perimeter of the nano-plate. According to one embodiment, the releasing the piezoelectric layer from the substrate may include any suitable process, such as an isotropic etching process.

According to one embodiment, the resonators may be fabricated by a two-mask microfabrication process. The two-mask fabrication process may include: disposing a piezoelectric layer over a substrate, disposing an electrode layer over the piezoelectric layer, patterning the electrode layer to form an interdigitated electrode, etching the piezoelectric layer to form a piezoelectric micro-plate, and releasing the micro-plate from the substrate. The substrate may comprise, or be, any suitable material, such as silicon. According to one embodiment, the disposing of the piezoelectric layer may include any suitable process, such as a sputter deposition process. The piezoelectric layer may include any suitable material, such as the aforedescribed piezoelectric materials. According to one embodiment, the disposing of the electrode layer may include any suitable process, such as a sputter deposition process. According to one embodiment, the patterning of the electrode layer to form an interdigitated electrode may include any suitable process, such as a lift-off process. According to one embodiment, the etching of the piezoelectric layer to form a piezoelectric micro-plate may include any suitable process, such as an ion conductive plasma (ICP) process. The forming of the micro-plate may include forming a perimeter of the nano-plate. According to one embodiment, the releasing the piezoelectric layer from the substrate may include any suitable process, such as an isotropic etching process.

TFE CLMRs were fabricated and their performance was experimentally characterized. A device was formed by a 4 μm thick AlN layer and two 0.1 μm thick platinum IDTs, one IDT disposed on the top surface of the AlN layer and the other IDT disposed on the bottom surface of the AlN layer. Platinum was chosen as the bottom IDT material due to the need of growing a high quality AlN film, and as the top IDT material in order to preserve high acoustic symmetry in the cross-section of the device. The coverage of the platinum IDTs (α) was 0.5. The pitch of the IDTs setting (i.e., λ_x/2) was chosen so as to match the AlN thickness.

Referring to FIGS. 11(a)-(e), steps of fabricating TFE CLMRs are shown according to one embodiment. The LFE CLMRs were fabricated using a four-mask microfabrication process. At step (a), a 100 nm thick Pt film was deposited on top of a high resistivity Silicon wafer 1120 through a 10 nm thick Ti layer used as adhesion layer, and was patterned through a lift-off process to form the bottom IDT 1110. At step (b), a 4 μm thick AlN film 1130 was sputter-deposited on the wafer. The AlN film 1130 may be wet etched to form vias. At step (c), a 100 nm thick Pt film was deposited on the AlN film 1130 through a 10 nm thick Ti layer, and was patterned through a lift-off process to form the top IDT 1140. At step (d), the AlN film 1130 was etched by ICP in Cl₂ based chemistry to define the width of the AlN plate. This was done through the use of a hard mask made out of 2 μm of SiO₂ so as to attain steep AlN sidewall angles (>75°). At step (e), the Silicon substrate underneath the resonator was released through XeF₂ isotropic etching. The SEM image of the fabricated device is shown in FIG. 12.

The electrical performance of the fabricated TFE CLMRs were measured, in air, by connecting the device, through Ground-Signal-Probes (GSG), to a commercial Network Analyzer (Agilent E5071C). Referring to FIG. 13, the measured admittance of the device and its fitting to the Butterworth-van Dyke (BVD) model are shown—they match well. The TFE CLMR used for the measurement has a T_AlN of 4 μm and a λ_x of 8 μm. The measured admittance was de-embedded from the directly extracted parasitic capacitance due to the probing pads (11 fF). The TFE CLMR shows a main resonant peak at about 1.02 GHz, a quality factor, Q, about 600, and a k_t² larger than 4.6%. This k_t² value is higher than the k_t² of other piezoelectric resonators using IDTs to actuate and sense mechanical vibrations in AlN plates. The experimental demonstration of such a high k_t² confirms the advantages of the disclosed CLMR technology, which is enabled by the combined use of the AlN e₃₁ and e₃₃ piezoelectric coefficients.

The fabricated device showed a relatively lower quality factor than the quality factor of other AlN resonators operating in the same frequency range. It may mostly be caused by a large amount of energy lost through anchor dissipations. It is believed that a refinement of the anchor design would help maximize the quality factor attained by TFE CLMRs. It is worth noting that the selected pitch size (4 μm) did not yield the highest possible k_t² value (˜5.7% for the given material stack and mode order number). This might be due to the presence of the metallic IDTs which causes a slight variation of the optimum T_AlN/λ_x value for the optimum transduction of the CLM and, consequently, the achievement of the highest electromechanical coupling coefficient. This was confirmed in FIG. 14.

FIG. 14 shows the simulated k_t² as a function of the acoustic wavelength λ_x for the TFE CLMRs. In the simulation, AlN TFE CLMR with a thickness of 4 μm is used. The IDTs are 0.1 μm thick platinum and the coverage a is set to 0.5. From FIG. 14, the highest k_t^2FEM(k_t^2max˜5.7%) was attained at a pitch size of 5 μm (i.e. λ_x=10 μm). It is also worth noting that a k_t^2FEM higher than 4% can be achieved for a relatively large range of λ_x values, thus not only when Equation (11) is strictly satisfied. As discussed above, this fact is enabled by the capability of exciting degenerate Lamé modes in AlN plates. The k_t^2max is larger than what is predicted when the presence of the metallic IDT is neglected. This may be due to the fact that the IDTs lower both the effective stiffnesses in the lateral (C_lat) and thickness (C_thic) directions, thus effectively increasing K_Chr[x]² and K_Chr[z]². FIG. 14 also shows the simulated admittance and the simulated mode-shape of the displacement relative to two adjacent fingers for λ_x=8 μm.

LFE CLMRs were fabricated and their performance was experimentally characterized. For example, a device was formed by a 1.5 μm thick AlN layer and a 0.2 μm thick aluminum IDT disposed on the top surface of the AlN layer. Different than the fabricated TFE CLMR discussed above, aluminum instead of platinum, which has a lower resistivity and a sound velocity closer to that of AlN, was chosen to be the IDT material. The coverage of the aluminum interdigitated electrode (α) was 0.5.

FIG. 15 shows the simulated k_t² as a function of the acoustic wavelength λ_x for the LFE CLMRs. FIG. 15 also shows the simulated admittance and the simulated mode-shape of the displacement relative to two adjacent fingers for λ_x=3.2 μm. Similar to the TFE CLMRs discussed above, due to the presence of the metallic IDT used to excite the electric field in the AlN layer, the highest k_t² value is expected at a T_AlN/λ_x value that is slightly different from what was found when the presence of the metal layers is neglected. Consequently, in order to maximize k_t², λ_x is adjusted through FEA. FIG. 15 shows that maximum k_t² is attained at λ_x close to 3.2 μm.

Referring to FIG. 16(a)-(d), steps of fabricating an AlN LFE CLMR are shown according to one embodiment. The LFE CLMRs were fabricated using a two-mask microfabrication process. At step (a), a 1.5 μm thick AlN film 1650 was sputter-deposited on top of a high resistivity silicon wafer substrate 1630. At step (b), a 200 nm thick Al film was deposited on top of the AlN film and patterned through etching to define the top interdigitated electrode 1610. At step (c), the AlN film was etched by ICP in Cl₂ based chemistry to define the width of the AlN plate. At step (d), the silicon substrate underneath the AlN resonator was released through XeF₂ isotropic etching. An SEM image of the fabricated device is shown in FIG. 17.

The electrical performance of the fabricated LFE CLMR was measured, in air, by connecting the device, through Ground-Signal-Ground probes (GSG), to a commercial Network Analyzer (Agilent E5071C). Referring to FIG. 18, the measured admittance of the device and its fitting to the Butterworth-van Dyke (BVD) model are shown they match well. The LFE CLMR used for the measurement has a T_AlN of 1.5 μm and a λ_x of 3.2 μm. The LFE CLMR shows a main resonant peak at about 2.82 GHz, a quality factor, Q, about 1855, and a k_t² about 2.48% which closely matches the FEM predicted value in FIG. 15. This k_t² value is slightly larger than what is predicted when the presence of the metallic IDT is neglected. It may be due to the fact that the aluminum lowers both the effective stiffnesses in the lateral (C_lat) and thickness (C_thic) direction, thus leading to higher K_Chr[x]² and K_Chr[z]². The measured k_t² and Q results in a k_t²·Q product (i.e. Figure of Merit (FoM)) close to 46, which is higher than the FoM of any LFE AlN MEM resonators disclosed before that employed a single top IDT fabricated through a 2-mask fabrication process.

Cross-Sectional Lamé Mode Transformers and Experimental Results

To date, transformers are generally large, lossy and hardly implementable at high frequency; hence, their use in low-power integrated applications has not be taken into consideration so far. In another aspect, the instant invention provides a novel class of MicroElectroMechanical (MEM) piezoelectric transformers capable of achieving high open-circuit voltage-gain in excess of 100 (for quality factors greater than 2000) between input and output of 2-port AlN CLMRs (FIG. 19). CLMRs with two physical ports can also be made by actuating and sensing the motion through different portions of top and bottom IDTs. In particular, when such portions are formed by the same number of identical strips and are electrically terminated on equal load (Z_load), 2-port CLMRs are reciprocal and symmetric. However, when these conditions are not verified, they behave as transformers, hence showing different voltage and current levels at their input and output ports.

FIG. 19 illustrates an embodiment of a transformer 2010 with an input portion 2012 and an output portion 2014 disposed on a piezoelectric layer 2013 having first and second opposed surfaces extending in a length direction and a width direction. The piezoelectric layer (AlN in FIG. 19) has a thickness T (T_AlN in FIG. 19) between the opposed surfaces. The input portion includes an input interdigital electrode 2016 with of a plurality of input conductive strips 2018 disposed on the surfaces of the piezoelectric layer. Each of the plurality of input conductive strips extends in the length direction and has a width, W_in, extending in the width direction. The input portion 2012 also includes an input port 2020 configured to receive an input signal and in electrical communication with the input interdigital electrode. The output portion 2014 includes an output interdigital electrode 2022 with a plurality of output conductive strips 2024 disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips 2018. Each of the plurality of output conductive strips 2024 extends in the length direction and has a width, W_out, extending in the width direction. The output portion 2014 also includes an output port 2026 configured to transmit an output signal and in electrical communication with the output interdigital electrode. The input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port. The input conductive strips can be interdigitated with the output conductive stripe with a pitch P in the width direction.

The transformation mechanism is originated from mechanical or electrical asymmetries altering the sound-velocity at the output-port with respect to its value at the input port. For instance, the use of different electrical loads connected at input and output ports produces a variation in the effective capacitance that, because of the piezoelectric effect, stiffs the device at each port. As a result, the magnitude of the displacement field at the two ports is significantly different. In particular, CLMTs show a larger displacement at the input port than that at the output port. As their insertion-loss is negligible, a lower displacement value at their output port must result into a larger voltage level (V_out) than that applied at the input port (V_in). Similarly, the introduction of mechanical asymmetries, such as the use of different strip size for the input and output portions of the IDTs, would also produce a discrepancy in the effective sound velocity relative to input and output ports, hence producing a voltage-gain. Differently from any other Lamb-wave piezoelectric transformer demonstrated to date, CLMTs use the special characteristics of two-dimensional modes of vibration to achieve much larger Gv than those based on contour-modes of vibration in piezoelectric plates, which was verified through Finite Element Analysis (FEA).

FIG. 20 shows that when the pitch of the IDTs is chosen to be close to the AlN thickness, Gv increases significantly with respect to what achieved by contour-mode resonators, which operate at much lower T_AlN/λ_x values. G_v is directly proportional to the device figure of merit (FoM), which is defined as the product between quality factor (Q) and k_t², and it is maximum when the output port is left floating. A simulation was conducted of the voltage level at the output terminal of a 920 MHz CLMT formed by a 4 μm thick AlN film sandwiched between two 100 nm thick platinum IDTs. The pitch of the IDTs was fine-tuned (W˜5 μm) so as to take into consideration the change of the T_AlN/λ_x value that enables the excitation of a non-degenerate Lamé mode when metallic electrodes are also used to actuate the motion in the device.

An experiment was conducted with simulated G_v value when assuming different values of quality factor, Q. As expected, G_v increases proportionally with respect to the assumed quality factor. In particular, a G_v value as high as 140 is expected by assuming a quality factor close to 4400, hence approaching the maximum demonstrated in AlN resonators operating in the same frequency range.

The electrical behavior of a CLMT can be modelled using an equivalent circuit (FIG. 21). This circuit has been widely used to analyze the performance of previously developed piezoelectric transformers. The equivalent circuit is composed by a series combination of motional resistance (R_m), motional inductance (L_m) and motional capacitance (C_m), two shunt capacitances representing the device input and output capacitances (C_in and C_out, respectively) and an electrical transformer. The turn-ratio (η) of the electrical transformer maps the ratio between the transduction coefficients relative to the output- and the input-ports. The open-circuit voltage gain (G_v) can be defined as the ratio between V_out and V_in, when the output port is kept open.

V_out is related to the voltage ({tilde over (V)}_out) across the input-port of the electrical transformer (FIGS. 21A-B) as follows:{tilde over (V)}_out=V_out·η  (25)

To compute an analytical expression for G_v, we first expressed {tilde over (V)}_out (FIGS. 21A-B) as a function of the magnitude of the input voltage, V_in, and of the frequency, f. To do so, we replaced the electrical transformer in FIG. 21 with the Thevenin impedance seen at its input-port (FIG. 21B). The resulting circuit in FIG. 21B was then represented through a 2-port admittance matrix (Y-matrix). The use of such representation permits to easily compute the expression of {tilde over (V)}_out/V_in and, by using Eq. (25), the desired expression of G as follows:

$\begin{array}{cc}{G}_v=\frac{v_{\mathrm{out}}}{v_{\mathrm{in}}}=\frac{{\tilde{v}}_{\mathrm{out}}}{\eta ·v_{\mathrm{in}}}=\frac{Y_{21}}{\eta ·Y_{22}}=\frac{8k_t^2{\eta \left(2\pi f_{\mathrm{res}}\right)}^2{C}_{\mathrm{in}}}{\sqrt{\frac{4096{\left(2f_{\mathrm{res}}\right)}^4{C_{\mathrm{out}}}^2}{{Q_{\mathrm{load}}}^2}+{\left(8k_t^2{{\eta }^2\left(2\pi f_{\mathrm{res}}\right)}^2{C}_{\mathrm{in}}+{\pi }^2\left(-{\left(2\pi f\right)}^2+{\left(2\pi f_{\mathrm{res}}\right)}^2\right)C_{\mathrm{out}}\right)}^2}}& \left(26\right)\end{array}$

From Eq. (26), it is straightforward to find the maximum G_v-value (G_v^(max)), which is attained when f is equal to the resonance frequency (i.e. f_res) relative to the device input-admittance (Y_in). The expression of G_v^(max) is reported in Eq. (27).

$\begin{array}{cc}{G_v}^{\left(\max \right)}=\frac{k_t^2{Q}_{\mathrm{load}}{C}_{\mathrm{in}}{\mathrm{\eta \pi }}^2}{8C_{\mathrm{out}}}& \left(27\right)\end{array}$

Y_in was instead computed as:Y_in=Y₁₁−Y₂₁Y₁₂/Y₂₂  (28)where Y₁₁, Y₁₂, Y₂₁, and Y₂₂ are the coefficients of the equivalent Y-matrix (FIG. 21C) relative to the circuit shown in FIG. 21B. It is important to point out that G_v^(max) (Eq. (27)) is proportional to the value of the resonator figure-of-merit (FoM∝Q_load·k_t²), which further highlights the advantage of CLMTs over transformers based on other laterally vibrating piezoelectric resonator technologies.A. Simulated Performance of 920 MHz 3-Fingers CLMTs through Finite Element Simulations

The design of a CLMT operating around 920 MHz was conducted through 2D-FEA. In particular, the dependence of both f_res (FIG. 22A) and G_v^(max) (FIG. 22B) from the pitch of the IDT (i.e. W) was investigated. Maximum G_v^(max)-value was attained when W was set to 5.5 μm. Such W-value (W(^(tx)) permits to maximize the input-k_t². In addition, sensitivity of G_v^(max) to the width of the metal-strip (W_el^(out)) forming the output portion of the IDT was detected. Therefore, in order to attain the highest G_v^(max)-value, W_el^(out) was chosen to be about half (W_el^(out)=2 μm) of the width of the metal strips forming the input portion of the IDT (W_el⁽ⁱⁿ⁾=3.5 μm). As verified by FEA, this choice enables a˜10% improvement in the G_v^(max)-value compared to the W_el^(out)=W_el⁽ⁱⁿ⁾ case. This improvement is originated from a slight increase of the device k_t². In addition f_res was found to decrease proportionally to W (FIGS. 22A-B). Furthermore, a k_t² in excess of 4.1% was estimated by 2D-FEA (FIG. 23). Such a high k_t²-value enables a G_v^(max)-value in excess of 40 when assuming a Q_load-value equal to 1000.

B. Measured Performance of a 920 MHz 3-Finger CLMT

A SEM-picture of a fabricated CLMT is shown in FIG. 24. The performance of a 3-finger 920 MHz CLMT was characterized (FIG. 25). The device was tested, in air, through a RF probe-station using GSG-probes. Values of k_t², Q_load and G_v^(max) in excess of 4%, 1000 and 39, respectively, were directly extracted from the measured Y-matrix. A value of η˜0.23 was instead experimentally attained by shorting the device's output-port and monitoring the induced variation, Δf_res, of

$f_{\mathrm{res}}\left(\Delta f_{\mathrm{res}}=f_{\mathrm{res}}·\left(-1+\sqrt{1+\frac{8k_t^2{\eta }^2{C}_{\mathrm{in}}}{{\pi }^2{C}_{\mathrm{out}}}}\right)\right).$ Both the FEA predictions (with Q_load=1000) and the analytical model (FIG. 6 and Eqs. 2-3), based on the experimentally extracted values of k_t², Q_load, η, C_in and C_out, fit well the measured data. However, a small variation of f_res was measured with respect to its predicted value through FEA. This discrepancy is due to fabrication uncertainties associated with the actual thickness of the IDTs with respect to the value considered in the FEA.

Claims

1. A micro-electromechanical piezoelectric transformer comprising: a piezoelectric layer having first and second opposed surfaces extending in a length direction and a width direction, and having a thickness, T, between the opposed surfaces;an input portion comprising: an input interdigital electrode comprising a plurality of input conductive strips disposed on the surfaces of the piezoelectric layer, each of the plurality of input conductive strips extending in the length direction and having a width, Win, extending in the width direction, andan input port configured to receive an input signal and in electrical communication with the input interdigital electrode; andan output portion comprising: an output interdigital electrode comprising a plurality of output conductive strips disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips, each of the plurality of output conductive strips extending in the length direction and having a width, Wout, extending in the width direction, andan output port configured to transmit an output signal and in electrical communication with the output interdigital electrode;wherein the input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port.

2. The micro-electromechanical piezoelectric transformer of claim 1, wherein the piezoelectric layer comprises at least one of aluminum nitride, lithium niobate, lithium tantalate, zinc oxide, gallium nitride, scandium nitride, and quartz.

3. The micro-electromechanical piezoelectric transformer of claim 1, wherein the piezoelectric layer comprises aluminum nitride.

4. The micro-electromechanical piezoelectric transformer of claim 1, wherein the input portion and the output portion are asymmetrical, the asymmetry configured to produce the voltage gain.

5. The micro-electromechanical piezoelectric transformer of claim 1, wherein an acoustic wavelength in the width direction is about equal to the thickness of the piezoelectric layer.

6. The micro-electromechanical piezoelectric transformer of claim 1, wherein the voltage gain is equal to or larger than about 100 for quality factors greater than about 2000.

7. The micro-electromechanical piezoelectric transformer of claim 1, wherein the thickness of the piezoelectric layer is from about 50 nanometers to about 100 micrometers.

8. The micro-electromechanical piezoelectric transformer of claim 1, wherein e31 and e33 piezoelectric coefficients of the piezoelectric layer are coherently combined to excite the two-dimensional Lamé mode of vibration.

9. The micro-electromechanical piezoelectric transformer of claim 1, wherein a frequency of the Lamé mode of vibration ranges from about 1 MHz to about 100 GHz.

10. A method of making the micro-electromechanical piezoelectric transformer of claim 1, the method comprising, lithographically patterning the input interdigital electrode and the output interdigital electrode in contact with the surfaces of the piezoelectric layer to provide a desired resonant frequency range.

11. The micro-electromechanical piezoelectric transformer of claim 1, wherein a resonant frequency of the micro-electromechanical piezoelectric transformer is lithographically defined.

12. The micro-electromechanical piezoelectric transformer of claim 1, wherein input and output portions are electrically asymmetrical.

13. The micro-electromechanical piezoelectric transformer of claim 12, wherein an effective capacitance at the input port differs from an effective capacitance at the output port.

14. The micro-electromechanical piezoelectric transformer of claim 1, wherein the input port and the output port are mechanically asymmetrical.

15. The micro-electromechanical piezoelectric transformer of claim 14, wherein the input conductive strips and the output conductive strips differ in one or more of strip size, width, length, and area.

16. The micro-electromechanical piezoelectric transformer of claim 1, wherein the plurality of input conductive strips are interdigitated with the plurality of output conductive strips with a pitch, P, in the width direction, and the pitch is so configured to maximize an electromechanical coupling coefficient for the resonator.

17. The micro-electromechanical piezoelectric transformer of claim 16, wherein the pitch is in a range of about 50 nm to about 100 μm.

18. The micro-electromechanical piezoelectric transformer of claim 16, wherein the thickness of the piezoelectric layer is about equal to the pitch.

19. The micro-electromechanical piezoelectric transformer of claim 16, wherein the thickness of the piezoelectric layer is equal to the pitch within about 0.5%.

20. A method of producing a voltage gain comprising: (a) providing a transformer comprising:a piezoelectric layer having first and second opposed surfaces extending in a length direction and a width direction, and having a thickness, T, between the opposed surfaces;an input portion comprising: an input interdigital electrode comprising a plurality of input conductive strips disposed on the surfaces of the piezoelectric layer, each of the plurality of input conductive strips extending in the length direction and having a width, Win, extending in the width direction, andan input port configured to receive an input signal and in electrical communication with the input interdigital electrode; andan output portion comprising: an output interdigital electrode comprising a plurality of output conductive strips disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips, each of the plurality of output conductive strips extending in the length direction and having a width, Wout, extending in the width direction, andan output port configured to transmit an output signal and in electrical communication with the output interdigital electrode;wherein the input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port; and(b) operating the transformer to provide the voltage gain.

21. A device comprising a transformer, the transformer comprising: a piezoelectric layer having first and second opposed surfaces extending in a length direction and a width direction, and having a thickness, T, between the opposed surfaces;an input portion comprising: an input interdigital electrode comprising a plurality of input conductive strips disposed on the surfaces of the piezoelectric layer, each of the plurality of input conductive strips extending in the length direction and having a width, Win, extending in the width direction, andan input port configured to receive an input signal and in electrical communication with the input interdigital electrode; andan output portion comprising: an output interdigital electrode comprising a plurality of output conductive strips disposed on the surfaces of the piezoelectric layer and interdigitated with the plurality of input conductive strips, each of the plurality of output conductive strips extending in the length direction and having a width, Wout, extending in the width direction, andan output port configured to transmit an output signal and in electrical communication with the output interdigital electrode;wherein the input portion and the output portion are configured to excite a two-dimensional Lamé mode of vibration in the piezoelectric layer in response to an input voltage applied to the input port to produce a voltage gain at the output port.

22. The device of claim 21, wherein the device is an AC-DC converter or a DC-DC converter.