Patent Application
20130241656
Subject matter described herein relates generally to radio frequency (RF) circuits and, more particularly, to techniques and circuits for linearizing the operation of RF amplifiers.
As is known in the art, a radio frequency (RF) amplifying device (such as a power amplifier used in RF systems, for example) has a first amplifying region in which signals are linearly amplified (often referred to as the amplifier's “linear region”) and a second amplifying region in which the amplifying device exhibits nonlinear characteristics (often referred to as the amplifier's “non-linear region” or “saturation region”).
When operating in its non-linear region, the amplifying device causes distortion in the phase and amplitude of an output signal. For example, an amplifier operating in its non-linear region may generate inter-modulation products. Such distortion is not desirable in most applications as it can lead to a degradation in performance of a system which includes the amplifier.
Such distortion components may be reduced or even eliminated by operating the amplifier in its linear region. One problem with the approach of operating the amplifier in its linear region, is that the amplifier may have only a limited range of RF input signal power levels over which it provides linear amplification. Furthermore, amplifiers are much more efficient (e.g., in terms of power added efficiency) when they are operating at or near their non-linear region, but less efficient when operating in their linear region. Thus, the approach of operating the amplifier in its linear region can be quite limiting and not appropriate for many RF applications.
Power amplifiers, for example, often operate near their saturation region where amplifiers work at the maximum efficiency and thus may exhibit strong nonlinear characteristics. Thus, in order to maximize output power and efficiency, the gains and phases of power amplifier output signals are distorted.
Consequently, power amplifiers often utilize linearization techniques to compensate for nonlinear characteristics of a power amplifier. So-called “feed-forward” linearizers and “pre-distortion” linearizers have been conventionally proposed.
In the case of a power amplifier using feed-forward linearization, signals are dividedly applied to a main path and a sub-path, and carrier signals (or a tone signal and its corresponding signals) on the main path are amplified to a predetermined level by a main amplifier as the power amplifier and then output.
Intermodulation signals of the main amplifier are selectively output by a 3 dB hybrid coupler and attenuated to a predetermined level by an attenuator. The 3 dB hybrid coupler offsets the attenuated signals and signals that are applied to the sub-path and delayed via a first delay loop, so that the intermodulation signals are synthesized.
The resulting signals that are synthesized by the 3 dB hybrid coupler are applied to an error amplifier so that errors of the synthesized signals are corrected and the corrected signals are amplified. Thereafter, the corrected and amplified signals are amplified on the main path and synthesized with signals, which are delayed by a predetermined time via a second delay loop, and output. In the synthesization process, intermodulation distortion (IMD) signals are offset and output.
Meanwhile, in the case of a power amplifier using a pre-distortion linearizer, an applied carrier signal is pre-distorted beforehand by a predetermined pre-distorter. The pre-distorted signal is amplified to a predetermined level by a main amplifier and output. In other words, a pro-distorted signal is generated beforehand and offset by a pre-distorted signal portion of an applied signal, and the remaining portion of the applied signal is amplified and output. In general, the power amplifier using the pre-distortion linearizer can have a small and lightweight structure with a broad bandwidth and a wide operating range at low cost.
While conventional techniques have been somewhat effective, they have utilized relatively complicated circuits and techniques and are relatively expensive to implement both in terms of dollar cost and manpower cost.
In accordance with the concepts, systems, circuits, and techniques described herein, a power amplification system includes a radio frequency (RF) linearization circuit driving a power amplifier. The linearization circuit is operative for shaping the input signal of the power amplifier before it reaches the amplifier input. In at least one implementation, the linearization circuit may be configured to increase the relative magnitude of higher power portions of the input signal and decrease the relative magnitude of lower power portions of the input signal in a manner that provides an overall increase in the linearity of the power amplification system. In addition, the linearization circuit may be implemented in a relatively simple, inexpensive fashion. In some embodiments, a Doherty amplifier is used as the power amplifier of the power amplification system. It has been found that the use of a Doherty amplifier in conjunction with a linearization circuit provides enhanced operating characteristics when amplifying signals using modulation schemes having high peak to average power ratios (PAPRs). In at least one embodiment, a preamplifier is used in connection with the linearization circuit to compensate for losses in the circuit.
In some embodiments, the linearization circuit comprises two parallel RF signal paths. The first path is a passive path that does not include any transistor amplifiers and the second path is an active path that includes an odd number of transistor amplifiers. A divider splits an input signal into two signal components and delivers one signal component to each RF signal path. The signal components propagate through the respective signal paths and are then combined in an output combiner. Since the active path includes an odd number of transistor amplifiers, the path maintains a relatively constant phase shift of 180 degrees over frequency plus a small amount of delay caused by the transistor and amplifier circuit. A delay line may be used in the non-amplification signal path to equalize the delay in the amplification path caused by the transistor and amplifier circuit, while maintaining the 180 degree phase shift between the first and second signal paths. Since the 180 degree phase shift is accomplished by the transistor(s), the 180 degree relation is maintained in a broad frequency band. When the output signals of both paths are combined in the combiner, the signals are partially or fully cancelled in a broad frequency band due to 180 degree phase difference.
The amount of cancellation in the combiner may depend on factors such as the gain of the amplifier path, which is determined by the amplifier gain, splitting/combining losses, attenuation in the amplification path, and any phase offsets between the paths that vary from 180 degrees. The linearizer may be designed with the signal from the amplification path being equal to the signal from non-amplification path when the amplifier is exhibiting the largest gain under small signal conditions. Thus, when low level input signals are being processed, a maximum amount of cancellation may be achieved in the combiner. When larger input signals are applied to the linearizer, the amplifier(s) in the active path may operate in saturation, reducing the gain thereof. In this case, the signal from the amplification path becomes much smaller than the signal from the non-amplification path and the linearizer will exhibit a reduced amount of cancellation. As a result, the overall linearizer may demonstrate a controlled gain expansion with input signal, which compensates the typical gain saturation characteristics of power amplifiers, when the linearizer is used as a pre-distorter. When the linearizer is used in conjunction with a power amplifier, a power amplifier may result that provides a relatively linear response over a relatively wide range of input power levels and a relatively wide range of RF frequencies.
In accordance with one aspect of the concepts, systems, circuits, and techniques described herein, a power amplification system comprises a Doherty power amplifier to amplify an input signal received at the input port to generate an amplified signal at the output port, the Doherty power amplifier including a carrier amplifier and a peaking amplifier coupled in a parallel arrangement, wherein both the carrier amplifier and the peaking amplifier contribute to RF signal amplification during peak portions of the input signal and only the carrier amplifier contributes to RF signal amplification during non-peak portions of the input signal; and a linearizer circuit coupled to the input port of the Doherty power amplifier to shape the input signal before it reaches the Doherty power amplifier, the linearizer circuit to increase the relative magnitude of higher power portions of the input signal and to decrease the relative magnitude of lower power portions of the input signal so that the power amplification system operates in a more linear manner than the Doherty power amplifier alone.
In one embodiment, the power amplification system further comprises a preamplifier coupled to the linearizer circuit to compensate for losses in the linearizer circuit.
In one embodiment, the preamplifier has an output that is operatively coupled to an input of the linearizer circuit.
In one embodiment, the preamplifier is located between the linearizer circuit and the Doherty power amplifier.
In one embodiment, the power amplification system is configured for use with modulation schemes having a high peak to average power ratio (PAPR).
In one embodiment, the power amplification system operates with a power added efficiency (PAE) of 30 percent or better when 64-QAM signals are being amplified.
In one embodiment, the Doherty power amplifier further comprises: a divider circuit to divide the input signal into first and second signal components for delivery to the carrier amplifier and the peaking amplifier, respectively; and a combiner circuit to combine output signals of the carrier amplifier and the peaking amplifier.
In one embodiment, the carrier amplifier is configured as a class B or class AB power amplifier and the peaking amplifier is configured as a class C power amplifier.
In accordance with another aspect of the concepts, systems, circuits, and techniques described herein, a power amplification system comprises: a Doherty power amplifier having an input port and an output port, the Doherty power amplifier to amplify an input signal received at the input port to generate an amplified signal at the output port, the Doherty power amplifier including a carrier amplifier and a peaking amplifier coupled in a parallel arrangement, wherein both the carrier amplifier and the peaking amplifier contribute to RF signal amplification during peak portions of the input signal and only the carrier amplifier contributes to RF signal amplification during non-peak portions of the input signal; and a linearizer circuit to shape the input signal before it reaches the Doherty power amplifier, the linearizer circuit comprising: a linearizer input port to receive an unshaped input signal; a linearizer output port to output a shaped input signal for delivery to the input port of the Doherty amplifier; a divider circuit coupled to the linearizer input port to divide the unshaped input signal into a first signal component and a second signal component; a first radio frequency (RF) signal path to process the first signal component, the first RF signal path comprising a passive delay line having a predetermined length; a second RF signal path to process the second signal component, the second RF signal path comprising an odd number of transistor amplifier stages; and a combiner circuit to combine output signals of the first and second RF signal paths to generate a combined signal at an output port of the combiner circuit, wherein the output port of the combiner circuit is coupled to the linearizer output port.
In one embodiment, an insertion phase of the first RF signal path is approximately 180 degrees different from an insertion phase of the second RF signal path within a frequency band of interest.
In one embodiment, the output signals of the first and second RF signal paths at least partially cancel each other within the combiner circuit.
In one embodiment, the first signal component and the second signal component output by the divider circuit are substantially in phase with one another.
In one embodiment, the first signal component and the second signal component output by the divider circuit are out of phase with one another by M degrees, wherein the combiner circuit is adapted to compensate for the M degree phase difference generated by the divider so that the output signals of the first and second RF signal paths are substantially 180 degrees out of phase when combined.
In one embodiment, the first RF signal path includes one or more adjustable phase shifters to adjust an insertion phase of the first RF signal path.
In one embodiment, the second RF signal path includes one or more adjustable phase shifters to adjust an insertion phase of the second RF signal path.
In one embodiment, the second RF signal path includes one or more adjustable attenuators to adjust a signal level within the second RF signal path.
In one embodiment, the power amplification system further comprises a preamplifier coupled to the linearizer circuit to compensate for losses in the linearizer circuit.
In one embodiment, the power amplification system further comprises a digital processor to adjust one or more controllable elements within the linearizer circuit based, at least in part, on an indication of non-linear content within an output signal of the Doherty amplifier.
In one embodiment, the digital processor is configured to adjust at least one of: one or more adjustable phase shifters within the first RF signal path, one or more adjustable phase shifters within the second RF signal path, one or more adjustable attenuators within the second RF signal path, or one or more amplifier bias levels associated with the transistor amplifier stages of the second RF signal path.
The foregoing features may be more fully understood from the following description of the drawings in which:
Referring now to
Linearizer circuit 12 includes a divider circuit 15 having an input coupled to the linearizer input and having a pair of outputs. Divider circuit 15 splits the RF input signal RFIN into first and second signal components to be applied to first and second RF signal paths 16, 18, respectively. Divider circuit 15 may be provided, for example, as an RF coupler (e.g., a 0 degree, 90 degree, or 180 degree coupler), an RF splitter (e.g., a Wilkinson type power divider, etc.), or any other type of circuit for dividing signals into multiple components.
RF signal path 16 is a non-amplification (passive) signal path and RF signal path 18 is an amplification (active) signal path. Each of the signal paths 16, 18 has an output coupled to a respective input of combiner circuit 19. The combiner circuit 19 has an output coupled to the output of the linearizer circuit 12 and thus to the input of RF amplifier 14. Divider circuit 15 and combiner circuit 19 may each include either a symmetric device/system (e.g., a 3 dB hybrid, etc.) or an asymmetric device/system (e.g., a 10 dB coupler, etc.).
Amplifying device 10 further includes a combiner 32 which receives a portion of the RF input signal RFIN and a portion of the RF output signal RFOUT. It should be appreciated that the RF input signal RFIN provided to combiner 32 is substantially “clean” (i.e., without sidebands), while the RF output signal RFOUT provided to combiner 32 includes sidebands 33 generated as a result of amplifier 14 being operated in or near its non-linear region.
Combiner 32 subtracts the clean RF input signal RFIN from the RF output signal RFOUT to isolate sidebands 33. The isolated sideband signals 33 are provided to a signal processing unit 34 (or more simply processor 34) that in some embodiments finds an optimum condition for attenuation and bias voltage settings and/or phase shifter settings of linearizer circuit 12. These conditions can be found by, for example, searching methods where the settings are varied to achieve the minimum sideband signal power.
As will be described in detail below in conjunction with
Referring now to
It should also be appreciated that a delay line delays a signal fed thereto by a specified time delay. This implies that the phase shift of a delay line is a linear function of frequency. A microstrip transmission line is an approximation to a delay line since the phase shift of a microstrip line is not precisely a linear function of frequency (i.e., microstrip lines are dispersive). A delay line does not introduce an impedance transformation in the transmission path. It should thus be appreciated that there are many ways to implement delay lines over limited bandwidths such as with lumped element L-C networks for example as used herein, the phrase “delay line” will include transmission line structures that approximate a delay line.
It should further be appreciated that portions, or in some cases all, of the delay provided in non-amplification signal path 16 may be provided by other circuit components in path 16 (e.g., phase adjuster circuits, amplitude adjustment circuits such as attenuators, etc.). In such cases, the delay line is said to be “absorbed” or “integrated” into the other circuit component(s).
Amplification signal path 18 may include a transistor amplifier 22 consisting of an odd number of gain stages and a first variable attenuator 21 coupled between the input of the second RF signal path 18 and an input of transistor amplifier 22. Transistor amplifier 22 may have a bias circuit 24 coupled thereto. Amplification signal path 18 may further include a phase adjustor 26 and a second variable attenuator 28 coupled between an output of transistor amplifier 22 and the output of the second RF signal path 18.
It should be appreciated that phase shifters placed in one or both of the amplifying and non-amplifying signal paths may be used to fine-adjust the relative phases of signals that are combined at the combiner 19. Nominally, the signals may be set opposite (180 degrees) in phase to cancel when combined. Introducing a small offset in the relative phase shift will bring the phase cancellation angle to change with the degree of cancellation. As a result, a small offset from 180 degrees will introduce positive or negative AM/PM characteristics depending upon the direction of the offset. Adjustment of this offset by phase shifters in both paths may be used to match the AM/PM of the linearizer to that of the RF amplifier, which may be positive or negative.
By utilizing a transistor amplifier consisting of an odd number of gain stages, the amplifier provides an odd multiple of a 180 degree phase shift to RF signals plus a small amount of additional delay. Thus, the RF signals at the outputs of signal paths 16, 18 are 180 degrees out of phase, after the small delay is corrected by, for example, a delay line in one of the paths 16, 18.
In one embodiment, linearizer circuit 12 includes a divider circuit 15 which creates a nominal 90 degrees of phase shift between the two RF output signals. In such an embodiment, the combiner circuit 19 may receive two substantially opposite phase RF signals with an additional nominal 90 degrees of phase shift and combine the signals while cancelling the 90 degree phase shift. In some embodiments, a 90 degree phase shifter may be included in one of the paths 16, 18 to account for the 90° phase shift. In the case where couplers are used for the divider circuit 15 and the combiner circuit 19 that provide an inherent 90 degree phase shift (or have embedded 90 degree phase shifters that generate the phase shift between the paths), there is no need for an additional phase shifter. If the divider 15 is a 90 degree hybrid coupler and the combiner 19 is a Wilkinson signal combiner, for example, then a 90 degree phase shifter may be added to one of the paths 16, 19 to obtain the desired phase relationship between the two paths. Other arrangements are also possible.
In at least one embodiment, the bias circuit 24 includes circuitry for adjusting a bias voltage level applied to the transistor amplifier 22 and circuitry for adjusting an attenuation level of at least one of the first and second attenuators 21, 28 placed before and after the amplifier 22. In one embodiment, the circuitry for adjusting the attenuation level comprises circuitry for adjusting the attenuation level in conjunction with a change in bias voltage applied to the amplifier 22.
In one embodiment, at least one of the first and second attenuators 21, 28 are provided as electronically tunable attenuators controllable in conjunction with a change in a voltage level of a bias voltage applied to the amplifier 22. In one embodiment, linearizer circuit 12 includes circuitry to electronically adjust a phase shift of at least one of the first and second RF signal paths 16, 18.
In one embodiment, the linearizer circuit 12 further includes circuitry for electronically adjusting an attenuation level of either or both of the first and second RF signal paths 16, 18 and circuitry for electronically adjusting a phase shift of either or both of the first and second RF signal paths. For example, in some embodiments processor 34 may serve to electronically adjust attenuation levels and phases of either or both of RF signal paths 16, 18.
In one embodiment, processor 34 calculates an optimum condition from an RF signal at the linearizer output and provides one or more control signals to circuitry for electronically adjusting an attenuation level of either or both of the first and second RF signal paths 16, 18. Processor 34 may also provide one or more control signals to circuitry for electronically adjusting a phase shift of either or both of the first and second RF signal paths 16, 18. In one embodiment, processor 34 is configured to generate signals reflective of one or more environmental conditions (e.g., weather, temperature, humidity, etc.) and to use the signals to calculate the optimum condition.
As mentioned above in conjunction with
Referring now to
Amplification signal path 18 may include a transistor amplifier 22 that includes an odd number of transistor stages, a phase adjuster 92, a first attenuator 94 disposed prior to transistor amplifier 22 and a second attenuator 96 disposed after transistor amplifier 22. One or more delay lines 98 (illustrated in
The transistor amplifier may also have a bias circuit coupled thereto (not shown in
As shown in
It may be desirable to measure the amplitude and/or phase of the sideband signals of the RF output signal and thus the cancellation circuit 100 may provide a cancellation signal to a detector 102. The detector 102 may detect an amplitude and/or phase of the cancellation signal provided thereto and generate a detector signal which can be used to adjust attenuators and/or phase shifters in both the amplification and non-amplification signal paths 16, 18 of the linearizer to reduce (or in some cases, eliminate) the sidebands in the RF output signal which give rise to the cancellation signal.
The detector 102 may provide the detector signal to a processor 104 which may be the same as or similar to processor 34 discussed above in connection with
In at least one embodiment, attributes of one or both of the amplification and non-amplification signal paths 16, 18 may be adjusted during operation so that maximum cancellation is achieved in the combiner 19 when smaller input signals are applied to the linearizer (e.g., under small signal conditions). Phase adjustments may be made to one or both of the RF signal paths 16, 18 so that the phases at the output of the paths 16, 18 are appropriate to produce cancellation in combiner 19 (e.g., 180 degree phase difference for a conventional in-phase combiner, etc.). Amplitude adjustments may also be made to one or both of the RF signal paths 16, 18 so that amplitude levels at the output of the paths 16, 18 are appropriate to produce cancellation (e.g., substantially equal amplitude levels for small signal input). In some implementations, phase delay adjustments are made in the non-amplification (i.e., passive) path and amplitude adjustments are made in the amplification path (by, for example, adjusting a bias voltage level on amplifier 22, adjusting variable attenuators 44, 44, and/or other ways). In other embodiments, phase and amplitude adjustments may be made in other ways. As the input power of the linearizer increases, the amount of signal cancellation that occurs in combiner 19 may decrease.
Referring now to
It should be appreciated that linearizer circuit 12′ may be used in the amplifying device 10 described above in conjunction with
Referring now to
It should be appreciated that linearizer circuit 12′ may be used in the amplifying device 10 described above in conjunction with
In some embodiments, the phase delay structure used in the non-amplification signal path 16 may be provided as a microstrip structure. In some other embodiments, co-planar waveguide (CPW) may be used. Other transmission structures may alternatively be used. In some embodiments, CPW may be used for both the non-amplification signal path 16 and the amplification path 18.
In the description above, various examples of the first and second RF signal paths 16, 18 are disclosed, with each describing different component combinations within the paths 16, 18. It should be appreciated that a wide variety of different combinations of delay lines, transmission lines, phase adjusters, variable attenuators, and/or other components may be used within the first and second RF signal paths 16, 18 in other embodiments. Both the number and the location of each different component may change from implementation to implementation.
As shown in
During operation of Doherty amplifier 56, both carrier amplifier 58 and peaking amplifier 60 will be operative during periods of high input signal level (i.e., peak periods) and both will contribute to RF signal amplification. During periods of lower input signal level, on the other hand, peaking amplifier 60 will be pinched off and will not contribute to RF signal amplification. Because peaking amplifier 60 is pinched off during this time, it consumes little to no DC power.
In general, carrier amplifier 58 may be configured as a class B or class AB amplifier and peaking amplifier 60 may be configured as a class C amplifier. Carrier amplifier 58 and peaking amplifier 60 may each be single stage or multi-stage amplifiers. In at least one implementation, both carrier amplifier 58 and peaking amplifier 60 are two stage amplifiers using a one device driving four device architecture. Any type of power transistors may be used within carrier amplifier 58 and peaking amplifier 60 including, for example, bipolar junction transistors (BJTs), field effect transistors (FETs), metal oxide semiconductor FETs (MOSFETs), laterally diffused MOSFETs (LDMOS), metal semiconductor FETs (MESFETs), heterojunction bipolar transistors (HBTs), high voltage HBTs (HV-HBTs), heterostructure FETs (HFETs), high electron mobility transistors (HEMTs), pseudomorphic HEMTs (pHEMTs), metamorphic HEMTs (mHEMTs), and/or others. In addition, transistors using any of a variety of different materials may be used including, for example, silicon, silicon carbide, gallium arsenide, gallium nitride, indium gallium arsenide, aluminum gallium arsenide, and/or others.
Divider 62 may include any type of device that is capable of splitting an RF signal. In at least one embodiment, a divider 82 is used that splits the input signal into two signal components having substantially equal signal amplitudes. However, dividers having unequal output levels may be used in some implementations. In the illustrated embodiment, divider 62 comprises a 90 degree hybrid coupler that generates two equal amplitude (or approximately equal amplitude) output signals that are 90 degrees out of phase. Because the 90 degree hybrid is a four port device, a termination 68 may be provided to terminate the fourth port of the device. Other types of divider circuits may be used in other implementations including, for example, hybrids having other phase shift values, Wilkinson dividers, and/or others.
In the illustrated embodiment, combiner 64 includes a quarter wavelength transmission line section 72 that acts as an impedance inverter at the output of carrier amplifier 58 to combine the output signals of carrier amplifier 58 and peaking amplifier 60. Other types of combiners may be used in other implementations including, for example, hybrid combiners, Wilkinson combiners, and/or others. Although not shown, in some implementations, one or more phase shifters may be provided within Doherty amplifier 56 to ensure that the signals are properly phased for combining. It should be appreciated that Doherty amplifiers may be implemented using any of a number of different architectures and the architecture shown in
As described above, linearizer circuit 54 is operative for processing or shaping the input signal of Doherty amplifier 56 in a manner that is intended to improve overall linearity. This may be accomplished by, for example, increasing the relative magnitude of some portions of the input signal while decreasing the relative magnitude of other portions of the input signal in a manner that complements the operation of Doherty amplifier 56. For example, Doherty amplifier 58 will typically have higher gain for lower power (i.e., small signal) input signals and lower gain for higher power input signals (due to, for example, gain compression and saturation). Thus, linearizer circuit 54 may be configured to increase the relative magnitude of higher power portions of the input signal while decreasing the relative magnitude of lower power portions of the input signal. The term “relative magnitude” is being used here to indicate the magnitude with respect to other portions of the input signal.
In some implementations, linearizer circuit 54 may comprise a linearizer that is substantially the same as or similar to the ones described above (e.g., in
The first and second RF signal paths may be configured in a manner that maintains a 180 degree phase difference between the two paths within a frequency range of interest. In this manner, the output signals of the two paths will subtract within the output combiner. As described previously, a transistor amplifier will typically maintain a relatively constant 180 degree phase shift between an input signal and an output signal across frequency. Therefore, by using an odd number of transistors amplifiers, the desired phase difference between the first and second paths is relatively easy to maintain. It should be appreciated, however, that in some implementations other techniques for maintaining the desired phase difference between the paths may be used.
In some embodiments, additional phase adjustment circuitry (e.g., adjustable phase shifters, etc.) may be provided within the first and/or second RF signal paths to make adjustments in the relative phase of the paths. Similarly, in some embodiments, additional amplitude adjustment circuitry (e.g., variable attenuators, etc.) may be provided within the first and/or second RF signal paths to provide signal amplitude adjustment at desired locations therein (e.g., at the input of an amplifier in the second path, at an output of one or both of the paths, etc.). These adjustable elements may be used to tune or calibrate the first and/or second RF paths to achieve an optimal or near optimal level of linearization for Doherty amplifier 56.
As is well known, a transistor amplifier typically provides higher gain under small signal input conditions. As the magnitude of the input signal increases, the gain of the transistor amplifier will compress and the amplifier will eventually enter saturation. In at least one implementation, linearizer circuit 54 is configured so that the gain of the second RF signal path (the amplified path) is similar to or the same as the gain of the first RF signal path (the non-amplified path) during small signal input conditions (i.e., during low power portions of the input signal). For this reason, a maximum amount of signal cancellation may be achieved in the combiner circuit of the linearizer during small signal conditions. During higher power portions of the input signal, the gain of the second RF signal path will be less than the gain of the first RF signal path, resulting in less signal cancellation in the combiner circuit. In this manner, the relative amplitudes of the lower power portions of the input signal are reduced and the relative amplitudes of the higher power portions of the input signal are increased in the linearizer 54. The linearizer circuit 54 may be configured so that the gain versus input power curve of the linearizer 54 complements the gain versus input power curve of the Doherty amplifier 56 in a manner that linearizes overall amplifier operation. Using this technique, a power amplification system may be achieved that has a relatively linear response characteristic over a relatively wide range of input power levels and a relatively wide RF bandwidth.
In some implementations, linearizer circuit 54 may add a significant amount of loss before the input port of Doherty amplifier 56. As described above, in some embodiments, preamplifier 52 may be provided to boost the input signal before it reaches Doherty amplifier 56 to compensate for the loss of the linearizer 54. When used, preamplifier 52 may be provided before or after linearizer circuit 54. In some implementations, preamplifier 52 may comprise a relatively linear small signal amplifier so that the overall linearity of power amplification system 50 is not degraded. In at least one implementation, a class A amplifier may be used as preamplifier 52, although other types of amplifiers having relatively linear responses may be used in other embodiments. In some other implementations, linearizer circuit 54 may be configured to linearize the combination of preamplifier 52 and Doherty amplifier 56. In at least one embodiment, the gain of preamplifier 52 is set so that the combined gain of preamplifier 52 and linearizer circuit 54 is zero dB or greater across an input power range of interest and a frequency range of interest.
In some implementations, power amplification system 50 of
In at least one implementation, difference unit 72 may subtract the input signal of power amplification system 80 from the output signal to isolate sidebands of the output signal that are indicative of nonlinear operation. The input signal may be coupled from the input of preamplifier 52 (when used) or the input of linearizer 54. The sideband information may then be delivered to digital processor 74 for use in modifying control signals delivered to elements within linearizer circuit 54. As will be appreciated, one or more devices or components may be provided within difference unit 72 (or elsewhere) to ensure that the amplitudes of the coupled input and output signals are of a comparable size before subtraction. This may include, for example, an adjustable attenuator, an automatic gain control (AGC) unit, or some other structure.
In at least one embodiment, processor 74 may include a digital processor and difference unit 72 may be replaced by a pair of analog to digital converters (DACs) that digitize the coupled input and output signals and deliver the resulting digital signals directly to digital processor 74. Processor 74 may then process the digital signals to develop the control signals for the linearizer circuit 54. In at least one approach, processor 74 may normalize the received signals and then perform a difference operation to isolate the sidebands of the output signal or some other indication of nonlinear behavior in the Doherty amplifier.
As described previously, in some embodiments, phase adjustment circuitry and/or amplitude adjustment circuitry may be provided within the first and/or second RF signal paths of linearizer circuit 54 for use in tuning the linearizer. In some implementations, this circuitry may be digitally controllable and processor 74 may adjust these elements in response to a changing non-linear characteristic of Doherty amplifier 56. In addition, or alternatively, processor 74 may adjust bias or power supply levels applied to one or more amplifiers within the second RF signal path of linearizer circuit 54 to adapt to changing nonlinear characteristics in the Doherty amplifier 56.
In some implementations, digital processor 74 may be configured to identify a change in the nonlinear content of the output signal of Doherty amplifier 56 before any changes are made to settings within linearizer circuit 54. In some other implementations, processor 74 may be configured to modify settings within linearizer circuit 54 in response to the non-linear content of the output signal exceeding a threshold level (e.g., a magnitude level of the sidebands, a total energy level of the sidebands, etc.). In some other implementations, adjustments may be made to the settings in linearizer circuit 54 in a periodic or continual manner regardless of a current sideband content of the output signal.
When used at high data rates, signal modulation schemes having a high peak to average power ratio (PAPR) typically make it difficult to provide linear power amplification in an efficient manner. Some signal modulation schemes having a high PAPR include, for example, quadrature phase shift keying (QPSK), 64-quadrature amplitude modulation (64-QAM), 128-QAM, orthogonal frequency division multiplexing (OFDM), and/or others. In various implementations, the power amplification systems described herein are capable of providing a high level of linearity in a very efficient manner when being used with high PAPR modulation schemes. In particular, power amplification systems that use a linearizer circuit feeding a Doherty power amplifier (such as, for example, power amplification systems 50, 80 of
In some implementations, the various circuits and systems described herein are implemented as monolithic microwave integrated circuits (MMICs). However, implementations using discrete circuit elements and implementations that are partially integrated and partially discrete may also be used.
In various embodiments described herein, processors and/or signal processing units may be used in connection with a linearizer circuit to adjust operational parameters of the linearizer circuit based on, for example, the output signal of an RF amplifier. In some implementations, these processors may include digital processing devices such as, for example, a general purpose microprocessor, a digital signal processor (DSP), a reduced instruction set computer (RISC), a complex instruction set computer (CISC), an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a programmable logic array (PLA), a microcontroller, an embedded controller, and/or others, including combinations of the above. In various embodiments, techniques and systems described herein may be implemented using any combination of hardware, software, and firmware.
Having described preferred embodiments which serve to illustrate various concepts, circuits, and techniques which are the subject of this patent, it will now become apparent to those of ordinary skill in the art that other embodiments incorporating these concepts, circuits, and techniques may be used. For example, described herein is a specific exemplary circuit topology and specific circuit implementation for achieving a desired performance. It is recognized, however, that the concepts and techniques described herein may be implemented using other circuit topologies and specific circuit implementations. Accordingly, it is submitted that that scope of the patent should not be limited to the described embodiments but rather should be limited only by the spirit and scope of the following claims.
The present application claims the benefit of U.S. Provisional Patent Application No. 61/612,473 filed on Mar. 19, 2012 and U.S. Provisional Patent Application No. 61/662,512 filed on Jun. 21, 2012, which are both hereby incorporated by reference herein in their entireties.
| Number | Date | Country | |
|---|---|---|---|
| 61662512 | Jun 2012 | US | |
| 61612473 | Mar 2012 | US |
