Patent Application
20170040948
Embodiments of the invention relate to electronic systems and, in particular, to power amplifiers in radio frequency systems.
A wireless device such as a smart phone, tablet, or laptop computer can communicate over multiple frequency bands using one or more common or shared antennas. A desire to transmit at wider bandwidth and/or over different communications networks has increased a demand for the number of bands that a wireless device can communicate over. For example, a wireless device may be specified to operate using one or more of a variety of communications standards including, for example, GSM/EDGE, IMT-2000 (3G), 4G, Long Term Evolution (LTE), Advanced LTE, IEEE 802.11 (Wi-Fi), Mobile WiMAX, Near Field Communication (NFC), Global Positioning System (GPS), GLONASS, Galileo, Bluetooth, and the like. Proprietary standards can also be applicable. The complexities of multi-band communication can be further exacerbated in configurations in which the wireless device is specified to use carrier aggregation.
A power amplifier is described. The power amplifier includes at least a first power amplifier stage coupled with a power supply. The first power amplifier stage is configured to adjust a first output power of the first power amplifier stage based on a power supply voltage of the power supply. The power amplifier also includes at least a first tunable loadline matching network. The first tunable loadline matching network is configured to adjust a loadline value of the power amplifier. The first tunable loadline matching network includes at least a first set of metal oxide semiconductor variable capacitor arrays.
Other features and advantages of embodiments of the present invention will be apparent from the accompanying drawings and from the detailed description that follows.
Embodiments of the present invention are illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like references indicate similar elements and in which:
Disclosed herein are power amplifiers with a tunable loadline matching network.
Power efficiency is one metric by which customers use to determine which power amplifier system to purchase, as power efficiency impacts battery life in mobile applications. There is a demand for manufacturers of power amplifier systems to continuously improve power efficiency, including both the efficiency of power amplifiers themselves and DC-to-DC converters used to generate a power amplifier supply voltage for the power amplifiers. Although improvements in fabrication processing can provide incremental improvements in power efficiency, little progress has been made on architectural and design fronts.
The challenges in improving power efficiency can be exacerbated in power amplifier systems that employ carrier aggregation to widen bandwidth. For example, in a power amplifier system operating using uplink carrier aggregation (mobile to base station), two or more power amplifiers can simultaneously transmit different waveforms associated with different frequency carriers. To provide high efficiency, envelope tracking (ET) can be employed, in which a voltage level of a power amplifier supply voltage of a power amplifier is controlled based on an envelope of an RF signal amplifier by the power amplifier. Using a shared envelope tracking DC-to-DC converter for two or more power amplifiers can be complicated to implement and/or can have a relatively low efficiency. However, using separate envelope tracking DC-to-DC converters for two or more power amplifiers can be prohibitive in cost and/or size. In one example, an envelope tracking DC-to-DC converter is relatively low in efficiency at 80 to 85%, yet a single power amplifier plus envelope converter can deliver system efficiencies (power amplifier+DC-to-DC converter) at about 40%.
Power amplifiers with tunable loadline matching networks are described herein. In certain configurations, a power amplifier includes a tunable loadline matching network implemented using one or more metal oxide semiconductor (MOS) variable capacitor arrays. Examples of MOS variable capacitor arrays can be as described in Ser. No. 14/559,783 and in U.S. Patent Publication No. 2014/0354348, each of which has been expressly incorporated by reference herein.
One or more MOS variable capacitor arrays can be used to implement a tunable matching network at the output of a power amplifier. The capacitance of the tunable matching network can be controlled by programming a capacitance value of the one or more MOS variable capacitor arrays, thereby adjusting the loadline value (real part of impedance in ohms) of the power amplifier. The capacitance of the one or more MOS variable capacitor arrays can be selected based on the output power and battery voltage of the power amplifier system using techniques including those described herein, thereby eliminating a need for a DC-to-DC converter and allowing a power amplifier to be powered directly by a battery voltage. For instance, in one example in which a power amplifier is implemented using a bipolar junction transistor, a collector of the bipolar junction transistor can be connected to the battery voltage rather than to a regulated voltage generated by a DC-to-DC converter.
As skilled artisans will appreciate, the loadline value at the output of a power amplifier impacts both the power amplifier's power-added efficiency (PAE) and the power amplifier's linearity. By controlling the capacitance of the one or more MOS variable capacitor arrays used to implement the tunable loadline matching network, the power amplifier can be operated with a desired loadline value and corresponding power amplifier performance.
In contrast, a power amplifier that operates with a fixed output matching network has a loadline that is fixed in value, such that expected output power can be delivered at a certain minimum collector voltage. For instance, a loadline value in the range of 3 to 5 ohms can be used for 2.8V operations at 30 dBm power. When such a system operates at lower output power levels, power efficiency can suffer since the loadline value is too low.
By implementing a power amplifier to have a tunable loadline matching network, power amplifier efficiency can be increased by increasing the loadline value at back-off power levels. In contrast, the loadline value cannot be increased at low output power levels in a power amplifier system that operates with a fixed output matching network.
A battery voltage of a mobile device can vary over time. Because the battery voltage varies over a wide range, the power amplifier system of
The overall system efficiency of the power amplifier system of
For various embodiments, the power amplifier 10b of
In contrast to the power amplifier system of
In one embodiment, a loadline voltage of a power amplifier can be expressed based on Equation 1 below, where the CollectorVoltage is the available voltage for the power amplifier (which can come be from, for instance, directly from a battery or from a DC-to-DC converter or other regulator), and the SaturationVoltage is the residual voltage across the transistor when fully turned on.
Although Equation 1 is described for a power amplifier implementation using bipolar transistors, the teachings herein are also applicable to power amplifiers implemented using field-effect transistors (FETs), such as complementary metal oxide semiconductor (CMOS) transistors or silicon on insulator (SOI) transistors. In such configurations, the collector voltage in Equation 1 can be replaced with a drain voltage of the FET. In one embodiment, a power amplifier is fabricated using a Gallium Nitride (GaN) process technology.
The plot illustrates loadline value versus output power and voltage as described in Equation 1 above. As shown in
In certain configurations described herein, a power amplifier operates with a programmable loadline matching network and is powered using a battery voltage according to techniques described herein. The efficiency of such a system can be enhanced in part due to the elimination of the DC-to-DC converter and in part due to the programmable aspect of the loadline. The power amplifier efficiency is increased significantly across a wide range of output power.
In the graph of
The graph includes a first plot 31 of normalized PAE versus output power for a power amplifier that receives power directly from a battery voltage and that includes a tunable loadline matching network as described herein. The graph further includes a second plot 32 of normalized PAE versus output power for a power amplifier that receives power from a DC-to-DC converter and that includes a fixed matching network. The graph further includes a third plot 33 of normalized PAE versus output power for a power amplifier that receives power directly from a battery voltage and that includes a fixed matching network.
As shown in
The power amplifier system illustrated in
In the illustrated configuration each of the power amplifiers 10c, 10d receives a power amplifier voltage from a separate DC-to-DC converter 19c, 19d, which can operate using ET or APT. Such an implementation can be require a relatively large area on an integrated circuit (for instance 15 to 25 mm2) and/or expensive.
The power amplifier system illustrated in
The power amplifier system illustrated in
In contrast to the power amplifier system of
The illustrated power amplifier system provides a high efficiency transmitter for uplink carrier aggregation while reducing cost and/or area on an integrated circuit relative to the configuration of
In certain configurations, a power amplifier can include two or power amplifier stages and one or more interstage matching networks to provide impedance matching between the power amplifier stages. In such configurations, the interstage impedance matching can additionally and/or alternatively be tuned using one or more MOS variable capacitor arrays.
Furthermore, power amplifier bias voltage and/or bias current can be adjusted to achieve higher efficiency. Additionally, one or more harmonic termination circuits at the output of the power amplifier can be implemented using MOS variable capacitor arrays, which can be controlled to provide output harmonic tuning.
Such tuning features of power amplifier circuitry are graphically illustrated using an arrow in the PA symbol for the power amplifier 4b of
Depending on the efficiency targets and the solution size, not all tuning features may be used and any combinations can be selected to achieve a desired performance.
The teachings herein can be used to eliminate a need for a DC-to-DC converter for both ET and APT power amplifier systems. Eliminating a DC-to-DC converter can increase system efficiency for a range of transmit output power levels and battery voltages to create an efficient power amplifier for LTE single carrier and uplink carrier aggregation. The teachings herein can enable handsets to support uplink carrier aggregation cost effectively and efficiently. Eliminating the DC-to-DC converter(s) and/or ET or APT can simplify transmitter calibration of handsets, which can reduce cost and/or complexity.
The input impedance transformer 11 can receive an RF input signal on the RF input IN, and can generate an impedance transformed signal 21. The input impedance transformer 11 can provide an impedance transformation from input to output. For example, in one embodiment, the input impedance transformer 11 transforms an input impedance of about 50Ω to an output impedance of about RL, where RL is less than 50Ω, for example, 8Ω.
Transforming the input impedance of the programmable filter 20 in this manner can result in the impedance transformed signal 21 having a smaller voltage level relative to a voltage level of the RF input signal received at the RF input IN. For example, when the programmable filter 20 has an input impedance of about 50Ω, the voltage level of the impedance transformed signal 21 can be smaller than the voltage level of the RF input signal by a factor of about √{square root over (50/RL)}.
The splitter transformer 12 can receive the impedance transformed signal 21 from the input impedance transformer 11, and can generate N split signals, where N is an integer greater than or equal to 2. In the illustrated configuration, the splitter transformer 12 generates a first split signal 22a, a second split signal 22b, and a third split signal 22c. Although an example with N=3 has been illustrated, the principles and advantages disclosed herein are applicable to a broad range of values for the integer N, including 2, 3, 4, 5, or 6 or more.
Splitting the impedance transformed signal 21 into N split signals can further decrease a voltage level of the RF input signal by a factor of N. Including the splitter transformer 12 can also reduce the impedance by a factor of N. For example, when the output impedance of the input impedance transformer 11 has a value of RL, the output impedance of each output of the splitter transformer 12 can have a value of RL/N.
As shown in
The illustrated RF signal processing circuit 13 can be used to process the split signals 22a-22c generated by the splitter transformer 12 to generate the processed signals 23a-23c, respectively. In certain configurations, the RF signal processing circuit 13 can include substantially identical circuitry in the signal paths between the RF signal processing circuit's inputs and outputs.
The combiner transformer 14 receives the processed signals 23a-23c, which the combiner transformer 14 can combine to generate a combined signal 24. The combiner transformer 14 can also provide an impedance transformation. For example, in a configuration in which each output of the RF signal processing circuit 13 has an output impedance of about RL/N, the combiner transformer 14 can have an output impedance of about RL.
The output impedance transformer 15 receives the combined signal 24 from the combiner transformer 14, and generates the RF output signal on the RF output OUT. In certain configurations, the combiner transformer 14 can have an output impedance RL that is less than 50Ω, and the output impedance transformer 15 can be used to provide the RF output signal at an output impedance of about 50Ω.
The illustrated programmable filter 20 provides filtering using the RF signal processing circuit 13, which processes the split signals 22a-22c at lower impedance relative to the programmable filter's input impedance. Thereafter, the processed signals 23a-23c are combined and transformed up in impedance. For example, in one embodiment, the programmable filter's output impedance is about equal to the programmable filter's input impedance.
Configuring the programmable filter 20 to process an RF input signal in this manner can increase the programmable filter's voltage handling capability. For example, when the programmable filter 20 has an input impedance of about 50Ω, the voltage level of the RF input signal can be decreased by a factor of about N√{square root over (50/RL)} before it is provided to the RF signal processing circuit 13, which may include circuitry that is sensitive to high voltage conditions. Accordingly, the illustrated programmable filter 20 can be used to process high voltage RF input signals and/or can have enhanced robustness to variations in voltage standing wave ratio (VWSR).
Furthermore, configuring the programmable filter 20 to process the RF signal at lower impedance can enhance the programmable filter's linearity. In one embodiment, the illustrated configuration can reduce the third-order inter-modulation distortion (IMD3) by a factor of about 40 log10 N√{square root over (50/RL)} relative to a configuration in which an RF input signal is provided directly to an RF signal processing circuit without impedance transformation or splitting. In one illustrative example, N can be selected to be equal to 8 and RL can be selected to be about equal to about 8Ω, and the programmable filter can provide a linearity improvement of about 52 dB. However, other configurations are possible.
As shown in
Although
The RF signal processing circuit 30 can be used to process RF input signals received on the first to third RF inputs I1-I3 to generate RF output signals on the first to third RF outputs O1-O3. As shown in
In one embodiment, the RF signal processing circuit 30 is configured to operate as a filter using techniques including those known in the art, and the control signal CNTL can be used to control a location in frequency of the filter's passband. However, other configurations are possible.
Although
Cascading LC circuits can increase a voltage handling capability of an RF signal processing circuit by limiting a voltage drop across individual circuit components of the LC circuits. For example, in certain implementations, the LC circuits 31a-31i are implemented using MOS capacitors, which can be damaged by large gate-to-drain and/or gate-to-source voltages. By arranging two or more LC circuits in a cascade, a voltage drop across the MOS capacitors during operation can be increased relative to a configuration including a single LC circuit between a particular input and output.
The RF signal processing circuit 30 illustrates one embodiment of the RF signal processing circuit 13 of
The RF signal processing circuit 30 includes a first signal path between the first RF input I1 and the first RF output O1, a second signal path between the second RF input I2 and the second RF output O2, and a third signal path between the third RF input I3 and the third RF output O3. In certain configurations, one or more electrical connections can be provided between corresponding positions along the first to third signals paths. For example, in certain implementations, the RF signal processing circuit 30 is used to process substantially identical RF input signals received on the first to third RF inputs I1-I3, respectively, to generate substantially identical RF output signals on the first to third RF outputs O1-O3. In such configurations, electrical connections can be provided along corresponding positions of signal paths, since the corresponding positions should have substantially the same voltage level. Examples of such electrical connections are illustrated in
The first variable capacitor array 500 receives the control signal CNTL, which can be used to control the first variable capacitor array's capacitance. The capacitance of the first variable capacitor array 500 can be used to control, for example, an input impedance of the RF circuit 1500 and/or to control a ratio of impedance transformation provided by the tunable input matching network 2100. Additionally, the capacitance of the second variable capacitor array 600 can be controlled by the control signal CNTL, thereby controlling, for example, an output impedance of the RF circuit 1500 and/or a ratio of impedance transformation provided by the tunable output matching network 2200.
In one embodiment, the control signal CNTL is received over an interface, such as a serial peripheral interface (SPI) or Mobile Industry Processor Interface radio frequency front end (MIPI RFFE) interface. Although two examples of interfaces have been provided, other interfaces can be used. Although
Including the tunable input matching network 2100 and the tunable output matching network 2200 can enhance performance in a variety of ways, such as improving performance under varying voltage standing wave ratio (VSWR). The first and second variable capacitor arrays 500, 600 can be implemented in accordance with the teachings herein to provide high RF voltage handling capabilities, high Q-factor, low insertion loss, and/or high linearity.
As described above, various embodiments of a filter and various embodiments of a matching network include one or more metal oxide semiconductor (MOS) variable capacitor arrays. For various embodiments, a variable capacitor array includes a plurality of variable capacitor cells electrically connected in parallel. Each of the variable capacitor cells can include a cascade of two or more pairs of anti-series metal oxide semiconductor (MOS) capacitors between an RF input and an RF output. The pairs of anti-series MOS capacitors include a first MOS capacitor and a second MOS capacitor electrically connected in anti-series. A bias voltage generation circuit generates bias voltages for biasing the MOS capacitors of the MOS variable capacitor cells.
A MOS capacitor, according to various embodiments, includes a gate that operates as an anode, and a source and drain that are electrically connected to one another and operate as a cathode. Additionally, a DC bias voltage between the MOS capacitor's anode and cathode can be used to control the MOS capacitor's capacitance. In certain configurations, two or more pairs of anti-series MOS capacitors are cascaded to operate as a variable capacitor cell. As used herein, a pair of MOS capacitors can be electrically connected in anti-series or inverse series when the pair of MOS capacitors is electrically connected in series with the first and second MOS capacitors' anodes electrically connected to one another or with the first and second MOS capacitors' cathodes electrically connected to one another.
The variable capacitor arrays disclosed herein can exhibit high RF signal handling and/or power handling capabilities. For example, including two or more pairs of anti-series MOS capacitors in a cascade can facilitate handling of RF signals with relatively large peak-to-peak voltage swings by distributing the RF signal voltage across multiple MOS capacitors. Thus, the variable capacitor array can handle RF signals of large voltage amplitude and/or high power without overvoltage conditions that may otherwise cause transistor damage, such as gate oxide punch through.
In certain configurations, the bias voltage generation circuit can bias the MOS capacitors of a particular variable capacitor cell at a voltage level selected from a discrete number of two or more bias voltage levels associated with high linearity. Thus, rather than biasing the MOS capacitors at a bias voltage level selected from a continuous tuning voltage range, the bias voltage generation circuit generates the MOS capacitors' bias voltages by selecting a particular cell's bias voltage level from a discrete set of bias voltage levels associated with high linearity. In one embodiment, the bias voltage generation circuit biases a particular MOS capacitor either at a first bias voltage level associated with an accumulation mode of the MOS capacitor or at a second bias voltage level associated an inversion mode of the MOS capacitor.
As used herein and as persons having ordinary skill in the art will appreciate, the terms MOS capacitors refer to any types of capacitors made from transistors with insulated gates. These MOS capacitors can have gates made from metals, such as aluminum, and dielectric regions made out of silicon oxide. However, these MOS capacitors can alternatively have gates made out of materials that are not metals, such as poly silicon, and can have dielectric regions implemented not just with silicon oxide, but with other dielectrics, such as high-k dielectrics. In certain embodiments, the MOS capacitors are implemented using fabricated using silicon on insulator (SOI) processes. For example, an integrated circuit can include a support substrate, a buried oxide (BOX) layer over the support substrate, and a device layer over the BOX layer, and the MOS capacitors can be fabricated in the device layer.
In certain embodiments, a variable capacitor array omits any switches in the signal path between the variable capacitor array's RF input and RF output. Switches can introduce insertion loss, degrade Q-factor, and/or decrease linearity. Thus, rather than providing capacitance tuning by opening and closing switches to set a number of active capacitors from a capacitor bank, capacitance tuning can be provided by biasing MOS capacitors of the variable capacitor cells at different bias voltage levels to provide a desired overall capacitance of the variable capacitor array. In certain configurations, the variable capacitor cells of the variable capacitor array can have the same or different weights or sizes, and the variable capacitor array's overall capacitance is based on a linear combination of the capacitances of the variable capacitor cells.
The variable capacitor arrays herein can have high RF voltage handling capability, while having a relatively small size, a relatively high Q-factor, a relatively high linearity, and/or a relatively low insertion loss. Furthermore, in certain implementations, a variable capacitor array can provide sufficient tuning range to provide filtering across a variety of different frequency bands. Accordingly, the variable capacitor array may be used to provide frequency tuning in a wide range of RF electronics, including, for example, programmable filters, programmable resonators, programmable antenna tuners, programmable impedance matching networks, programmable phase shifters, and/or programmable duplexers.
A wireless device such as a smart phone, tablet, or laptop computer can communicate over multiple frequency bands using one or more common or shared antennas. A desire to transmit at wider bandwidth and/or over different communications networks has increased a demand for the number of bands that a wireless device can communicate over. For example, a wireless device may be specified to operate using one or more of a variety of communications standards including, for example, GSM/EDGE, IMT-2000 (3G), 4G, Long Term Evolution (LTE), Advanced LTE, IEEE 802.11 (Wi-Fi), Mobile WiMAX, Near Field Communication (NFC), Global Positioning System (GPS), GLONASS, Galileo, Bluetooth, and the like. Proprietary standards can also be applicable. The complexities of multi-band communication can be further exacerbated in configurations in which the wireless device is specified to use carrier aggregation.
The metal oxide semiconductor (MOS) capacitors, which can offer enhanced performance over certain other tunable capacitance structures. For instance, certain microelectromechanical systems (MEMS) capacitors can exhibit low Q-factor, poor reliability, and/or limited tuning range. Additionally, other approaches such as coupled resonators can suffer from large size and/or cost, and thus can be unsuitable for certain applications, including smart phones.
The first variable capacitor array 61 includes a first variable capacitor cell 71a, a second variable capacitor cell 71b, and a third variable capacitor cell 71c. The first to third capacitors cells 71a-71c are electrically connected in parallel between the first RF input RFIN1 and the first RF output RFOUT1. The second variable capacitor array 62 includes a first variable capacitor cell 72a, a second variable capacitor cell 72b, and a third variable capacitor cell 72c. The first to third capacitors cells 72a 72c are electrically connected in parallel between the second RF input RFIN2 and the second RF output RFOUT2. The third variable capacitor array 63 includes a first variable capacitor cell 73a, a second variable capacitor cell 73b, and a third variable capacitor cell 73c. The first to third capacitors cells 73a 73c are electrically connected in parallel between the third RF input RFIN3 and the third RF output RFOUT3.
Although
Additionally, although
The bias voltage generation circuit 64 receives the control signal CNTL, and generates a first bias voltage VBIAS1, a second bias voltage VBIAS2, and a third bias voltage VBIAS3. As shown in
The bias voltage generation circuit 64 can be used to control the voltage levels of the first, second, and third bias voltages VBIAS1-VBIAS3 to control the capacitances of the first to third variable capacitor arrays 61-63.
The illustrated variable capacitor cells can be implemented using MOS capacitors. For example, in certain configurations, two or more pairs of anti-series MOS capacitors are cascaded to operate as a variable capacitor cell. Additionally, the first to third bias voltages VBIAS1-VBIAS3 can be used to bias the MOS capacitors at two or more bias voltages associated with a small amount of capacitance variation, and thus with high linearity. For example, in one embodiment, the first to third bias voltages VBIAS1 VBIAS3 can be selectively controlled to bias the MOS capacitors in accumulation or inversion to control the overall capacitance of the arrays.
In certain configurations, the MOS capacitors can be fabricated using silicon on insulator (SOI) processes. However, other configurations are possible, including, for example, implementations in which the MOS capacitors are fabricated using deep sub-micron (DSM) complementary metal oxide semiconductor (CMOS) processes.
In certain configurations herein, a variable capacitor cell can include pairs of MOS capacitors implemented using anti-series configurations. Configuring a variable capacitor cell in this manner can help cancel the second-order intermodulation tones (IM2) and/or control the variation in the cell's capacitance in the presence of RF signals.
As shown in
The IC 60 includes a first signal path from the first RF input RFIN1 to the first RF output RFOUT1 through the first variable capacitor array 61. Additionally, the IC 60 includes a second signal path from the second RF input RFIN2 to the second RF output RFOUT2 through the second variable capacitor array 62, and a third signal path from the third RF input RFIN3 to the third RF output RFOUT3 through the third variable capacitor array 63.
In certain embodiments, the IC 60 does not include any switches in the signal paths between the IC's inputs and outputs through the variable capacitor arrays. By configuring the variable capacitor arrays in this manner, the variable capacitor arrays can have lower insertion loss and/or higher linearity relative to a configuration in which capacitance is provided by selecting discrete capacitors via switches.
As shown in
The first graph 91 includes a high frequency capacitance-voltage (CV) plot 93 for one example of an n-type MOS capacitor. As shown in the CV plot 93, the capacitance of the MOS capacitor can increase with bias voltage level. The increase in capacitance can be associated with the MOS capacitor transitioning between operating regions or modes. For example, at low bias voltage levels, the MOS capacitor can operate in an accumulation mode in which a majority carrier concentration near the gate dielectric/semiconductor interface is greater than a background majority carrier concentration of the semiconductor. Additionally, as the voltage level of the bias voltage increases, the MOS capacitor can transition from the accumulation mode to a depletion mode in which minority and majority carrier concentrations near the gate dielectric/semiconductor interface are less than the background majority carrier concentration. Furthermore, as the voltage level of the bias voltage further increases, the MOS capacitor can transition from the depletion mode to an inversion mode in which the minority carrier concentration near the gate dielectric/semiconductor interface is greater than the background majority carrier concentration.
The first graph 91 has been annotated to include an AC signal component 94 when biasing the MOS capacitor at a bias voltage level VB. When the AC signal component 94 is not present, the MOS capacitor can have a capacitance C. However, as shown by in
With reference to
As shown in
When biased at the first bias voltage level VB1 or the second bias voltage level VB2, the MOS capacitor can nevertheless have a capacitance that varies in the presence of AC signals. However, the first and second bias voltage levels VB1, VB2 can be associated with DC bias points of the MOS capacitor having relatively small capacitance variation or change.
Accordingly, in contrast to the capacitance variation 95 of
In certain embodiments herein, a variable capacitor array includes MOS capacitors that are biased at bias voltages associated with small capacitance variation. By biasing the MOS capacitors in this manner, a variable capacitor array can exhibit high linearity.
Such a variable capacitor array can also have less capacitance variation when operated in a system using multiple frequency bands. For example, when included in a tunable filter, or a tunable matching network, the variable capacitor array can provide relatively constant capacitance even when tuned to frequency bands that are separated by a wide frequency.
In certain embodiments, the first bias voltage level VB1 is selected to operate in the MOS capacitor in an accumulation mode, and the second bias voltage level VB2 is selected to operate the MOS capacitor in an inversion mode. In certain configurations, biasing a MOS capacitor in this manner can achieve a capacitance tuning range of 3:1 or more. However, other tuning ranges can be realized, including, for example, a tuning range associated with a particular manufacturing process used to fabricate the MOS capacitor.
The variable capacitor array 101 includes a first variable capacitor cell 111a, a second variable capacitor cell 111b, and a third variable capacitor cell 111c, which have been electrically connected in parallel between an RF input RFIN and an RF output RFOUT. Although the illustrated variable capacitor array 101 includes three variable capacitor cells, the variable capacitor array 101 can be adapted to include more or fewer variable capacitor cells.
The bias voltage generation circuit 104 receives the control signal CNTL, and generates a first bias voltage 105a for the first variable capacitor cell 111a, a second bias voltage 105b for the second variable capacitor cell 111b, and a third bias voltage 105c for the third variable capacitor cell 111c.
In the illustrated configuration, the control signal CNTL can be used to set the voltage level of the first bias voltage 105a to a first bias voltage level VB1 or to a second bias voltage level VB2. Similarly, the control signal CNTL can be used to set the voltage level of the second bias voltage 105b to the first bias voltage level VB1 or to the second bias voltage level VB2, and to set the voltage level of the third bias voltage 105c to the first bias voltage level VB1 or to the second bias voltage level VB2.
By controlling the voltage levels of the bias voltages to the first or second bias voltage levels VB1, VB2, the variable capacitor array 101 can exhibit a small variation in capacitance in the presence of an RF signal at the RF input RFIN. Accordingly, the variable capacitor array 101 can exhibit high linearity in the presence of RF signals.
The control signal CNTL can control an overall capacitance of the variable capacitor array 101. For example, the size of the first, second, and third MOS capacitors cells 111a 111c can be weighted relative to one another, and an overall capacitance of the variable capacitor array 101 can be based on a sum of the capacitances of the array's variable capacitor cells.
In one embodiment, the variable capacitor array's variable capacitor cells are scaled by a factor of 2, and each of the variable capacitor cells includes k pairs of anti-series MOS capacitors connected in a cascade. For example, a second variable capacitor cell of the variable capacitor array can have a size that is about a factor of 2 relative to a first variable capacitor cell of the variable capacitor array. Additionally, an nth variable capacitor cell in the array can have a size that is about 2n-1 that of the first variable capacitor cell, where n is an integer greater than or equal to 2. Although one possible variable capacitor array sizing scheme has been described, other configurations are possible.
When a variable capacitor array includes n variable capacitor cells that are scaled by a factor of 2 relative to one another and that include k pairs of anti-series MOS capacitors in a cascade, the bias voltage generation circuit 104 can control the array's first variable capacitor cell to a capacitance of C1/2k or C2/2k by biasing the first variable capacitor cell with the first bias voltage level VB1 or the second bias voltage level VB2. Additionally, the bias voltage generation circuit 104 can control the array's second variable capacitor cell to a capacitance of 21*C1/2k or 21*C2/2k by biasing the second variable capacitor cell with the first bias voltage level VB1 or the second bias voltage level VB2. Furthermore, the bias voltage generation circuit 104 can control the array's nth variable capacitor cell to a capacitance of 2n-1*C1/2k or 2n-1*C2/2k by biasing the nth variable capacitor cell with the first bias voltage level VB1 or the second bias voltage level VB2.
Configuring the bias voltage generation circuit 104 to control a bias voltage to one of two voltage levels can simplify a coding scheme associated with the control signal CNTL. For example, in such a configuration, the control signal CNTL can comprise a digital control signal, and individual bits of the digital control signal can be used to control the array's bias voltages to a particular bias voltage level. Although one possible coding scheme of the control signal CNTL has been described, other configurations are possible.
The variable capacitor array 121 includes a first variable capacitor cell 121a, a second variable capacitor cell 121b, and a third variable capacitor cell 121c, which have been electrically connected in parallel between an RF input RFIN and an RF output RFOUT. The first variable capacitor cell 121a includes a cascade of a first pair of anti-series MOS capacitors 141a, a second pair of anti-series MOS capacitors 141b, and a third pair of anti-series MOS capacitors 141c. The second variable capacitor cell 121b includes a cascade of a first pair of anti-series MOS capacitors 142a, a second pair of anti-series MOS capacitors 142b, and a third pair of anti-series MOS capacitors 142c. The third variable capacitor cell 121c includes a cascade of a first pair of anti-series MOS capacitors 143a, a second pair of anti-series MOS capacitors 143b, and a third pair of anti-series MOS capacitors 143c. Although the illustrated variable capacitor array 121 includes three variable capacitor cells, the variable capacitor array 121 can be adapted to include more or fewer variable capacitor cells. Additionally, although the illustrated variable capacitor cells each include a cascade of three pairs of anti-series MOS capacitors, the variable capacitor cells can include more or fewer pairs of anti-series MOS capacitors.
The bias voltage generation circuit 124 receives the control signal CNTL, and generates a first bias voltage VBIAS1 for the first variable capacitor cell 131a, a second bias voltage VBIAS2 for the second variable capacitor cell 131b, and a third bias voltage VBIAS3 for the third variable capacitor cell 131c. In certain configurations, the bias voltage generation circuit 124 can also be used to generate a body bias voltage VBODY, which can be used to control the body voltages of MOS capacitors of the variable capacitor array 121.
Additional details of the integrated circuit 120 can be similar to those described earlier.
Although the variable capacitor cell 150 is illustrated as including three pairs of anti-series MOS capacitors, the teachings herein are applicable to configurations including more or fewer pairs of anti-series MOS capacitors. For example, in one embodiment, a variable capacitor cell includes a cascade of between 2 and 18 pairs of anti-series MOS capacitors.
In the illustrated configuration, each of the pairs of anti-series MOS capacitors 151-153 includes two MOS capacitors electrically connected in anti-series or inverse series. For example, the first pair of anti-series MOS capacitors 151 includes a first MOS capacitor 161 and a second MOS capacitor 162. The first and second MOS capacitors 161, 162 have anodes associated with transistor gates and cathodes associated with transistor source and drain regions. As shown in
As shown in
Arranging two or more pairs of anti-series MOS capacitors in a cascade can increase a voltage handling capability of a variable capacitor cell relative to a configuration including a single pair of anti-series MOS capacitors. For example, arranging two or more pairs of anti-series MOS capacitors in a cascade can increase a voltage handling and/or power handling capability of the variable capacitor cell by distributing RF signal voltage across multiple MOS capacitors.
Accordingly, cascading several pairs of anti-series MOS capacitors can achieve high voltage operation of a variable capacitor cell.
Additionally, the illustrated variable capacitor cell 150 includes pairs of MOS capacitors that are electrically connected in anti-series, which can decrease capacitance variation in the presence of RF signals. For example, when the first and second variable capacitors are each biased with a particular bias voltage, the variable capacitors' capacitance may change when an RF input signal is received on the RF input RFIN. However, a capacitance variation ΔC between MOS capacitors in a given pair can have about equal magnitude, but opposite polarity.
For instance, in the presence of an RF input signal that generates a capacitance variation having a magnitude ΔC, a first MOS capacitor of a pair of anti-series MOS capacitors may have a capacitance CV+ΔC, while the second MOS capacitor may have a capacitance CV−ΔC. Thus, the total capacitance of the anti-series combination of the first and second MOS capacitors 121, 122 can be about equal to ½CV−½ΔC2/CV. Since ½ΔC2 is typically much smaller than ΔC, the anti-series MOS capacitors can exhibit small capacitance variation when RF signals propagate through the variable capacitor cell.
Accordingly, the illustrated variable capacitor cell 150 can provide reduced capacitance variation in the presence of RF signals.
In the illustrated configuration, the first to fourth DC biasing resistors 171-174 have been used to bias the cathodes of the MOS capacitors 161-166 with the first voltage V1, which can be a ground, power low supply, or other reference voltage in certain implementations. Additionally, the first to third control biasing resistors 181-183 are used to bias the anodes of the MOS capacitors 161-166 with the bias voltage VBIAS.
In one embodiment, the DC biasing resistors 171-174 have a resistance selected in the range of 10 kΩ to 10,000 kΩ, and the control biasing resistors 181-183 have a resistance selected in the range of 10 kΩ to 10,000 kΩ. Although one example of resistance values have been provided, other configurations are possible. For example, choosing relatively low resistance values for the biasing resistors can increase control over DC biasing conditions, but can also undesirably increase signal loss and/or degrade linearity since the resistors operate in shunt to an RF signal propagating through the variable capacitor cell. Accordingly, resistance values can vary depending on application, fabrication process, and/or desired performance specifications.
The bias voltages across the MOS capacitors 161-166 can be based on a voltage difference between the bias voltage VBIAS and the first voltage V1. Additionally, a bias voltage generation circuit, such as the bias voltage generation circuit 64 of
In certain configurations, the bias voltage generation circuit can control the bias voltage VBIAS to a voltage level selected from a discrete number of two or more bias voltage levels associated with high linearity. Thus, rather than biasing the MOS capacitors at a bias voltage level selected from a continuous tuning voltage range, the bias voltage generation circuit generates the MOS capacitors' bias voltages by selecting a particular cell's bias voltage level from a discrete set of bias voltage levels associated with high linearity. In one embodiment, the bias voltage generation circuit biases a particular MOS capacitor either at a first bias voltage level associated with an accumulation mode of the MOS capacitor or at a second bias voltage level associated an inversion mode of the MOS capacitor.
Biasing the MOS capacitors 161-166 in this manner can improve linearity relative to a configuration in which the MOS capacitors 161-166 are biased at a bias voltage level selected from a continuous tuning voltage range. For example, a MOS capacitor can exhibit a change in capacitance in response to changes in an applied RF signal, and a magnitude of the capacitance change can vary with the MOS capacitor's bias voltage level.
Accordingly, the illustrated variable capacitor cell 150 can provide high linearity between the RF input RFIN and the RF output RFOUT.
The variable capacitor cell 160 of
In particular, in contrast to the variable capacitor cell 150 of
As shown in
In the illustrated configuration, the first to fourth DC biasing resistors 171174 are used to bias the anodes of the MOS capacitors 201-206 with the first voltage V1, which can be a ground, power low supply, or other reference voltage in certain implementations. Additionally, the first to third control biasing resistors 181-183 are used to bias the cathodes of the MOS capacitors 201-206 with the bias voltage VBIAS.
In certain configurations, the variable capacitor cell 150 of
For example, the RF input RFIN and RF output RFOUT of a variable capacitor cell may be electrically connected to input and output pins of an IC on which the variable capacitor cell is fabricated. Since a MOS capacitor's source and drain regions typically can withstand a greater voltage relative to the MOS capacitor's gate region when fabricated using certain manufacturing processes, the variable capacitor cell 150 of
Additional details of the variable capacitor cell 160 can be similar to those described earlier.
As shown in
Including the diodes 221-226 can enhance the performance in the presence of RF signaling conditions, including, for example, enhanced performance in the presence of voltage changes to an RF signal over a signal cycle. For example, the diodes 221-226 can increase voltage headroom of the MOS capacitors 161-166 relative to a configuration in which the diodes 221-226 are omitted. Additionally, the diodes 221-226 can aid in better distributing an RF signal voltage across the MOS capacitors 161-166, thereby preventing large voltage build-up across a particular MOS capacitor in the cascade. Thus, the illustrated configuration can exhibit greater signal handling and/or power handling capability relative to a configuration that omits the diodes 221-226.
Additional details of the variable capacitor cell 220 can be similar to those described earlier.
As shown in
Additional details of the variable capacitor cell 230 can be similar to those described earlier.
The body biasing resistor 241-246 are used to bias the bodies of the MOS capacitors 161-166 with a body bias voltage VBODY. Including the body biasing resistors 241-246 can aid in increasing the voltage headroom of the MOS capacitors 161-166 in the presence of RF voltage swing. In certain configurations, the body bias voltage VBODY is generated by a bias voltage generation circuit, such as the bias voltage generation circuit 124 of
The body biasing resistors 241-246 can have any suitable resistance value. In one embodiment, the body biasing resistors 241-246 have a resistance selected in the range of 10 kΩ to 10,000 kΩ. Although one example of resistance values have been provided, other configurations are possible, such as resistance values selected for a particular application, fabrication process, and/or desired performance specifications.
Additional details of the variable capacitor cell 240 can be similar to those described earlier.
As shown in
Additional details of the variable capacitor cell 250 can be similar to those described earlier.
As shown in
The signal swing compensation capacitors 261-263 can be used to balance or compensate for differences in voltage, current, and/or phase between pairs of anti-series MOS capacitors. Absent compensation, variation in voltage, current, and/or phase between MOS capacitors may degrade the variable capacitor cell's linearity.
In certain configurations, the capacitance values of the signal swing compensation capacitors 261-263 can be individually selected to improve voltage, current, and/or phase balancing between MOS capacitors 161-166. For example, even when the MOS capacitors 161-166 are implemented with the same size and/or geometry, the capacitance values of the signal switch compensation capacitors 261-263 can be individually selected to provide improve compensation in the presence of RF signaling conditions. In one embodiment, the first signal swing compensation capacitor 261 has a capacitance value that is greater than that of the second signal swing compensation capacitor 262, and the second signal swing compensation capacitor 262 has a capacitance value that is greater than that of the third signal swing compensation capacitor 263. Sizing the signal swing compensation capacitors in this manner may provide enhanced balancing in certain configurations, such as configurations in which large amplitude RF signals are received at the RF input RFIN.
Additional details of the variable capacitor cell 260 can be similar to those described earlier.
As shown in
The signal swing compensation capacitors 261-263 can be included to balance differences in voltage, current, and/or phase between adjacent MOS capacitors, thereby improving linearity of the variable capacitor cell.
Additional details of the variable capacitor cell 270 can be similar to those described earlier.
Additional details of the variable capacitor cell 280 can be similar to those described earlier.
Additional details of the variable capacitor cell 290 can be similar to those described earlier.
Additional details of the variable capacitor cell 300 can be similar to those described earlier.
Additional details of the variable capacitor cell 310 can be similar to those described earlier.
As shown in
The drift protection resistor 321-323 can be used to balance DC operating points across the MOS capacitors 161-166, thereby enhancing performance in the presence of RF amplitude variation or swing. As described earlier, a capacitance provided by a MOS capacitor changes with a voltage difference across the MOS capacitor's anode and cathode. Accordingly, balancing the DC operating point across the MOS capacitors 161-166 can help prevent the capacitances values of the MOS capacitors 161-166 from drifting and becoming unstable in the presence of RF signaling conditions.
In one embodiment, the drift protection resistors 321-323 have a resistance selected in the range of 5 kΩ to 1,000 kΩ. Although one example of resistance values have been provided, other configurations are possible. For example, choosing relatively low resistance values for the drift protection resistors can reduce capacitance value drift due to RF signal swing, but can also impact signaling performance since the resistors are electrically in series between the RF input RFIN and the RF output RFOUT. Accordingly, resistance values can vary depending on application, fabrication process, and/or desired performance specifications.
Additional details of the variable capacitor cell 320 can be similar to those described earlier.
As shown in
The drift protection resistors 321-323 can be included to prevent the capacitances values of the MOS capacitors 201-206 from drifting and becoming unstable in the presence of RF signaling conditions.
Additional details of the variable capacitor cell 330 can be similar to those described earlier.
Additional details of the variable capacitor cell 340 can be similar to those described earlier.
Additional details of the variable capacitor cell 350 can be similar to those described earlier.
Additional details of the variable capacitor cell 360 can be similar to those described earlier.
Additional details of the variable capacitor cell 370 can be similar to those described earlier.
As shown in
The feed forward capacitors 381-383 can be used to balance or compensate for differences in voltage, current, and/or phase between MOS capacitors. For example, the feed forward capacitors 381-383 can be used to balance an RF voltage drop across the MOS capacitors 161-166, thereby improving the linearity of the variable capacitor cell.
In certain configurations, the feed forward capacitors 381-383 can be individually selected to improve voltage, current, and/or phase balancing between MOS capacitors 161-166. For example, even when the MOS capacitors 161-166 are implemented with the same size and/or geometry, the capacitance values of the feed forward capacitors 381-383 can be individually selected to provide improve compensation in the presence of RF signaling conditions. In one embodiment, the first feed forward capacitor 381 has a capacitance value that is greater than that of the second feed forward capacitor 382, and the second feed forward capacitor 382 has a capacitance value that is greater than that of the third feed forward capacitor 383. Sizing the feed forward capacitors in this manner may provide enhanced balancing in certain configurations, such as configurations in which large amplitude RF signals are received at the RF input RFIN.
Additional details of the variable capacitor cell 380 can be similar to those described earlier.
As shown in
The feed forward capacitors 381383 can be included to balance differences in voltage, current, and/or phase between MOS capacitors, thereby improving linearity of the variable capacitor cell.
Additional details of the variable capacitor cell 390 can be similar to those described earlier.
Additional details of the variable capacitor cell 400 can be similar to those described earlier.
Additional details of the variable capacitor cell 410 can be similar to those described earlier.
Additional details of the variable capacitor cell 420 can be similar to those described earlier.
Additional details of the variable capacitor cell 430 can be similar to those described earlier.
As described earlier, the drift protection resistor 321-323 can be used to balance DC operating points across the MOS capacitors 161-166, thereby enhancing performance in the presence of RF amplitude variation or swing. In the illustrated configuration, the first to fourth DC biasing resistors 171-174 have been omitted in favor of controlling the DC bias voltage at the cathodes of the MOS capacitors 161-166 using the drift protection resistors 321-323. For example, in the illustrated configuration, the DC bias voltage at the cathodes of the MOS capacitors 161-166 can be controlled to a DC bias voltage of the RF input RFIN and RF output RFOUT. Additionally, one of the terminals RFIN or RFOUT may be grounded when used in a shunt configuration, thus eliminating the need of first to fourth DC biasing resistors 171-174.
Additional details of the variable capacitor cell 440 can be similar to those described earlier.
As shown in
Additional details of the variable capacitor cell 450 can be similar to those described earlier.
Although
Additionally, although various embodiments of variable capacitor cells are shown in
Unless the context clearly requires otherwise, throughout the description and the claims, the words “comprise,” “comprising,” and the like are to be construed in an inclusive sense, as opposed to an exclusive or exhaustive sense; that is to say, in the sense of “including, but not limited to.” The word “coupled”, as generally used herein, refers to two or more elements that may be either directly connected, or connected by way of one or more intermediate elements. Likewise, the word “connected”, as generally used herein, refers to two or more elements that may be either directly connected, or connected by way of one or more intermediate elements. Additionally, the words “herein,” “above,” “below,” and words of similar import, when used in this application, shall refer to this application as a whole and not to any particular portions of this application. Where the context permits, words in the above Detailed Description using the singular or plural number may also include the plural or singular number respectively. The word “or” in reference to a list of two or more items, that word covers all of the following interpretations of the word: any of the items in the list, all of the items in the list, and any combination of the items in the list.
Moreover, conditional language used herein, such as, among others, “can,” “could,” “might,” “can,” “e.g.,” “for example,” “such as” and the like, unless specifically stated otherwise, or otherwise understood within the context as used, is generally intended to convey that certain embodiments include, while other embodiments do not include, certain features, elements and/or states. Thus, such conditional language is not generally intended to imply that features, elements and/or states are in any way required for one or more embodiments or that one or more embodiments necessarily include logic for deciding, with or without author input or prompting, whether these features, elements and/or states are included or are to be performed in any particular embodiment.
The above detailed description of embodiments of the invention is not intended to be exhaustive or to limit the invention to the precise form disclosed above. While specific embodiments of, and examples for, the invention are described above for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the relevant art will recognize. For example, while processes or blocks are presented in a given order, alternative embodiments may perform routines having steps, or employ systems having blocks, in a different order, and some processes or blocks may be deleted, moved, added, subdivided, combined, and/or modified. Each of these processes or blocks may be implemented in a variety of different ways. Also, while processes or blocks are at times shown as being performed in series, these processes or blocks may instead be performed in parallel, or may be performed at different times.
The teachings of the invention provided herein can be applied to other systems, not only the system described above. The elements and acts of the various embodiments described above can be combined to provide further embodiments.
While certain embodiments of the inventions have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the disclosure. Indeed, the novel methods and systems described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the methods and systems described herein may be made without departing from the spirit of the disclosure. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the disclosure.
This application claims the benefit of U.S. Provisional Application No. 62/201,233 filed on Aug. 5, 2015, which is hereby incorporated by reference in its entirety.
| Number | Date | Country | |
|---|---|---|---|
| 62201233 | Aug 2015 | US |
