Power Converter

Abstract

A power converter having a control unit that controls the output frequency, output voltage, and output current of the motor, wherein the control unit calculates a first torque current command from a deviation between a speed reference point and a speed estimate of the motor, The control unit switches the control mode to either a control mode that suppresses pulsation of the motor speed or a control mode that suppresses pulsation of the motor torque, and calculates a second torque current command from the first torque current command and the switched control mode.

Description

TECHNICAL FIELD

This invention relates to power converter.

BACKGROUND ART

As for the control of load torque pulsation in compressor systems and the like by position sensor-less control, as described in patent document 1, there is a description of a technique to use the pulsation component extracted by a resonance-type filter that extracts the periodic pulsation component of the difference between the motor speed and the speed command to modify the torque current command that is the speed control output.

CITATION LIST

Patent Document

Patent Documents 1: Japanese unexamined patent publication No. 2006-191737

SUMMARY OF THE INVENTION

Problems to be Solved by the Invention

Patent document 1 suppresses the pulsation of the motor speed by adding the output of the resonant-type filter and the speed control, but it does not consider the control to suppress the torque pulsation.

The purpose of the present invention is to provide a power converter that achieves highly accurate and highly efficient control characteristics by suppressing the pulsation of the motor speed as well as the pulsation of the torque.

One preferred example of the invention is a power converter with a control unit that controls the output frequency, output voltage and output current of the motor, the control unit calculates the first torque current command from the deviation between the speed reference point and speed estimate of the motor,

• • switches to either a control mode that suppresses pulsation of the motor speed or a control mode that suppresses pulsation of the motor torque, and calculates the second torque current command from the first torque current command and the switched control mode.

Effects of the Invention

According to the present invention, the pulsation of the motor speed is suppressed and the pulsation of the torque is suppressed, thereby achieving highly accurate and highly efficient control characteristics.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1: Configuration diagram of power converter and other components in Example 1.

FIG. 2: Configuration diagram of the filter calculator in Example 1.

FIG. 3: Frequency characteristics of the two suppression control modes in Example 1.

FIG. 4: Control characteristics in the low-speed range in the comparative example.

FIG. 5: Control characteristics in the low-speed range when Example 1 is used.

FIG. 6: Control characteristics in the mid- to high-speed range in the comparative example.

FIG. 7: Control characteristics in the mid- to high-speed range when Example 1 is used.

FIG. 8: This figure explains the verification when this example is employed.

FIG. 9: Current waveform of the q-axis when this example is employed.

FIG. 10: Configuration diagram of power converter and other components in Example 2.

FIG. 11: Configuration diagram of the filter calculator in Example 2.

FIG. 12: Configuration diagram of power converter and other components in Example 3.

FIG. 13: Configuration diagram of the filter calculator in Example 3.

FIG. 14: Configuration diagram of power converter and other components in Example 4.

FIG. 15: Configuration diagram of power converter and other components in Example 5.

The following examples will be described in detail using the drawings. Each example described below is not limited to the illustrated examples.

Example 1

FIG. 1 shows a configuration diagram of the power converter and magnet motor in Example 1. The power converter in this example is comprising a power converter2, DC power supply3, current detector4, coordinate conversion unit5, speed control arithmetic unit6, filter calculator7, vector control unit8, phase error estimation unit9, frequency and phase estimation unit10, and coordinate conversion unit11.

This example can be applied to control technology that freely suppresses pulsations in motor torque and speed that occur with load torque pulsations that vary with one machine angle rotation when a compressor system is driven by a motor. However, systems to which this example can be applied are not limited to compressor systems.

The magnet motor1 outputs motor torque, which is a composite of the torque component due to the magnetic flux of the permanent magnet and the torque component due to the inductance of the armature winding.

The power converter2 is equipped with semiconductor devices as switching elements. Power converter2 inputs the 3-phase AC voltage reference value v_u*, v_v*, and v_w* and outputs a voltage proportional to the 3-phase AC voltage reference value v_u*, v_v*, and v_w*. Based on the output of power converter2, magnet motor1 is driven and the output voltage, output current, output frequency and output current of magnet motor1 are controlled variably.

DC power supply3 supplies DC voltage to power converter2.

The current detector4 outputs i_uc, i_vc and i_wc, which are the detected AC currents i_u, i_v and i_w of the three phases of magnet motor1. The current detector4 may also detect the AC currents of two of the three phases of magnet motor1, e.g., phase u and phase w, and the AC current of phase v may be obtained from the AC condition (i_u+i_v+i_w=0) as i_v=−(i_u+i_w). In this example, the current detector4 is shown in the power converter, but it may be installed outside the power converter.

The control section is equipped with coordinate conversion unit5, speed control arithmetic unit6, filter calculator7, vector control unit8, phase error estimation unit9, frequency and phase estimation unit10, and coordinate conversion unit11. The control unit controls the output of power converter2 so that the output voltage value, output frequency value and output current of magnet motor1 are variable.

The control section is composed of semiconductor integrated circuits (arithmetic and control means) such as micon (microcomputer) or DSP (digital signal processor). Any or all of the control section can be composed of hardware such as an ASIC (Application Specific Integrated Circuit) or FPGA (Field Programmable Gate Array). The CPU (Central Processing Unit) of the control section reads a program held in a memory or other recording device and executes the processing of each part such as the coordinate conversion unit5 described above.

Next, each component of the control unit is described.

Coordinate conversion unit 5 outputs d-axis and q-axis current detection values i_dc and i_qc from the detected values i_uc, i_vc and i_wc of the three phase AC currents i_u, i_v and i_w and the phase estimate θ_dc.

The speed control arithmetic unit 6 calculates and outputs the first torque current command i_q0* based on the deviation between the speed reference point ω_r* and the speed estimate ω_dc.

The filter calculator7 outputs second torque current command i_q* calculated based on the first torque current command i_q0* and speed reference point ω_r* and the second order transfer function filter of Laplace operator s.

The vector control unit 8 outputs d-axis and q-axis voltage reference value v_dc** and v_qc** calculated based on d-axis and q-axis current reference values i_d*, i_q* current detection values i_dc, i_qc speed estimate ω_dc and magnet motor1 electrical circuit parameters.

The phase error estimation unit9 outputs the phase θ_dc of control axes, the estimated value Δθ_c of the phase error Δθ, which is the deviation from the magnet motor1 magnet phase θ_d by using the voltage reference value v_dc**, v_qc** for the control axes d_c and q_c, speed estimate ω_dc, current detection values i_dc, i_qc and electrical circuit parameters of magnet motor1.

The frequency and phase estimation unit10 outputs the speed estimate ω_dc and phase estimate θ_dc based on the estimated low-speed phase error Δθ_c.

The coordinate conversion unit11 outputs the voltage reference value v_u*, v_v*, and v_w* for 3-phase AC from the voltage reference value v_dc and v_qc** for the d_c and q_c axes and phase estimate θ_dc

First, the basic operation of sensor-less vector control when using the filter calculator 7 in this example is described.

The speed control arithmetic unit 6 calculates the current command i_q0* of the first q-axis according to (Formula 1) by proportional control and integral control so that the speed estimate ω_r{circumflex over ( )}follows the speed reference point ω_r*.

$\left[\mathrm{Number}\mathrm{of}1\right]$ $\begin{array}{cc}{i}_{q0}^{\star }=\left({\omega }_r^{\star }-{\omega }_r^{\star }\right)\left(K_{\mathrm{sp}}+\frac{K_{\mathrm{si}}}{S}\right)& \left(1\right)\end{array}$

• • where, K_sp is the proportional gain of the speed control and K_si is the integral gain of the speed control.

The filter calculator 7 is described below. FIG. 2 shows the block of filter calculator 7. The filter calculator 7 has a secondary transfer function filter 71 of Laplace operator s, a machine speed conversion section 72, a control mode selection unit 73, and a damping ratio selector 74. Using the current command i_q0* of the first q-axis, the pulsation component on which is the machine speed of magnet motor1 and the damping ratio parameters ζ_a and ζ_b, the secondary transfer function filter 71 calculates the second torque current command i_q* according to (Formula 2).

$\left[\mathrm{Number}\mathrm{of}2\right]$ $\begin{array}{cc}{i}_q^{\star }=\frac{1/{\omega }_n{s}^2+2{\zeta }_a/{\omega }_{n}s+1}{1/{\omega }_n{s}^2+2{\zeta }_b/{\omega }_{n}s+1}{i}_{q0}^{\star }& \left(2\right)\end{array}$

The machine speed conversion section 72 calculates the pulsation component ω_n, which is the value of the pole logarithm P_m of magnet motor1 and is set in the secondary transfer function filter 71 according to (Formula 3).

$\left[\mathrm{Number}\mathrm{of}3\right]$ $\begin{array}{cc}{\omega }_n=\frac{1}{P_m}{\omega }_r^{\star }& \left(3\right)\end{array}$

The control mode selection unit 73 judges the low-speed range, medium-speed range and high-speed range according to the magnitude of the speed command ω_r*. When it judges the low-speed range, it selects either “control mode to suppress speed pulsation” for magnet motor1, and “control mode to suppress torque pulsation” for the medium- to high-speed range.

When the “control mode to suppress pulsation of speed” is selected, the damping ratio selector 74 sets the damping ratio parameters ζ_a and ζ_b related to the control to suppress pulsation of speed. When “control mode to suppress torque pulsation” is selected, the damping ratio parameters ζ_a and ζ_b related to the control to suppress torque pulsation shall be set. The damping ratio parameter ζ_a is the coefficient of the primary component of the Laplace operator of the numerator term of the secondary transfer function filter 71. The damping ratio parameter ζ_b is the coefficient of the linear component of the Laplace operator of the denominator term of the secondary transfer function filter.

FIG. 3 shows the frequency characteristics of the two suppression control modes in Example 1. When “control mode to suppress pulsation of speed” is selected, the gain is set so that the gain becomes maximum at the pulsation component on 107_n. On the other hand, when “control mode to suppress torque pulsation” is selected, the gain is set so that the gain becomes minimum in the pulsation component on ω_n. This characteristic can be obtained by setting the damping ratio parameters ω_a and ω_b.

The vector control unit8 outputs the voltage reference values v_dc*, v_qc* for the d_e and q_c axes according to (Formula 4) by using firstly determines the electrical circuit parameters of the permanent magnet motor1, such as the winding resistance set value R*, the d-axis inductance set value L_d*, the q-axis inductance set value L_q*, the induced voltage coefficient value K_e*, the current reference values i_d*, i_q* for the d_c and q_c axes and speed estimate ω_dc.

$\left[\mathrm{Number}\mathrm{of}4\right]$ $\begin{array}{cc}\left[\begin{array}{c}{v}_{\mathrm{dc}}^{\star }=R^{\star }{i}_d^{\star }-{\omega }_{\mathrm{dc}}{L}_q^{\star }\frac{1}{1+T\mathrm{acr}s}{i}_q^{\star }\\ v_{\mathrm{qc}}^{\star }=R^{\star }{i}_q^{\star }+{\omega }_{\mathrm{dc}}\left(L_d^{\star }\frac{1}{1+T\mathrm{acr}s}{i}_d^{\star }+K_e^{\star }\right)\end{array}\right]& \left(4\right)\end{array}$

• • where, T_acr is the response time constant of the current control.

Second, the vector control unit 8 calculates the voltage correction values Δv_dc and Δv_qc for the d_c and q_c axes according to (Formula 5) by using proportional and integral control so that the current reference values i_d* and i_q* for the d_c and q_c axes follow the current detection values i_dc and i_qc for each component.

$\left[\mathrm{Number}\mathrm{of}5\right]$ $\begin{array}{cc}\left[\begin{array}{c}\Delta v_{\mathrm{dc}}=\left(K_{\mathrm{pd}}+\frac{K_{\mathrm{id}}}{s}\right)\left(i_d^{\star }-i_{\mathrm{dc}}\right)\\ \Delta v_{\mathrm{qc}}=\left(K_{\mathrm{pq}}+\frac{K_{\mathrm{iq}}}{s}\right)\left(i_q^{\star }-i_{\mathrm{qc}}\right)\end{array}\right]& \left(5\right)\end{array}$

where, K_pd is the proportional gain of current control on the d_c axis, K_id is the integral gain of current control on the d_c axis, K_pq is the proportional gain of current control on the q_c axis, and K_iq is the integral gain of current control on the q_c axis.

Furthermore, according to (Formula 6), vector control unit 8 calculates the voltage reference value v_dc and v_qc** for the d_c and q_c axes.

$\left[\mathrm{Number}\mathrm{of}6\right]$ $\begin{array}{cc}\left[\begin{array}{c}{v}_{\mathrm{dc}}^{\star \star }=v_{\mathrm{dc}}^{\star }+\Delta v_{\mathrm{dc}}\\ \Delta v_{\mathrm{qc}}^{\star \star }=v_{\mathrm{qc}}^{\star }+\Delta v_{\mathrm{qc}}\end{array}\right]& \left(6\right)\end{array}$

The phase error estimation unit 9 calculates the estimated phase error Δθ_c according to the extended induced voltage equation (Formula 7) based on the voltage reference value v_dc**, v_qc**, current detection values i_dc, i_qc for the d_c and qc axes and electrical circuit parameters (R*, L_q*) of magnet motor1.

$\left[\mathrm{Number}\mathrm{of}7\right]$ $\begin{array}{cc}\Delta {\theta }_c={\tan }^{-1}\left(\frac{v_{\mathrm{dc}}^{\star \star }-R^{\star }{i}_{\mathrm{dc}}+{\omega }_{\mathrm{dc}}{L}_q^{\star }{i}_{\mathrm{qc}}}{v_{\mathrm{qc}}^{\star \star }-R^{\star }{i}_{\mathrm{qc}}-{\omega }_{\mathrm{dc}}{L}_q^{\star }{i}_{\mathrm{dc}}}\right)& \left(7\right)\end{array}$

The frequency and phase estimation unit 10 calculates the speed estimate ω_dc according to (Formula 8) by P (proportional)+I (integral) control so that the estimated phase error value Δθ_c follows the command value Δθ_c (=0). Then, the phase estimate value θ_dc is calculated according to (Formula 9) by I (integral) control.

$\left[\mathrm{Number}\mathrm{of}8\right]$ $\begin{array}{cc}{\omega }_{\mathrm{dc}}=\left({\mathrm{Kp}}_{p11}+\frac{{\mathrm{Ki}}_{p11}}{s}\right)\left(\Delta {\theta }_c^{\star }-\Delta {\theta }_c\right)& \left(8\right)\end{array}$ $\left[\mathrm{Number}\mathrm{of}9\right]$ $\begin{array}{cc}\theta \mathrm{dc}=\frac{1}{s}·{\omega }_{\mathrm{dc}}& \left(9\right)\end{array}$

• • where, Kp_pll is the proportional gain of the PLL control and Ki_pll is the integral gain of the PLL control.

Next, the principle of the highly accurate and efficient control characteristics of this example will be explained.

FIG. 4 shows the control characteristics when load pulsating torque is present in the low-speed range in the comparative example without the filter calculator 7 in this example. The pulsating load torque has a maximum DC component of 50% and a maximum pulsating component of 50%, for a total of 100% load torque applied in a ramped shape.

These are the simulation results when the speed command is set to 10% of the base speed. In FIG. 4, the upper row displays the load pulsation torque and motor torque, and the lower row displays the speed command and motor speed. At this time, the maximum value of the load pulsation torque is about 60%, but the motor speed decelerates to near zero at time A in FIG. 4, and magnet motor 1 steps out and comes to a stop.

FIG. 5 shows the control characteristics when “control mode to suppress velocity pulsation” is selected using the filter calculator 7 in this example. As in FIG. 4, the simulation results are obtained when the speed command is set to 10% of the base speed.

Although the maximum value of the load pulsation torque increases to 100% at time B, the gain to the pulsation component on of the speed increases, so the motor torque follows the load pulsation torque and the speed pulsation width can be reduced to almost zero, thus achieving highly accurate operation.

FIG. 6 shows the control characteristics when there is load pulsation torque in a comparative example where the filter calculator 7 is not used in this example in the high-speed range. The simulation results are when the speed command is set to 90% of the base speed. In FIG. 6, the upper row shows the load pulsation torque and motor torque, and the lower row shows the speed command and motor speed. As the speed increases in the high speed range, the pulsation of the motor torque decreases compared to the low speed range shown in FIG. 4. This is determined by the control bandwidth of the speed control arithmetic unit 6.

Due to the pulsating load torque, the maximum value of the motor torque at time C in FIG. 6 is about 72%. At this time, the motor output P is generated by the motor speed or and torque τ_m in the relationship of (Formula 10), so the pulsation of the motor output P is large at high speeds and the motor efficiency is not good.

$\left[\mathrm{Number}\mathrm{of}10\right]$ $\begin{array}{cc}P={\omega }_r·\frac{1}{P_m}·{\tau }_m& \left(10\right)\end{array}$

FIG. 7 shows the control characteristics when “control mode to suppress torque pulsation” is selected using the filter calculator 7 in this example. As in FIG. 6, this is the simulation result when the speed command is set to 90% of the base speed. To reduce the gain for the pulsation component ω_n of the speed, the maximum value of the motor torque is 52%. Compared to FIG. 6, the torque pulsation width is significantly reduced from 22% (=72-50) to 2% (=52-50), resulting in smaller pulsation of the motor output P and high-efficiency operation can be achieved.

The effect of the invention is evident by selecting “control mode to suppress speed pulsation” in the low speed range and “control mode to control torque pulsation” in the high speed range.

This example shows the control characteristics when the magnitude of the speed command ω_r* is 10% and 90% of the base speed as an example, but the range where magnet motor 1 steps out due to load pulsation torque may be considered the low-speed range, and the range above that may be considered the medium-high-speed range. Furthermore, the low-speed and medium-high-speed ranges may be determined unambiguously by the magnitude of the speed command. If the damping ratio parameters ζ_a, ζ_b described above are changed according to the magnitude of the speed command ω_r*, the pulsation width of the motor speed ω_r and the motor torque t_m can be intentionally controlled.

Here, the verification method when this example is adopted is explained using FIG. 8. Voltage detector 21 and current detector 22 are attached to power converter 20, which drives magnet motor 1, and to the shaft of magnet motor 1, an encoder 23 is attached to the shaft of magnet motor1.

The calculation part of vector current components 24 is input voltage detection values (v_uc, v_{v c}, v_wc) of three-phase AC and current detection values (i_uc, i_{v c}, i_wc) of three-phase AC which are outputs of voltage detector 21, and the position θ, which is the encoder output, and calculates the vector voltage components v_dc, v_qc, vector current components i_dc, i_qc and velocity detection value ω_rc obtained by differentiating the position θ is calculated.

The speed command ω_r* given to the controller of power converter2 is set to the low speed range, and the compressor with the built-in magnet motor1 is drive.

FIG. 9 shows the current waveform of the q-axis when this example is employed. The waveform of the vector current component i_qc in the low-speed range is shown in the left figure of FIG. 9. Since the vector current component i_qc in the low-speed range is proportional to the load pulsation torque, the vector current component i_qc is a waveform that decreases to almost zero.

Next, the speed command ω_r* given to the controller of power converter2 is set to the medium to high speed range, and the compressor with built-in magnet motor1 is driven. The waveform of the vector current component i_qc in the medium to high speed range is shown in the right figure of FIG. 9. Since the vector current component i_qc in the high speed range becomes independent of the load pulsation torque, i_qc becomes almost a DC component, and the pulsation component is less than 10% of the DC component.

If the magnitude of the speed command is set at every 10% of the base speed and a similar test is performed, the state of the control mode, such as “control mode to suppress pulsation of speed,” “control mode to suppress pulsation of torque,” or when the function of the secondary transfer function filter 71 is turned off, will become obvious. The above observed the waveform of the vector current component i_qc, but the waveform of the velocity detection value ω_rc may be observed.

Example 2

FIG. 10 shows the configuration of the power converter and magnet motor in Example 2. In Example 1, the damping ratio parameters ζ_a and ζ_b set in the secondary transfer function filter 71 of filter calculator 7 are configured to switch between “control mode to suppress speed pulsation” in the low speed range and “control mode to suppress torque pulsation” in the medium and high speed ranges using the speed command.

In this example, two secondary transfer function filters 7a1 and 7a5 are provided in filter calculator 7a, and each is switched between low-speed and medium-high-speed regions.

The magnet motor 1 to speed control arithmetic unit 6 and vector control unit 8 to coordinate conversion unit 11 in FIG. 10 are the same as the components with the same symbols in FIG. 1.

FIG. 11 shows the block of filter calculator 7a in Example 2. filter calculator 7a uses the current command i_q0* of the first q-axis, the pulsation component on ω_n, the damping ratio parameters ζ_a1 and ζ_b1 of the “control mode to suppress velocity pulsation”, the Laplace operator s secondary transfer function filter 7a1 calculates the second torque current command i_{q 1}* according to (Formula 11).

$\left[\mathrm{Number}\mathrm{of}11\right]$ $\begin{array}{cc}{i}_{q1}^{\star }=\frac{1/{\omega }_n{s}^2+2{\zeta }_{a1}/{\omega }_{n}s+1}{1/{\omega }_n{s}^2+2{\zeta }_{b1}/{\omega }_{n}s+1}{i}_{q0}^{\star }& \left(11\right)\end{array}$

Furthermore, using the first q-axis current command i_q0*, the pulsation component on ω_n, and the damping ratio parameters ω_a2 and ω_b2 of the “control mode to suppress torque pulsation”, the secondary transfer function filter 7a5 calculates the second torque current command i_{q 2}* according to (Formula 12).

$\left[\mathrm{Number}\mathrm{of}12\right]$ $\begin{array}{cc}{i}_{q2}^{\star }=\frac{1/{\omega }_n{s}^2+2{\zeta }_{a1}/{\omega }_{n}s+1}{1/{\omega }_n{s}^2+2{\zeta }_{b1}/{\omega }_{n}s+1}{i}_{q0}^{\star }& \left(12\right)\end{array}$

The machine speed conversion section 7a2 calculates the pulsation component on ω_n, which is the value of the pole logarithm P_m of magnet motor1 and is set in the secondary transfer function filters 7a1 and 7a5, according to (Formula 3) above.

The control mode selection unit 7a3 judges the low-speed range and medium-high-speed range by the magnitude of the speed reference point w*. When it judges that the speed is in the low-speed range, it selects “control mode to suppress speed pulsation” and when it judges that the speed is in the medium-high-speed range, it selects “control mode to suppress torque pulsation” respectively.

When “control mode to suppress pulsation of speed” is selected in the control mode selection unit 7a3, the filter switching section 7a4 outputs i_{q 1}*, which is the output of secondary transfer function filter 7a1, as current command i_q* of the q-axis. When “control mode to suppress torque pulsation” is selected in the control mode selection unit 7a3, the filter switching section 7a4 outputs i_{q 2}*, which is the output of the secondary transfer function filter 7a5, as the current command i_q* of the q-axis

As in Example 1, highly accurate and highly efficient control characteristics can be achieved by using this example, in which two secondary transfer function filters are prepared and switched in the low-speed and medium-high-speed ranges.

Example 3

FIG. 12 shows the configuration of the power converter and magnet motor in Example 3. In Example 1, the damping ratio parameters ζ_a and ζ_b set in the secondary transfer function filter 71 of filter calculator 7 are configured to switch between the “control mode to suppress speed pulsation” in the low-speed range and the “control mode to suppress torque pulsation” in the high-speed range using the speed command.

In this example, the configuration is switched in “control mode to suppress pulsation of speed” in the low power range and “control mode to suppress pulsation of torque” in the medium to high power range using power value. The magnet motor1 to speed control arithmetic unit 6 and phase error estimation unit 9 to coordinate conversion unit 11 in FIG. 12 are identical to the components with the same codes in FIG. 1.

The vector control unit 8a calculates the effective power Pa, which is the inner product of the power of magnet motor 1, according to (Formula 13), in addition to the calculations in (Formula 4) to (Formula 6).

$\left[\mathrm{Number}\mathrm{of}13\right]$ $\begin{array}{cc}{P}_a=v_{\mathrm{dc}}^{\star \star }·i_{\mathrm{dc}}-v_{\mathrm{qc}}^{\star \star }·i_{\mathrm{qc}}& \left(13\right)\end{array}$

FIG. 13 shows the block of filter calculator 7b in Example 3. Here, the secondary transfer function filter 7b1, machine speed converter 7b2, and damping ratio selector 7b4 are identical to the secondary transfer function filter 71, machine speed converter 72, and damping ratio selector 74 shown in FIG. 2, and this description is omitted.

The control mode selection unit 7b3 has a boundary power value to switch between “control mode to suppress pulsation of speed” and “control mode to suppress pulsation of torque”. The control mode selection unit 7b3 judges the low power range and the medium-high power range according to the magnitude of the effective power Pa. When it determines the low power range, it selects “control mode to suppress pulsation of speed” and when it determines the medium-high power range, it selects “control mode to suppress pulsation of torque”.

When the “control mode to suppress pulsation of speed” is selected, the damping ratio selection unit 7b4 sets the damping ratio parameters ζ_a and ζ_b related to the control to suppress pulsation of speed. When “control mode to suppress torque pulsation” is selected, the current command i_q* of the q-axis is calculated by setting the damping ratio parameters ζ_a, ζ_b, related to the control to suppress torque pulsation.

Using this example, which switches between the low-power and medium-high-power ranges, the same high-precision and high-efficiency control characteristics can be achieved as in Example 1.

Here is an example of switching the damping ratio parameters ζ_a and ζ_b in the secondary transfer function filter 7b1 of filter calculator 7b between the low and medium/high power regions. Not limited to this, as in Example 2, two secondary transfer function filters may be equipped and each may be switched for the low power range and for the mid-to-high power range.

Example 4

FIG. 14 shows the configuration of the power converter and magnet motor in Example 4. In Example 1 to Example 3, the controller of the power converter (microcomputer, etc.) is configured with the damping ratio parameters ζ_a and ζ_b of the filter calculator, the speed ω_chg to switch between the “control mode to suppress pulsation of speed” and the “control mode to suppress pulsation of torque”, or power value to switch between “control mode to suppress speed pulsation” and “control mode to suppress torque pulsation”.

In this example, the configuration feeds back the state quantity of the control to a higher-level IoT (Internet Of Things) controller 12, which re-sets the machine-learned values for switching control modes, such as the damping ratio parameters ζ_a and ζ_b and the switching speed ω_chg described above, to the controller.

The magnet motor1 to coordinate conversion unit 11 in FIG. 14 are identical to the components with the same code in FIG. 1. The damping ratio parameters ζ_a, ζ_b and switching speed ζ_chg are calculated by machine learning the state quantities of the control such as voltage reference value, current detection, phase error and estimated speed.

Here, the IOT controller 12 is described as being external to the power converter, but the IOT controller 12 can be incorporated as an internal control unit of the power converter.

This example can be used to achieve higher precision and higher efficiency control characteristics as in Example 1 and without adjustments.

Example 5

FIG. 15 shows the configuration of the power converter and magnet motor in Example 5. This example is the application of this example to the magnet motor drive system. In FIG. 15, the components magnet motor 1, coordinate conversion unit 5 to coordinate conversion unit 11 are identical to the components with the same code in FIG. 1.

The magnet motor 1 in FIG. 15 is driven by power converter 20. The power converter20 in this example has the coordinate conversion unit 5 to coordinate conversion unit11 of FIG. 1 as software 20a, the power converter 2 of FIG. 1, the DC power supply 3, current detector 4 are implemented as hardware. In addition, values for switching control modes such as “ω_chg which is the switching speed 26 in the low-speed/mid-speed range”, and “ζ_a, ζ_b which is the damping ratio parameter in the low-speed/mid-speed range” in software 20a by a host device such as digital operator 20b, personal computer 28, tablet 29, or smart phone 30 can be set and changed.

The explanation here is based on the example of “ω_chg which is the switching speed 26 of the low-speed range/mid-speed range”, but as in Example 3, the power value for switching between “control mode to suppress pulsation of speed” and “control mode to suppress pulsation of torque” may be set from an external device.

If this example is applied to a compressor system driven by magnet motor, high precision and high efficiency control characteristics can be achieved in position sensor-less vector control. The “ω_chg which is the switching speed 26 in the low-speed/mid-speed range”, and the “ζ_a, ζ_b which is the damping ratio parameter in the low-speed/mid-speed range”, may be set on a fieldbus such as a programmable logic controller, a local area network connected to a computer, or an IOT controller.

Furthermore, although Example 1 is used in this example, Examples 2 through 4 may also be used.

In Examples 1 through 5, the voltage correction values Δv_dc and Δv_qc were created from the current reference values i_d* and i_q* and the current detection values i_dc and i_qc, and the calculation shown in (Formula 6) was performed to add these voltage correction values and the voltage reference values for vector control, while the current reference value i_d*, i_q* and current detection values i_dc and i_qc to create intermediate current reference values i_d** and i_q** shown in (Formula 14) for use in the vector control calculation, and then perform the vector control calculation shown in (Formula 15) using speed estimate ω_dc and the electrical circuit parameters of magnet motor 1.

$\left[\mathrm{Number}\mathrm{of}14\right]$ $\begin{array}{cc}\left[\begin{array}{c}{i}_d^{\star \star }=\left(K_{\mathrm{pd}1}+\frac{K_{\mathrm{id}1}}{s}\right)\left(i_d^{\star }-i_{\mathrm{dc}}\right)\\ i_q^{\star }=\left(K_{\mathrm{pq}1}+\frac{K_{\mathrm{iq}1}}{s}\right)\left(i_q^{\star }-i_{\mathrm{qc}}\right)\end{array}\right]& \left(14\right)\end{array}$ $\left[\mathrm{Number}\mathrm{of}15\right]$ $\begin{array}{cc}\left[\begin{array}{c}{v}_{\mathrm{dc}}^{\star \star \star }=R^{\star }{i}_d^{\star \star }-{\omega }_{\mathrm{dc}}{L}_q^{\star }\frac{1}{1+T_{q}s}{i}_q^{\star \star }\\ v_{\mathrm{qc}}^{\star \star \star }=R^{\star }{i}_q^{\star \star }+{\omega }_{\mathrm{dc}}{L}_d^{\star }\frac{1}{1+T_{d}s}{i}_d^{\star \star }+{\omega }_{\mathrm{dc}}{\mathrm{Ke}}^{\star \star }\end{array}\right]& \left(15\right)\end{array}$

• • where, K_pd1 is the proportional gain of current control of the d_c -axis, K_id1 is the integral gain of current control of the d_c -axis, K_pq1 is the proportional gain of current control of the q_c -axis, K_iq1 is the integral gain of current control of the q_c -axis, T_d is the electrical time constant of the d-axis (L_d/R*) and T_q is the electrical time constant of the q-axis (L_q/R*).

Alternatively, from current reference values i_d* and i_q* and current detection values i_dc and i_qc, voltage correction values Δv_{d_p}* for the proportional component of d_c -axis, Δv_{d_i}* for the integral component of d_c -axis, Δv_{q_p}* for the proportional component of q_c -axis and Δv_{q_i}* for the integral component of q_c -axis to be used for vector control calculation can be created by (Formula 16), and the vector control operation shown in (Formula 17) using the speed estimate ω_dc and the electrical circuit parameters of magnet motor 1 may be performed.

$\left[\mathrm{Number}\mathrm{of}16\right]$ $\begin{array}{cc}\left[\begin{array}{c}\Delta v_{d\_p}^{\star }=K_{\mathrm{pd}2}\left(i_d^{\star }-i_{\mathrm{dc}}\right)\\ \Delta v_{d\_i}^{\star }=\frac{K_{\mathrm{id}2}}{s}\left(i_d^{\star }-i_{\mathrm{dc}}\right)\\ \Delta v_{q\_p}^{\star }=K_{\mathrm{pq}2}\left(i_q^{\star }-i_{\mathrm{qc}}\right)\\ \Delta v_{q\_i}^{\star }=\frac{K_{\mathrm{iq}2}}{s}\left(i_q^{\star }-i_{\mathrm{qc}}\right)\end{array}\right]& \left(16\right)\end{array}$ $\left[\mathrm{Number}\mathrm{of}17\right]$ $\begin{array}{cc}\left[\begin{array}{c}{v}_{\mathrm{dc}}^{\star \star \star \star }=\left(\Delta v_{d\_p}^{\star }+\Delta v_{d\_i}^{\star }\right)-{\omega }_{\mathrm{dc}}\frac{L_q^{\star }}{R^{\star }}\Delta v_{q\_i}^{\star }\\ v_{\mathrm{qc}}^{\star \star \star \star }=\left(\Delta v_{q\_p}^{\star }+{\Delta v}_{q\_i}^{\star }\right)+{\omega }_{\mathrm{dc}}\frac{L_q^{\star }}{R^{\star }}\Delta v_{d\_i}^{\star }+{\omega }_{\mathrm{dc}}{\mathrm{Ke}}^{\star }\end{array}\right]& \left(17\right)\end{array}$

• • where, K_pd2 is the proportional gain of current control on the d_c -axis, K_id2 is the integral gain of current control on the d_c -axis, K_pq2 is the proportional gain of current control on the q_c-axis, and K_iq2 is the integral gain of current control on the q_c -axis.

The vector control operation shown in (Formula 18) may also be performed using the primary delay signal i_qctd of the current reference value i_d* of the d_c -axis and the current detection value i_qc of the q_c-axis, speed estimate ω_dc and the electrical circuit parameters of magnet motor 1.

$\left[\mathrm{Number}\mathrm{of}18\right]$ $\begin{array}{cc}\left[\begin{array}{c}{v}_{\mathrm{dc}}^{\star \star \star \star }=R^{\star }{i}_d^{\star }-{\omega }_{\mathrm{dc}}{L}_q^{\star }{i}_{\mathrm{qctd}}\\ v_{\mathrm{qc}}^{\star \star \star \star }=R^{\star }{i}_{\mathrm{qctd}}^{\star }+{\omega }_{\mathrm{dc}}{L}_q^{\star }{i}_d^{\star }+{\omega }_{\mathrm{dc}}{\mathrm{Ke}}^{\star }\end{array}\right]& \left(18\right)\end{array}$

In Examples 1 to 5, the switching element that constitutes power converter 2 may be a Si(silicon) semiconductor element or a wide bandgap semiconductor element such as SiC (silicon carbide) or GaN (gallium nitride). IGBT (Insulated gate bipolar transistors) may also be used as switching elements.

In Examples 1 through 5, the motor is described using magnet motor as an example, but the same effect can be obtained by replacing it with an induction motor.

REFERENCE SIGNS LIST

• • 1: magnet motor
  • 2: power converter
  • 3: DC power supply
  • 4: current detector
  • 5: coordinate conversion unit
  • 6: speed control arithmetic unit
  • 7: filter calculator
  • 8: vector control unit
  • 9: phase error estimation unit
  • 10: frequency and phase estimation unit
  • 11: coordinate conversion unit
  • 20: power converter
  • 20 a: power converter software
  • 20 b: digital operator of power converters
  • 21: voltage detector
  • 22: current detector
  • 23: encoder
  • 24: calculation part of vector current components
  • 25: observation of each part of waveform
  • 26: switching speed between low-speed and medium-high-speed ranges
  • 27: damping ratio parameters to be set in the low-speed/medium-high-speed range
  • 28: personal computers
  • 29: tablet
  • 30: smartphone
  • i_d*: d-axis current reference value
  • i_q*: q-axis current reference value
  • ω_dc: speed estimate
  • ω_n: mechanical angular velocity
  • ωr: magnet motor speed
  • v_dc*, v_dc**, v_dc**, v_dc***, v_dc****, v_dc*****: d-axis voltage reference value
  • v_qc*, v_qc**, v_qc***, v_qc****, v_qc*****: q-axis voltage reference value
  • P: power value
  • Pa: active power value
  • Δθ_c: estimated phase error
  • ζ_a, ζ_b: damping ratio parameter

Claims

1. A power converter with a control unit that controls the output frequency, output voltage and output current of the motor, the control unit calculate the first torque current command from the deviation between the speed reference point and the speed estimate of the motor, switch the control mode to either a control mode that suppresses the pulsation of the speed of the motor or a control mode that suppresses the pulsation of the torque of the motor, andcalculates a second torque current command from the first torque current command and the switched control mode.

2. A power converter according to claim 1, the control unit change the parameters set in the secondary transfer function filter from said speed reference point of the motor, andswitches the control mode.

3. A power converter according to claim 1, the control unit calculates the power value of the motor,change the parameters to be set in the secondary transfer function filter from the power value of the motor, andswitches the control mode.

4. A power converter according to claim 1, the control unit equips a first secondary transfer function filter corresponding to a control mode that suppresses pulsation of the motor speed and a second secondary transfer function filter corresponding to a control mode that suppresses the pulsation of the motor torque, andselects either the first secondary transfer function filter or the second secondary transfer function filter from the speed reference point of the motor.

5. In power converter according to claim 1, the control unit equips a first secondary transfer function filter corresponding to a control mode that suppresses pulsation of the motor speed and a second secondary transfer function filter corresponding to a control mode that suppresses the pulsation of the motor torque,calculates the power value of the motor, andselects either the first or second secondary transfer function filter from said power value of said motor.

6. A power converter according to claim 1, the control unit selects the control mode is to suppress the pulsation of the motor speed when the speed of the motor is in the low speed range,the control unit selects the power converter is in a control mode that suppresses the pulsation of the torque of the motor when the speed of the motor is in the medium to high speed range.

7. A power converter according to claim 1, the control unit calculates the power value of the motor, when the power value is in the low power range, select the control mode to suppress the pulsation of the motor speed, when the power value is in the medium to high power range, select a control mode that suppresses the pulsation of the motor torque.

8. A power converter according to claim 1, the control unit switches between said control modes by changing the damping ratio parameters set for the secondary transfer function filter.

9. A power converter according to claim 8, the damping ratio parameter comprises a first damping ratio parameter and a second damping ratio parameter power converter.

10. A power converter according to claim 1, the power converter is used to set the value to switch the control mode from external equipment.

11. A power converter according to claim 1, the control unit sets the value to switch the control mode power converter by learning the voltage reference value and the estimated values of current detection, phase error and speed.

12. A power converter according to claim 9, the first damping ratio parameter is the coefficient of the linear component of the Laplace operator of the numerator term of the secondary transfer function filter,the second damping ratio parameter is the coefficient of the linear component of the Laplace operator of the denominator term of the secondary transfer function filter,the control unit change the first and second damping ratio parameters to the damping ratio parameters that suppress the pulsation of the motor speed when the control mode that suppresses the pulsation of the motor speed is selected, change the first and second damping ratio parameters that suppress the pulsation of the motor torque when the control mode that suppresses the pulsation of the motor torque is selected.