Patent Grant
9831857
The present application may be related to U.S. Pat. No. 6,804,502, issued on Oct. 12, 2004 and entitled “Switch Circuit and Method of Switching Radio Frequency Signals”, the disclosure of which is incorporated herein by reference in its entirety. The present application may also be related to U.S. Pat. No. 7,910,993, issued on Mar. 22, 2011 and entitled “Method and Apparatus for use in Improving Linearity of MOSFET's using an Accumulated Charge Sink”, the disclosure of which is incorporated herein by reference in its entirety. The present application may also be related to International Publication No. WO2009/108391 A1, published Sep. 3, 2009, entitled “Method and Apparatus for use in digitally tuning a capacitor in an integrated circuit device”, the disclosure of which is incorporated herein by reference in its entirety. The present application may also be related to US Published Application No. 2013/0222075-A1, published Aug. 29, 2013 entitled “Methods and Apparatuses for Use in Tuning Reactance in a Circuit Device”, the disclosure of which is incorporated herein by reference in its entirety.
1. Field
The present application relates to radio frequency (RF) systems and circuits. In particular, the present application relates to methods and systems for providing an RF power splitter with outputs having a desired phase difference (e.g. relative phase).
2. Description of Related Art
In the field of radio and telecommunications, such as, for example, transmission and/or manipulation of RF signals (e.g. in the range of 100 MHz to 100 GHz), it may be desirable to split an input RF signal of a given power amplitude in two signals with a desired phase relationship between the two signals. Such phase relationship can be obtained by shifting each of the two signals by a predetermined phase with respect to the input RF signal, so that the phase difference between the two signals provides the desired phase relationship. In some cases it can be desirable that the two split signals have a desired power amplitude relationship, based on the given power amplitude of the input RF signal, which can result in split signals with a desired relative phase and power relationships. In some cases it can be desirable that the power amplitude relationship provides two split signals of different power amplitudes. In some cases it can be desirable that the power amplitude relationship provides two split signals of same or similar power amplitudes. In some cases it can be desirable to minimize power loss (e.g. insertion loss) during such splitting of the RF signal in two signals, so that the combined power of the two signals is substantially the same (e.g. within 2 dB) as the power of the RF signal. An exemplary embodiment of such splitting function is provided by a hybrid coupler, which as known to the person skilled in the art uses transmission line properties at RF frequencies to divide (e.g. split) an input RF signal of a given power amplitude in two signals of a same power amplitude (e.g. half the power amplitude of the input RF signal) and a desired fixed phase relationship. In some cases, such signals obtained by splitting an RF signal can be used to drive different amplifiers whose outputs can be combined to provide an output RF signal with a desired characteristic, such as, for example, in a case of a Doherty amplifier. The Doherty amplifier may use two signals in quadrature (e.g. 90° phase difference between the two signals) to feed each of its two constituent amplifiers, the carrier amplifier and the peaking amplifier, where the signal feeding the peaking amplifier is at 90° phase with respect to the signal feeding the carrier amplifier. Given the fixed phase relationship between the signals feeding the two constituent amplifiers of the Doherty amplifier, an input RF signal to the latter amplifier can be split using a hybrid coupler, so that each output of the hybrid coupler is connected to a specific constituent amplifier input, and therefore physically linking the outputs of the coupler and the inputs of the constituent amplifiers and thereby imposing certain design and layout rules for a corresponding circuital implementation. More information about a Doherty power amplifier can be found, for example, in reference [1], which is a paper by W. H. Doherty: “A new High-Efficiency Power Amplifier for Modulated Waves”, presented before the Annual Convention of the Institute of Radio Engineers, May 11-13, 1936, at Cleveland, Ohio, which is incorporated herein by reference in its entirety.
By programmatically controlling a phase relationship between two RF signals of a power divider, such as for example, reversing the phase relationship (e.g. same absolute phase difference but opposite sign), added flexibility in usage of the power divider can be obtained. In some embodiments, such RF signals can be obtained via splitting of an input RF signal of a given power amplitude. In some embodiments such split RF signals can have a same power amplitude based on the power amplitude of the input RF signal. For example, in the exemplary case of the Doherty amplifier fed by a hybrid coupler discussed in the previous section of the present application, programmability of the output signals of the hybrid coupler can relax the amount of discrete tuning at a final manufacturing and test phase of the Doherty amplifier. Such programmability can be provided by a power splitter with programmable output phase shift as presented in the various embodiments of the present disclosure.
According to a first aspect of the present disclosure, an integrated circuit (IC) device configured for operation within a desired operating frequency range is presented, the IC device comprising: an input port configured to receive an input radio frequency (RF) signal; a first output port configured to output a first output RF signal based on the input RF signal; and a second output port configured to output a second output RF signal based on the input RF signal, wherein during operation within the desired operating frequency range and with respect to an input power level of the input power signal: the IC device is configured to operate in one of two modes of operation: a first mode of operation and a second mode of operation, a power of the first output RF signal in the first mode of operation is equal to a power of the first output RF signal in the second mode of operation, a power of the second output RF signal in the first mode of operation is equal to a power of the second output RF signal in the second mode of operation, a sum of the power of the first output RF signal and the power of the second output RF signal is constant, and a relative phase of the first output RF signal to the second output RF signal in the first mode of operation, ΔPhase_m1, is opposite of a relative phase of the first output RF signal to the second output RF signal in the second mode of operation, ΔPhase_m2, such that ΔPhase_m1=−ΔPhase_m2.
According to a second aspect of the present disclosure, an integrated circuit (IC) device configured for operation within a desired operating frequency range is presented, the IC device comprising: an input port configured to receive an input radio frequency (RF) signal; a first output port configured to output a first output RF signal based on the input RF signal; and a second output port configured to output a second output RF signal based on the input RF signal, wherein during operation within the desired operating frequency range and with respect to an input power level of the input RF signal: the IC device is configured to operate in one of two modes of operation: a first mode of operation and a second mode of operation, a power of the first output RF signal in the first mode of operation is equal to a power of the first output RF signal in the second mode of operation, a power of the second output RF signal in the first mode of operation is equal to a power of the second output RF signal in the second mode of operation, a sum of the power of the first output RF signal and the power of the second output RF signal is constant, and an absolute value of a difference between a relative phase of the first output RF signal to the second output RF signal in the first mode of operation, ΔPhase_m1, and a relative phase of the first output RF signal to the second output RF signal in the second mode of operation, ΔPhase_m2, is a desired phase offset, K, such that |ΔPhase_m1|−|ΔPhase_m2|=K.
The accompanying drawings, which are incorporated into and constitute a part of this specification, illustrate one or more embodiments of the present disclosure and, together with the description of example embodiments, serve to explain the principles and implementations of the disclosure.
Splitting of an RF signal into two signals of known power amplitudes and with a desired phase relationship can be performed using a two stage approach, as depicted by the block diagram in
With further reference to the exemplary embodiment of
By denoting Φ1 the phase shift provided by the phase shift module (120) and Φ2 the phase shift provided by the phase shift module (130), the absolute value of the phase difference, ΔPhase, between the two RF signals at output terminals (145, 155) can be provided by the expression |ΔPhase|=|Φ1−Φ2|. In some exemplary embodiments, each phase shift module (120, 130) can have a similar reduced power attenuation (e.g. insertion loss). In some exemplary embodiments, power attenuation of the phase shift modules (120, 130) can be within 1 dB of each other. This means that a power of an RF signal at an output terminal (145, 155) of the phase shift module (120, 130) can be nearly the same as a power of an RF signal at an input terminal (125, 136) of the corresponding phase shift module.
The person skilled in the art readily understands the concept of power loss of an RF signal through a circuital arrangement (e.g. at a corresponding input/output) and various methods for calculating such power loss and various corresponding units for measuring/expressing such power loss (e.g. percentage, decibels, absolute, relative, etc.). The person skilled in the art also knows that the notion of power loss and phases shift may be associated to a frequency of operation of a corresponding/affected signal, such as, for example, the phase shift (Φ1, Φ2) of the phase shift modules (120, 130) and/or the power of the split RF signals provided by the power divider module (110) may be constant for a given frequency range of a corresponding RF signal and may vary for frequencies outside of said range. The various circuital arrangements presented in the present disclosure are therefore designed to operate at a desired frequency range (e.g. contained within the frequencies of 100 MHz to 100 GHz) corresponding to a desired frequency of operation of an RF signal, and within which frequency range the circuital arrangements provide a desired frequency response such as to affect the RF signal's power attenuation and phase shift according to a desired characteristic. Such characteristic can be, for example, for the case of the power divider module (110), a fixed and equal phase shift and power attenuation for each of the split signals, and for the case of the phase shift modules (120, 130), a fixed and opposite phase shift (e.g. Φ1=−Φ2) with a fixed reduced power attenuation. In some embodiments, the fixed reduced power attenuation of the two phase shift modules (120, 130) can be nearly equal (e.g. within 1 dB).
With continued reference to
According to an embodiment of the present disclosure,
A person of ordinary skills readily realizes that in each of the first mode of operation and the second mode of operation the block diagram of
With further reference to such first/second modes of operation and denoting (Φ1A, Φ2A, Φ2B) phase shift values of the phase shift modules (520a, 520b, 530a, 530b), when configured to operate in the first mode, the relative phase difference between the phase of the output RF signal at output port (OP1) and the phase of the output RF signal at output port (OP2) can be provided by the expression ΔPhase_m1=Φ2A−Φ1B, and when configured to operate in the second mode, such relative phase difference can be provided by the expression ΔPhase_m2=Φ2B−Φ1A.
According to an exemplary embodiment of the present disclosure, phase shift values (Φ1A, Φ1B, Φ2A, Φ2B) can be chosen such as Φ1A=−Φ1B, and Φ2A=−Φ2B, and therefore ΔPhase_m1=Φ2A=Φ1B, and ΔPhase_m2=Φ2B−Φ1A=−Φ2A=Φ1B=−ΔPhase_m1. Such exemplary embodiment according to the present disclosure with Φ1A=−Φ1B, and Φ2A=−Φ2B can provide opposite relative phase shifts based on a selected mode of operation (e.g. first/second mode).
According to a further exemplary embodiment of the present disclosure, opposite relative phase shifts based on a selected mode of operation can be provided by choosing Φ1A=Φ2A, and Φ1B=Φ2B, and therefore ΔPhase_m1=Φ2A−Φ1B=Φ1A=Φ1B, and ΔPhase_m2=Φ2B−Φ1A=Φ1B−Φ1A=−ΔPhase_m1.
According to yet further exemplary embodiments of the present disclosure, the phase shift values (Φ1A, Φ1B, Φ2A, Φ2B) can be chosen to satisfy the equation ΔPhase_m2=−ΔPhase_m1. The person skilled in the art realizes that such equation possesses infinite number of values for the phase shifts (Φ1A, Φ1B, Φ2A, Φ2B), including values where |Φ1A|≠|Φ2A|≠|Φ1B|≠|Φ2B|. For example, (Φ1A, Φ1B, Φ2A, Φ2B)=(60, 30, 50, 40) is one possible solution for ΔPhase_m2=−ΔPhase_m1=−20°, and (Φ1A, Φ1B, Φ2A, Φ2B)=(70, 10, 30, 50) is yet another solution for same relative phase output of −20°.
The various embodiments according to the present disclosure allow for the flexibility to choose the phase shift values (Φ1A, Φ1B, Φ2A, Φ2B) in a manner suitable to other related design parameters (e.g. related to filters (520a, 520b, 530a, 530b)) while keeping a constraint related to the phase difference between signals at the output ports (OP1, OP2), such as, for example, ΔPhase_m2=−ΔPhase_m1. According to some embodiments of the present disclosure, and as previously noted, the absolute value of the phase difference between signals at the output ports (OP1, OP2) can be constant in the first and the second modes of operation, such as |ΔPhase_m1|=|ΔPhase_m2|. According to further embodiments of the present disclosure, the absolute value of the phase difference of signals at the output ports (OP1, OP2) in the first mode of operation can be offset by a desired phase offset value with respect to the absolute value of the phase difference of the signals at the output port (OP1, OP2) in the second mode of operation, as expressed by: |ΔPhase_m1|−|ΔPhase_m2|=K, where K is the desired phase offset value. According to some embodiments the value of K can be any value comprised in a range of 0° to 180°.
In each of the two modes of operations, a conduction path provided to an input RF signal at the input terminal of the power divider module (510), can include conduction paths associated to the switches (550, 555, 560, 565). For example, in the first mode of operation, the input RF signal can have two distinct conduction paths for each of the first output RF signal at the (OP1) port and the second output RF signal at the (OP2) port. The first conduction path can include items (510, 550, 520b, 555) and the second conduction path can include items (510, 560, 530a, 565). Switches (550, 555, 560, 565) can be chosen to have a negligible effect on an RF signal through a corresponding conduction path at a frequency/band of operation of the RF signal, such as, for example, a power loss and/or a phase shift of the RF signal due to a switch (550, 555, 560, 565) can be negligible.
By way of further example and not limitation, switches (550, 555, 560, 565) and/or other switches used for implementation of the various embodiments according to the present disclosure of a power splitter with programmable output phase shift can be implemented using transistors, stacked transistors (FETs), diodes, or any other devices or techniques known to or which can be envisioned by a person skilled in the art. In particular, such switching circuitry can be constructed using CMOS technology and various architectures known to the skilled person, such as, for example, architecture presented in U.S. Pat. No. 7,910,993, issued on Mar. 22, 2011 and entitled “Method and Apparatus for use in Improving Linearity of MOSFET's using an Accumulated Charge Sink”, and in U.S. Pat. No. 6,804,502, issued on Oct. 12, 2004 and entitled “Switch Circuit and Method of Switching Radio Frequency Signals”, both incorporated herein by reference in their entirety.
According to an embodiment of the present disclosure, the phase shift modules (520a, 520b, 530a, 530b) can be designed using various types of filters. The person skilled in the art is well aware of design techniques for implementing filters operative at RF frequencies (e.g. 100 MHz to 100 GHz) with given amplitude and phase responses. By way of example and not limitation, such filter circuits can be designed using lumped elements models when a frequency band of operation (e.g. wavelength) is much larger than the circuits' length (e.g. characteristic length).
The phase shift module represented by the second order low pass filter (600A) of
According to an embodiment of the present disclosure, capacitive elements (620, 630) and inductive elements (610, 640) of the filters (600A, 600B) can be chosen so as to provide a desired positive phase shift (e.g. filter 600B) and a desired negative phase shift (e.g. filter 600A) at a frequency band of operation. According to further embodiments of the present disclosure, difference between such positive phase shift and negative phase shift can provide a desired nearly constant phase difference across the entire frequency band of operation. It should be noted that such desired negative/positive phase shifts can be different (e.g. in absolute value) and yet provide a nearly constant phase difference over the frequency band of operation, as depicted, for example, in
Phase=Time_delay*360*Frequency
and characteristics of the time delay elements can be chosen so as the desired relative phase shift (e.g. nearly constant phase difference) is provided by the two time delay elements within the desired frequency band of operation.
According to an exemplary embodiment of the present disclosure, by choosing such elements to provide a phase shift near −45° via the low pass filter (600A) and a phase shift near +45° via the high pass filter (600B) at the frequency band of operation and using filters (600A, 600B) as the phase shift modules (120, 130) of
According to a further exemplary embodiment of the present disclosure, by using a low pass filter (600A) with −45° or near phase shift for the phase shift modules (520b, 530b) and a high pass filter (600B) with +45° or near phase shift for the phase shift modules (520a, 530a), in quadrature signals of opposite polarity can be obtained at ports (OP1, OP2) of the power splitter with programmable output phase shift IC (400) depicted in
An exemplary second order low pass (LP) filter (600AC) and second order high pass (HP) filter (600BC) for providing −/+45° relative phase shift respectively (e.g. near −45° and near)+45° at an operating frequency band of 1.8-2.0 GHz are depicted in
Fc=constant*1/√(LC) and Zo=constant*√(L/C),
where Fc is a cut off frequency of the filter. Using such scaling technique can allow the person skilled in the art to design similar filters for RF signals operating within different frequency bands, such as, for example, a frequency band of [3.6 GHz, 4.0 GHz], [7.2 GHz, 8.0 GHz], etc. As mentioned in the above sections, the present teachings can enable the person skilled in the art to design a power splitter with programmable output phase shift which can operate at any frequency band contained within the frequencies of 100 MHz to 100 GHz, via usage of known filter design techniques, using either lumped element model and/or distributed element models (e.g. transmission lines).
Frequency plots of the amplitude response (e.g. magnitude of ratio output_signal_amplitude/input_signal_amplitude) of the filters (600AC, 600BC) of
As shown in
The inventors of the present disclosure have realized similar filters to filters (600AC, 600BC) for use as phase shift modules of the power splitter with programmable output phase shift according to the various embodiments of the present disclosure for different relative phase shift values (e.g. phase difference) at the corresponding output ports (e.g. OP1, OP2). For example, by using inductance values of (1.61 nH, 10.896 nH) for inductors (610, 640) and capacitance values of (0.644 pF, 4.358 pF) for capacitors (620, 630) of the filters (600AC, 600BC) depicted in
It should be noted that in some exemplary cases of the power splitter with programmable output phase shift device according to the various embodiments of the present disclosure, the absolute value of the phase shift=|Φ| (e.g. with respect to an input RF signal at the input port (IP)) at a given frequency provided by two conductions paths of the split RF signals at ports (OP1, OP2) can be different while the relative phase shift (e.g. its absolute value) of the RF signals at the same output ports can be substantially the same (e.g. less than +/−2°) in the first and the second modes of operation of the device. For example, the filters (600AC, 600BC) of
According to further embodiments of the present disclosure, the difference in the absolute value of the phase shift provided by the two conduction paths of the split RF signals at ports (OP1, OP2) by the selectable phase shift modules (570, 580), as described in the previous paragraph, can be set to provide a desired performance of other related RF parameters of the modules, such as, for example, the insertion loss, the return loss and a die area of the modules. The design flexibility provided by the choice of the absolute value of the phase shifts where the only constraint can be their relative phase (e.g. as obtained at the output ports (OP1, OP2)), can allow, for example, setting a uniform insertion loss and/or a uniform return loss in the corresponding conduction paths of the two modes of operation. Such design flexibility can also allow reducing a corresponding die area used for the integrated circuit (400) by optimizing values (e.g. physical sizes) of corresponding filter (e.g. 600A, 600B) capacitors and inductors.
The RF power splitter with programmable output phase shift according to the teachings of the present disclosure depicted in
The circuital arrangement (700) depicted in
With further reference to
With further reference to
According to an embodiment of the present disclosure,
As depicted in
As depicted in
As mentioned in the prior sections of the present disclosure, the power divider module (510) can split the input RF signal of a given power amplitude provided at the input port (IP) of the IC (800) into two equal powers (based on the power amplitude of the input RF signal) and equal phase signals (e.g. in phase), each split signal being fed to one of the configurable phase shift modules (700, 700) as depicted in
In an exemplary case according to the present disclosure, by having the magnitude |Φ1A|=|Φ1B|=|Φ2A|=|Φ2B| of the configurable phase shift module (700) be equal to 45°, a relative phase of +/−90° at output ports (OP1, OP2) in the first/second mode of operation of the IC (800) can be obtained. According to yet another exemplary case of the present disclosure, by having the magnitude |Φ1| of the configurable phase shift module (700) be equal to 90°, a relative phase of +/−180° at output ports (OP1, OP2) in the first/second mode of operation of the IC (800) can be obtained. It is within the abilities of the person skilled in the art to use teachings according to the present disclosure to design/implement a power splitter with programmable output phase shift where the relative output phase shift (e.g. between output ports (OP1, OP2)) is according to a desired set of values. According to some embodiments of the present disclosure, the absolute value of such relative output phase shift can be within a range of 0° to 180°, or similarly, the absolute value |Φ1A|=|Φ1B|=|Φ2A|=|Φ2B| of the configurable phase shift module (700) be within a range of 0° to 90°.
As used within the present disclosure, the term integrated circuit can refer to a circuit comprising various passive and/or active electrical elements that are monolithically integrated on a same substrate using one of the many semiconductor device fabrication processes and technologies known to the skilled person. Such technologies can include complementary metal-oxide-semiconductor (CMOS) fabrication technology using bulk silicon or silicon on insulator (SOI) substrates. An exemplary case of an SOI substrate is silicon on sapphire (SOS).
As known to the person skilled in the art, due to the nature of a semiconductor device fabrication technology used for IC fabrication, corresponding reactive components can have fabrication tolerances of up to +/−20% of a desired reactive component value. In some cases such tolerances can be the effect of parasitic elements inherent to the fabrication technology which couple to the designed reactive component. Therefore, in a case of a filter fabricated using such fabrication technology, a corresponding filter response (e.g. amplitude/phase response with respect to frequency) can vary according to the corresponding fabrication tolerances. In the exemplary cases of the various phase shift modules according to the various teachings of the present disclosure discussed in the previous sections, such variation in a filter response can translate to offsets in a pass band region position (e.g. bandwidth) of a corresponding filter (e.g. 600A, 600B, 700) and therefore cause offsets in the phase shift magnitude and amplitude response of the filter within a desired RF frequency range. In an exemplary case, such offsets can create phase imbalance of the power splitter with programmable output phase shift, such as a magnitude of the relative phase shift at the output ports (OP1, OP2) is different when the device is operating in the first mode and the second mode. Therefore, it is an object of the present disclosure to provide methods and devices for compensating for such offsets due to the IC device fabrication technology.
According to an embodiment of the present disclosure,
With further reference to IC (900) of
Configuration control (e.g. first/second mode of operation) and adjustments of phase shift and amplitude attenuation of modules (910, 920, 930, 940) of the IC (900) can be performed via the digital interface/control module (950) depicted in the
According to some embodiments of the present disclosure, the tunable reactive elements used in module (700A) depicted in
As mentioned in the previous sections of the present disclosure, adjustability of the phase shift and signal amplitude at output ports (OP1, OP2) of the power splitter with programmable output phase shift IC (900) via modules (910, 920, 930, 940) can be used to compensate for offsets introduced via parasitic elements inherent to the fabrication process of the IC device. The person skilled in the art readily appreciates the flexibility of such adjustability which can be used to compensate for other unintended variations in design parameters. In some exemplary embodiments, variation in a load subjected to output ports (OP1, OP2) at an operating frequency band of interest can be compensated via the same compensation means, regardless of the nature and origin of the variation. For example, load variations can be caused by variations in an input impedance of a device coupled to the ports (OP1, OP2), by imbalance between input impedances of devices coupled to the ports (OP1, OP2), or simply caused by poor layout techniques of the circuit coupling output ports (OP1, OP2) of the IC (900) to corresponding loads.
According to an exemplary embodiment of the present disclosure, the power splitter with programmable output phase shift IC (900) can be configured in the first/second mode of operation to provide split RF signals with a relative phase of +/−Φ1 and a same power amplitude at input ports of two devices coupled to output ports (OP1, OP2) irrespective of parasitic elements introduced via IC (900) manufacturing, input impedance imbalance and layout issues.
According to a further exemplary embodiment of the present disclosure, the two devices coupled to the output ports (OP1, OP2) can be the constituent amplifiers of a Doherty power amplifier and the relative phase of the split RF signals can be from 0° to +/−90°. The person skilled in the art is readily aware of the critical nature of the phase and amplitude balance of the split RF signals at the input of a Doherty power amplifier. In some cases, best performance of the Doherty amplifier as measured by a corresponding characteristics of an RF output signal of the Doherty amplifier (e.g. efficiency of the amplifier) is not obtained by perfectly balanced split RF signals at the input of the constituent Doherty amplifiers, but rather by providing a controlled offset (e.g. in relative phase) between the split RF signals, as depicted in
The person skilled in the art readily appreciates the flexibility provided in the Doherty configuration depicted in
In the configuration depicted in
As mentioned in the previous sections and according to some embodiments of the present disclosure, time delay elements can be used to provide a desired phase difference of RF signals at output ports (OP1, OP2). A corresponding exemplary embodiment is depicted in
The exemplary circuit depicted in
Although the exemplary embodiments of the present disclosure depicted in
The examples set forth above are provided to give those of ordinary skill in the art a complete disclosure and description of how to make and use the embodiments of the present disclosure, and are not intended to limit the scope of what the inventors regard as their disclosure. Modifications of the above described modes for carrying out the disclosure may be used by persons of skill in the art, and are intended to be within the scope of the following claims. All patents and publications mentioned in the specification may be indicative of the levels of skill of those skilled in the art to which the disclosure pertains. All references cited in this disclosure are incorporated by reference to the same extent as if each reference had been incorporated by reference in its entirety individually.
It is to be understood that the disclosure is not limited to particular methods or systems, which can, of course, vary. It is also to be understood that the terminology used herein is for the purpose of describing particular embodiments only, and is not intended to be limiting. As used in this specification and the appended claims, the singular forms “a”, “an”, and “the” include plural referents unless the content clearly dictates otherwise. The term “plurality” includes two or more referents unless the content clearly dictates otherwise. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which the disclosure pertains.
A number of embodiments of the disclosure have been described. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the present disclosure. Accordingly, other embodiments are within the scope of the following claims.
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| Number | Date | Country | |
|---|---|---|---|
| 20160269008 A1 | Sep 2016 | US |
