FIELD OF THE INVENTION
The present invention is related generally to a DC-DC voltage converter and, more particularly, to a PWM controller and method for a DC-DC voltage converter.
BACKGROUND OF THE INVENTION
In conventional constant on-time or hysteretic mode self-clocking DC-DC voltage converters, the generation of the PWM signal is triggered by the ripple of the output voltage. For a DC-DC voltage converter, an output capacitor having a smaller equivalent series resistance will result in a smaller ripple of the output voltage; however, if the ripple is too small, the loop stability will be impaired. Conversely, an output capacitor having a greater equivalent series resistance contributes to increasing loop stability, and yet the resultant output ripple may be so large as to exceed the specification of the voltage converter. Hence, there is always a trade-off between a small output voltage ripple and enough loop stability.
There have been two approaches to achieve a small output voltage ripple while remaining loop stability, both relying on a resistor-capacitor network to generate a signal synchronous or substantially synchronous with the inductor current for stabilizing the loop. As shown in FIG. 1, a conventional DC-DC converter using a ceramic capacitor for the output capacitor COUT has a very small equivalent series resistance RC thereof and therefore, the output voltage VOUT has a small ripple and can be regarded as a DC signal. A resistor voltage divider 1 divides the output voltage VOUT to generate a feedback signal VFB. Due to the small ripple of the output voltage VOUT, the feedback signal VFB also has a small ripple. The output inductor L is connected in parallel with a resistor-capacitor network which includes an external resistor R3 and an external capacitor C3 connected in series. The voltage across the capacitor C3 is equal to the voltage across the equivalent series resistance RL of the output inductor L, which is equal to the product of the inductor current IL and the equivalent series resistance RL. This voltage is converted into a synchronous triangular-wave signal to add to the feedback signal VFB to thereby generate a signal VFB1 having a larger ripple, in place of the original feedback signal VFB, for the PWM controller. The signal VFB1 is equivalent to a feedback signal obtained by dividing an output voltage having a large ripple and thus the loop remains stable despite the fact that the output voltage VOUT has a small ripple. Alternatively, as shown in FIG. 2, an external resistor RS is serially connected to the low-side element Q2 to extract a signal related to the inductor current IL to retrieve a larger ripple for loop stability. While the above two approaches realize a small output voltage ripple and loop stability by using external elements, higher component costs will be needed.
Therefore, it is desired a PWM controller and method for realizing a small output voltage ripple and loop stability without the use of additional external elements.
SUMMARY OF THE INVENTION
An object of the present invention is to provide a PWM controller and method for a DC-DC voltage converter to have a small output voltage ripple as well as loop stability.
Another object of the present invention is to provide a PWM controller and method not relying on additional external elements to realize a small output voltage ripple and loop stability.
According to the present invention, a DC-DC voltage converter includes at least one pair of high-side and low-side elements connected by a phase node to control an inductor current to charges an output capacitor to generate an output voltage, and from a phase-node voltage at the phase node, a signal related to the inductor current is extracted for loop stabilization control.
In one embodiment of the present invention, a PWM controller includes a synchronous-signal generator to generate a first signal according to the phase-node voltage and a PWM comparator circuit to generate a PWM signal according to the first signal and a feedback signal related to the output voltage. The first signal is synchronous and out-of-phase with the inductor current. Preferably, the synchronous-signal generator includes a first low-pass filter to filter out the high-frequency component of the phase-node voltage to generate a second signal synchronous and in phase with the inductor current, and an error amplifier to generate the first signal according to the difference between a first reference voltage and the second signal. Preferably, the first reference voltage is generated by a second low-pass filter by filtering out the AC component of the second signal.
Alternatively, the first signal is synchronous and in phase with the inductor current. Preferably, the synchronous-signal generator includes a first low-pass filter to filter out the high-frequency component of the phase-node voltage to generate a second signal synchronous and in phase with the inductor current, and an error amplifier to generate the first signal according to the difference between a first reference voltage and the second signal. Preferably, the first reference voltage is generated by a second low-pass filter by filtering out the AC component of the second signal.
BRIEF DESCRIPTION OF THE DRAWINGS
These and other objects, features and advantages of the present invention will become apparent to those skilled in the art upon consideration of the following description of the preferred embodiments of the present invention taken in conjunction with the accompanying drawings, in which:
FIG. 1 is a circuit diagram of a conventional DC-DC voltage converter;
FIG. 2 is a circuit diagram of another conventional DC-DC voltage converter;
FIG. 3 is a circuit diagram of a first embodiment according to the present invention;
FIG. 4 is a circuit diagram of a second embodiment according to the present invention;
FIG. 5 is a circuit diagram of a third embodiment according to the present invention;
FIG. 6 is a circuit diagram of a fourth embodiment according to the present invention;
FIG. 7 is a circuit diagram of a fifth embodiment according to the present invention;
FIG. 8 is a circuit diagram of a sixth embodiment according to the present invention;
FIG. 9 is a circuit diagram of a seventh embodiment according to the present invention;
FIG. 10 is a diagram of a simulated output from a constant on-time DC-DC voltage converter using the circuit of FIG. 9.
FIG. 11 is a circuit diagram of an eighth embodiment according to the present invention.
DETAILED DESCRIPTION OF THE INVENTION
According to the present invention, as shown in FIG. 3, a DC-DC voltage converter includes a MOSFET Q1 serving as a high-side element, a MOSFET Q2 serving as a low-side element, and an output inductor L, all connected to a phase node P. When Q1 is on and Q2 is off, the phase-node voltage VP is equivalent to the input voltage VIN; conversely, when Q1 is off, and Q2 is on, the phase-node voltage VP is equivalent to zero. Therefore, the phase-node voltage VP is similar to a square-wave signal. Q1 and Q2 are alternatively switched to control the inductor current IL to charge an output capacitor COUT to generate an output voltage VOUT. The output capacitor COUT may employ an element having a very small equivalent series resistance RC such that the output voltage VOUT has a very small ripple and thus can be regarded as a DC signal. A PWM controller 11 includes a synchronous-signal generator 13 and a PWM comparator circuit 15. According to the phase-node voltage VP, the synchronous-signal generator 13 generates a first signal S12 synchronous and out-of-phase with the inductor current IL. The synchronous-signal generator 13 includes a first low-pass filter 3 and an error amplifier 5. The first low-pass filter 3 includes a resistor R4 and a capacitor C4 configured for filtering out a high-frequency component of the phase-node voltage VP so as to generate a second signal S1 at a node X. The second signal S1 is synchronous and in-phase with the inductor current IL and similar to a triangular wave. The error amplifier 5 multiplies the difference between a first reference voltage VREF1 and the second signal S1 by an appropriate multiplication factor to generate the first signal S12. The output voltage VOUT of this DC-DC voltage converter is divided by a resistor voltage divider 1 to produce a feedback signal VFB, and based on the feedback signal VFB and the first signal S12, the PWM comparator circuit 15 generates a PWM signal to supply to a driver. The PWM comparator circuit 15 includes an adder 7 and a comparator 9. The adder 7 generates a third signal S13 by summing the first signal S12 with a second reference voltage VREF2, and in consequence, the third signal S13 has a ripple synchronous and out-of-phase with the inductor current IL. The feedback signal VFB and the third signal S13 are compared with each other by the comparator 9 to generate the PWM signal. Since the third signal S13 has a ripple synchronous and out-of-phase with the inductor current IL, the comparison between the feedback signal VFB and the third signal S13 is equivalent to a comparison between a large-ripple output voltage and a DC reference voltage. Hence, even though the output voltage VOUT has a small ripple, the loop can still function stably without using additional external elements. Furthermore, since a DC-DC voltage converter has a pin connected to the phase node P in the first place for the purpose of current detection, there is no need to add any pins to the controller chip.
In a second embodiment as shown in FIG. 4, the MOSFETs Q1 and Q2, the output inductor L, the output capacitor COUT, and the resistor voltage divider 1 are configured as in the circuit of FIG. 3, and the PWM controller 21 also includes a synchronous-signal generator 23 and a PWM comparator circuit 25, but has a slightly different configuration from its counterpart shown in FIG. 3. In the synchronous-signal generator 23, the second signal 51 is also generated by the first low-pass filter 3 by filtering out the high-frequency component of the phase-node voltage VP and thus synchronous and in phase with the inductor current IL and similar to a triangular wave, but is provided for the positive input of the error amplifier 5 whose negative input is the first reference voltage VREF1. Likewise, the error amplifier 5 multiplies the difference between the second signal S1 and the first reference voltage VREF1 by an appropriate multiplication factor to generate a first signal S22 which is synchronous and in phase with the inductor current IL. In the PWM comparator circuit 25, the feedback signal VFB is one of the inputs of the adder 7 and to add to the first signal S22 to generate a third signal S23. The third signal S23 is then provided for the negative input of the comparator 9 whose positive input is the second reference voltage VREF2, thereby generating the PWM signal. After the feedback signal VFB having a small ripple is superimposed on the first signal S22 which is synchronous and in phase with the inductor current IL, the resultant third signal S23 is equivalent to a signal generated by dividing an output voltage having a large ripple. Therefore, the comparison of the second reference voltage VREF2 with the third signal S23 is equivalent to the comparison of an output voltage having a large ripple with a DC reference voltage.
The third embodiment shown in FIG. 5 has a same configuration as that of FIG. 3, except that a synchronous-signal generator 33 is slightly different from its counterpart shown in FIG. 3. Likewise, the synchronous-signal generator 33 is in a PWM controller 31 including the PWM comparator circuit 15 and, however, includes a second low-pass filter 30 in addition to the first low-pass filter 3 and the error amplifier 5. The second low-pass filter 30 composed of a resistor R5 and a capacitor C5 filters out most of the AC component of the second signal S1 to generate a first reference voltage VREF1 which is similar to a DC voltage and represents the DC component of the output voltage VOUT. Therefore, the first signal S12 generated by the error amplifier 5 from the difference between the first reference voltage VREF1 and the second signal S1 is proportional to the slope of the inductor current IL and, in an IC package where no pins are provided for feedback of the output voltage VOUT, serves to provide the PWM controller 31 with a signal representing the output voltage VOUT.
The fourth embodiment shown in FIG. 6 has a same configuration as that of FIG. 4, except that a synchronous-signal generator 43 is slightly different from its counterpart shown in FIG. 4. Likewise, the synchronous-signal generator 43 is in a PWM controller 41 including the PWM comparator circuit 25 and, however, includes the second low-pass filter 30 in addition to the first low-pass filter 3 and the error amplifier 5, to filter out most of the AC component of the second signal S1 to generate the first reference voltage VREF1 similar to a DC voltage representing the DC component of the output voltage VOUT. This embodiment has the same advantages as those of the circuit shown in FIG. 5.
The fifth embodiment shown in FIG. 7 has a same configuration as that of FIG. 3, except that a synchronous-signal generator 53 is slightly different from its counterpart shown in FIG. 3. Likewise, the synchronous-signal generator 53 is in a PWM controller 51 including the PWM comparator circuit 15 and, however, includes a second low-pass filter 57 in addition to a first low-pass filter 55 and the error amplifier 5. The first low-pass filter 55 and the second low-pass filter 57 are both connected to the phase node P but have different time constants. The first low-pass filter 55 filters out the high-frequency component of the phase-node voltage VP to generate the first signal S1, and the second low-pass filter 57 filters out most of the AC component of the phase-node voltage VP to generate a first reference voltage VREF1 similar to a DC voltage. The error amplifier 5 multiplies the difference between the first reference voltage VREF1 and the second signal S1 by an appropriate multiplication factor to generate a first signal S12 which is synchronous and out-of-phase with the inductor current IL. This embodiment has the same advantages as those of the circuit shown in FIG. 5.
The sixth embodiment shown in FIG. 8 has a same configuration as that of FIG. 4, except that a synchronous-signal generator 63 is slightly different from its counterpart shown in FIG. 4. Likewise, the synchronous-signal generator 63 is in a PWM controller 61 including the PWM comparator circuit 25 and, however, includes the second low-pass filter 57 in addition to the first low-pass filter 55 and the error amplifier 5. The first low-pass filter 55 and the second low-pass filter 57 are both connected to the phase node P but have different time constants. The first low-pass filter 55 filters out the high-frequency component of the phase-node voltage VP to generate the first signal S1, and the second low-pass filter 57 filters out most of the AC component of the phase-node voltage VP to generate a first reference voltage VREF1 similar to a DC voltage. The error amplifier 5 multiplies the difference between the second signal S1 and the first reference voltage VREF1 by an appropriate multiplication factor to generate a first signal S22 which is synchronous and in phase with the inductor current IL. This embodiment has the same advantages as those of the circuit shown in FIG. 5.
The seventh embodiment shown in FIG. 9 has a same configuration as that of FIG. 5, except that in a PWM controller 71, a synchronous-signal generator 73 uses a transconductance amplifier 77 to implement the error amplifier 5 shown in FIG. 5, which converts the difference VXY between the second signal S1 and the first reference voltage VREF1 into a first signal I1 equal to gm×VXY and synchronous and out-of-phase with the inductor current IL. On the other hand, a PWM comparator circuit 75 uses a resistor R6 to implement the adder 7 shown in FIG. 5, which is connected between the positive input terminal Z of the comparator 9 and the input terminal receiving the second reference voltage VREF2 to generate a first voltage V1=gm×VXY×R6 responsive to the first signal 11. The first voltage V1, which is synchronous and out-of-phase with the inductor current IL, is superimposed on the second reference voltage VREF2 to generate a second voltage VZ1=VREF2+gm×VXY×R6. The comparator 9 compares the second voltage VZ1 with the feedback signal VFB and generates a PWM signal accordingly. In this embodiment, the second voltage VZ1 includes a ripple which is synchronous and out-of-phase with the inductor current IL, and has a desired ripple amplification factor by adjusting the transconductance gm of the transconductance amplifier 77 and the resistance of the resistor R6.
FIG. 10 is a graph of simulated output from a constant on-time DC-DC voltage converter using the circuit of FIG. 9. Despite the equivalent series resistance RC of the output capacitor COUT is very small, the system could remain stable operation and has a very small output voltage ripple during a load transient.
The eighth embodiment shown in FIG. 11 has a same configuration as that of FIG. 6, except that in a PWM controller 81, a synchronous signal generator 83 uses the transconductance amplifier 77 to implement the error amplifier 5 shown in FIG. 6, which converts the difference VXY between the second signal S1 and the first reference voltage VREF1 into a first current signal 12=gm×VXY and synchronous and in phase with the inductor current IL. On the other hand, a PWM comparator circuit 85 uses the resistor R6 to implement the adder 7 shown in FIG. 6, which is connected between the negative input terminal Z of the comparator 9 and the input terminal receiving the feedback signal VFB to generate a first voltage V2=gm×VXY×R6 responsive to the first signal 12. The first voltage V2, which is synchronous and in phase with the inductor current IL, is superimposed on the feedback signal VFB to generate a second voltage VZ2=VFB+gm×VXY×R6. The comparator 9 compares the second voltage VZ2 with the second reference voltage VREF2 and generates a PWM signal accordingly. In this embodiment, the second voltage VZ2 includes a ripple synchronous and in phase with the inductor current IL, and has a desired ripple amplification factor by adjusting the transconductance gm of the transconductance amplifier 77 and the resistance of the resistor R6.
While the present invention has been described in conjunction with preferred embodiments thereof, it is evident that many alternatives, modifications and variations will be apparent to those skilled in the art. Accordingly, it is intended to embrace all such alternatives, modifications and variations that fall within the spirit and scope thereof as set forth in the appended claims.
Patent Application
20100283441
