RADAR APPARATUS

Abstract

A radar transmitter Txs (s=1) generates a baseband transmission signal by modulating a first code sequence having a prescribed code length on the basis of a first transmission timing signal and gives a first transmission phase shift corresponding to each transmission cycle to the transmission signal. A radar receiver Txs (s=2) generates a baseband transmission signal by modulating a second code sequence having the prescribed code length on the basis of a second transmission timing signal and gives, to the transmission signal, a second transmission phase shift that correspond to each transmission cycle and opposite to the first transmission phase.

Description

TECHNICAL FIELD

The present disclosure relates to a radar apparatus which detects a target by receiving pulse signals of reflection waves reflected from the target.

BACKGROUND ART

Radar apparatus using a pulse signal radiate radio waves to the space on the basis of a pulse signal to be transmitted, receive a pulse signal of reflection waves reflected from a target, and measure at least one of a distance between a measuring site and the target and a direction of the target. In recent years, radar apparatus which can detect targets including automobiles and pedestrians by performing a high-resolution measurement using short-wavelength radio waves including microwaves and millimeter waves have been being developed.

For example, Patent document 1 is known as disclosing a radar apparatus which suppresses interference by reducing measurement times of respective sector radars. The radar apparatus disclosed in Patent document 1 will be outlined with reference to FIG. 23. FIG. 23 is an explanatory diagram (timing chart) for description of how the conventional radar apparatus operates.

The radar apparatus of Patent document 1 is equipped with two radar devices, that is, an A radar device and a B radar device. The A radar device is equipped with a sync unit for controlling the timing of an A pulse signal to be transmitted from the A radar device and an I/F unit for receiving a B sync trigger signal that is synchronized with a B pulse signal transmitted from the B radar device. The A radar device receives the B trigger signal from the B radar device through the I/F unit. The A radar device controls the emission timing of the A pulse signal to be emitted from the A radar device on the basis of the received B sync signal.

Therefore, as shown in FIG. 23, the arrival time of an interference wave signal that the B radar device receives from the A radar device always exists in a time interval Tx that is outside an effective reception period of the B radar device. The interference signal that the B radar device receives from the A radar device does not affect a measurement of the B radar device.

The arrival time of an interference wave signal that the A radar device receives from the B radar device exists in an effective reception period of the A radar device. However, the A radar device can eliminate an interference signal effectively by performing restrictive interference suppression processing or gate processing on the interference wave signal coming from the B radar device. In FIG. 23, parameter Tm represents the effective reception period, parameter Tx represents the time interval between effective reception periods, parameter Td represents a time that elapses to arrival of interference waves from the other radar device.

For example, Patent document 2 is known as disclosing a radar apparatus which suppresses occurrence of interference even if reflection signals reflected from a target are received in an asynchronous manner, by using complementary codes (P1, P2) and (Q1, Q2) which are complete complementary codes.

Two radar systems of Patent document 2 transmit and receive different coded pulses (P1, P2, Q1, Q2) as coded pulses of a complete complementary code using carrier waves in the same frequency band.

When receiving plural coded pulses transmitted from the self radar system, one radar system outputs one of autocorrelation function signals R_P1P1(τ), R_P2P2(τ), R_Q1Q1(τ), and R_Q2Q2(τ) corresponding to the plural respective coded pulses (P1, P2, Q1, Q2). When receiving plural coded pulses transmitted from the other radar system, the one radar system outputs one of cross-correlation function signals R_Q1P1(τ), R_Q2P2(τ), R_P1Q1(τ), and R_P2Q2(τ) corresponding to the plural coded pulses (P1, P2) or (Q1, Q2).

Because of the properties of the complete complementary code, the sum of plural outputs autocorrelation function signals (R_P1P1(τ)+R_P2P2(τ) or R_Q1Q1(τ)+R_Q2Q2(τ)) is equal to 0 except for τ being equal to 0 and the sum of plural outputs cross-correlation function signals (R_Q1P1(τ)+R_Q2P2(τ) or R_P1Q1(τ)+R_P2Q2(τ)) is equal to 0 irrespective of τ.

The reception side performs reception processing of calculating plural autocorrelation function signals corresponding to plural respective coded pulses (P1, P2, Q1, Q2) transmitted from the self radar system. As a result, compressed pulses that are free of sidelobes are obtained. Even when plural coded pulses transmitted from the other radar system are received, signal components of the other radar system can be made zero in a process of calculating the sum of autocorrelation function signals. That is, plural radar systems that do not interfere with each other can be provided even if the same frequency band is used between adjoining frequency bands.

PRIOR ART DOCUMENTS

Patent Documents

• Patent document 1: JP-A-7-333328
• Patent document 2: JP-A-61-096482

SUMMARY OF THE INVENTION

Problems to Be Solved by the Invention

However, in Patent document 1, it is necessary that transmission cycles of pulse signals of the A radar device and the B radar device need to be synchronized with each other. Furthermore, for the A radar device to suppress an interference wave signal coming from the B radar device, the A radar device needs to be provided with an additional circuit (e.g., filter circuit) for suppressing interference and the configuration of the receiver of the A radar device is thus complicated. If, alternatively, the A radar device performs gate processing on an interference wave signal coming from the B radar device, an unmeasurable slot corresponding to a reception time of an interference wave coming from the B radar device occurs in an effective reception period Tm of the A radar device.

In Patent document 2, since transmission cycles of coded pulses (P1, P2, Q1, Q2) need to be synchronized with each other, it is necessary to synchronize transmission cycles of coded pulses of the radar systems P and Q.

The present disclosure has been made in the above circumstances, and an object of the disclosure is to provide a radar apparatus which, in the case where plural sector radars are installed being opposed to each other, suppresses interference between the sector radars with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other.

Means for Solving the Problems

This disclosure provides a radar apparatus as mentioned above comprising a first radar transmitter for transmitting a first radar transmission signal generated using a first code sequence having a prescribed code length from a first transmission antenna as a first radio-frequency signal; and a second radar transmitter for transmitting a second radar transmission signal generated using a second code sequence having a prescribed code length from a second transmission antenna as a second radio-frequency signal, wherein the first radar transmission signal is a signal generated by modulating a first baseband signal that has been phase-shifted on the basis of a first transmission timing signal; the second radar transmission signal is a signal generated by modulating a second baseband signal that has been phase-shifted on the basis of a second transmission timing signal; and a phase shift given to the first baseband signal is opposite to a phase shift given to the second baseband signal.

Advantages of the Invention

According to this disclosure, in the case where plural sector radars are installed being opposed to each other, interference between the sector radars can be suppressed with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1( a) illustrates an autocorrelation calculation result of one of a pair of complementary code sequences, FIG. 1(b) illustrates an autocorrelation calculation result of the other of the pair of complementary code sequences, and FIG. 1(c) illustrates addition values of the autocorrelation calculation results of the pair of complementary code sequences.

FIG. 2 illustrates sector radars which constitute a radar apparatus according to a first embodiment and measurement ranges of the respective sector radars.

FIG. 3 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SR_s (s=1, 2) of the radar apparatus according to the first embodiment.

FIG. 4 is a block diagram showing, in detail, the internal configuration of each sector radar SR_s (s=1, 2) of the radar apparatus according to the first embodiment.

FIG. 5 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of each sector radar Sly (s=1, 2) of the radar apparatus according to the first embodiment.

FIG. 6 is an explanatory diagram illustrating a measurement period of the sector radar SR₁ in a case that each transmission interval of a radar transmission signal transmitted from the sector radar SR₂ includes the start of the corresponding transmission cycle of a radar transmission signal transmitted from the sector radar SR₁.

FIG. 7 is an explanatory diagram illustrating a measurement range of the sth sector radar Sly.

FIG. 8( a) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal of a conventional radar apparatus, FIG. 8(b) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal, before reception phase shifting, of each sector radar ST_s of the radar apparatus of this disclosure, and FIG. 8(c) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal, after the reception phase shifting, of each sector radar ST_s of the radar apparatus of this disclosure.

FIG. 9 is a graph showing a relationship between the phase rotation amount due to the frequency error between the sector radars SR₁ and SR₂ and the interference suppression amount of an interference signal coming from the other sector radar in the self sector radar.

FIG. 10( a) is an explanatory diagram illustrating a measurement range of the sector radar SR₁ in a case that transmission codes used in the respective sector radars SR₁ and SR₂ have different code lengths, and FIG. 10(b) is an explanatory diagram illustrating a measurement range of the sector radar SR₂ in a case that the transmission codes used in the respective sector radars SR₁ and SR₂ have different code lengths.

FIG. 11 is a block diagram showing, in detail, the internal configuration of a sector radar SRa_s of a radar apparatus according to a modification of the first embodiment.

FIG. 12 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRb_s (s=1, 2) of a radar apparatus according to the second embodiment.

FIG. 13 is a block diagram showing, in detail, the internal configuration of each sector radar SRb_s (s=1, 2) of the radar apparatus according to the second embodiment.

FIG. 14 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of a radar transmission signal of each sector radar SRb_s (s=1, 2) of the radar apparatus according to the second embodiment.

FIG. 15 is a block diagram showing, in detail, the internal configuration of a sector radar SRc_s of a radar apparatus according to a modification of the second embodiment.

FIG. 16 is an explanatory diagram showing sector radars constituting a radar apparatus according to a third embodiment.

FIG. 17 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRd_s (s=1, 2) of a radar apparatus according to a fourth embodiment.

FIG. 18 is a block diagram showing, in detail, the internal configuration of each sector radar SRd_s (s=1, 2) of the radar apparatus according to the fourth embodiment.

FIG. 19 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission codes of radar transmission signals of a first radar transmitter and a second radar transmitter of each sector radar SRd_s (s=1, 2) of the radar apparatus according to the fourth embodiment.

FIG. 20 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRe_s (s=1, 2) of a radar apparatus according to a fifth embodiment.

FIG. 21 is a block diagram showing, in detail, the internal configuration of each sector radar SRe_s (s=1, 2) of the radar apparatus according to the fifth embodiment.

FIG. 22 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission codes of radar transmission signals of a first radar transmitter and a second radar transmitter of each sector radar SRe_s (s=1, 2) of the radar apparatus according to the fifth embodiment.

FIG. 23 is an explanatory diagram (timing chart) for description of how a conventional radar apparatus operates.

FIG. 24 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of transmission signals in a case that different pairs of complementary code sequences are used in the sector radars SRb_s (s=1, 2) which constitute the radar apparatus according to the second embodiment.

MODES FOR CARRYING OUT THE INVENTION

Background of Embodiments

Radar apparatus receive a signal that is a mixture of reflection waves coming from a nearby target and reflection waves coming from a distant target. Range sidelobes occur due to a signal of reflection waves coming from a nearby target. Where range sidelobes and a main lobe of a signal of reflection waves coming from a distant target exist in mixture, the accuracy of detection of the distant target by a radar apparatus is lowered.

Therefore, radar apparatus which use a pulse signal and are required to perform high-resolution measurements on plural targets are required to transmit a pulse wave or a pulse-modulated wave having an autocorrelation characteristic with low range sidelobe levels (hereinafter referred to as a low range sidelobe characteristic).

When an automobile and a pedestrian are located at the same distance from a measuring site, a radar apparatus receives a signal that is a mixture of signals of reflection waves coming from the automobile and the pedestrian which have different radar cross sections (RCSs). In general, the radar cross section of a pedestrian is smaller than that of an automobile.

Radar apparatus are required to properly receive reflection wave signals coming from an automobile and a pedestrian even if they are located at the same distance from a measuring site. Since the output level (reception level) of a reflection wave signal varies depending on the distance or type of a target, radar apparatus are required to have a reception dynamic range that enables reception of reflection wave signals of various reception levels.

Among radar apparatus as described above are ones which are provided with plural radar units for detecting targets existing in plural different measurement areas, respectively. In the following description, radar units for measurements for different measurement areas to detect targets will be referred to as sector radars. Although the measurement areas of the respective sector radars are different from each other, they may overlap with each other in the case where they are close to each other.

Where the measurement areas of the respective sector radars are close to each other, interference occurs between transmission signals transmitted from the respective sector radars. When interference has occurred, the SNIR (signal to interference and noise power ratio) decreases. In conventional radar apparatus, this means a problem that the target positioning estimation accuracy is lowered.

To solve this problem, the following methods are being studied as measures for suppressing interference between sector radars in conventional radar apparatus.

A first method is a method in which sector respective radars use plural different frequency bands or prescribed narrow frequency bands (subbands) and transmit transmission signals by frequency division multiplication (FDM).

Although the first method can suppress interference between the sector radars by using different frequency bands, it is still associated with the following problem. In the former case in which plural different frequency bands are used, many frequency sources are necessary. In the latter case in which narrow frequency bands are used, the time resolution (which corresponds to the distance resolution) of target positioning estimation of each sector radar lowers.

A second method is a method in which sector radars transmit transmission signals in order in a time-divisional manner. However, in the second method, the measurement time increases because it is necessary to transmit a transmission signal repeatedly to make the SNR of a reflection wave signal coming from a target larger than a prescribed value. Therefore, where there is a limitation on the measurement time, it is difficult to transmit a transmission signal repeatedly so that a prescribed SNR value is attained and hence the target detection accuracy lowers.

A third method is a method in which each sector radar transmit a transmission signal by code division multiplexing (CDM) using plural code sequences that are low in cross-correlation. According to the third method, it is not necessary to add new frequency bands or subbands and the time resolution of target positioning estimation of each sector radar does not lower.

However, where a transmission signal is transmitted from each sector radar by code division multiplexing, reflection wave signals, reflected from a target, of transmission signals transmitted from respective other sector radars are received in an asynchronous manner to cause interference in the self sector radar. The SNR and the target detection accuracy of the self sector radar lower more as the reception level of a reflection wave signal increases.

The following embodiments, which have been conceived in view of the above, are each directed to a radar apparatus which, in the case where plural sector radars are installed being opposed to each other, suppresses interference between the sector radars with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other.

Before describing radar apparatus according to the respective embodiments of the disclosure, the complementary code will be described below briefly as a technique that is a base of the embodiments.

(Complementary Code)

FIG. 1( a) illustrates an autocorrelation calculation result of one of a pair of complementary code sequences. FIG. 1(b) illustrates an autocorrelation calculation result of the other of the pair of complementary code sequences. FIG. 1(c) illustrates addition values of the autocorrelation calculation results of the pair of complementary code sequences.

The complementary code is a code which uses plural complementary code sequences, for example, a pair of complementary code sequences (A_n, B_n). The complementary code has a property that the range sidelobes are made zero when autocorrelation calculation results of the one complementary code sequence A_n and the other complementary code sequence B_n are added together with the same delay time τ(s). Parameter n takes values 1 to L, and parameter L represents a code sequence length or merely a code length.

A method for generating a complementary code is disclosed in the following Referential non-patent document 1, for example:

(Referential non-patent document 1) BUDISIN, S. Z, “NEW COMPLEMENTARY PAIRS OF SEQUENCES” Electron. Lett., 26, (13), pp. 881-883 (1990).

An autocorrelation calculation result of the one complementary code sequence A_n between the complementary code sequences (A_n, B_n) is obtained according to Equation (1). An autocorrelation calculation result of the other complementary code sequence B_n is obtained according to Equation (2). Parameter R represents an autocorrelation calculation result. It is assumed that each of the complementary code sequences A_n and B_n is zero when n>L or n<1 (i.e., A_n=0 and B_n=0 when n>L or n<1). The asterisk “*” is a complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}1\right]& \\ R_{\mathrm{AA}}\left(\tau \right)=\sum _{n=1}^LA_nA_{n+\tau }^{*}& \left(1\right)\\ \left[\mathrm{Formula}2\right]& \\ R_{\mathrm{BB}}\left(\tau \right)=\sum _{n=1}^LB_nB_{n+\tau }^{*}& \left(2\right)\end{array}$

The autocorrelation calculation result R_AA(τ) of the complementary code sequence A_n calculated according to Equation (1) has a peak when the delay time (or shift time) τ is equal to 0 and has range sidelobes for the delay times τ being not equal to 0. Likewise, the autocorrelation calculation result R_BB(τ) of the complementary code sequence B_n calculated according to Equation (2) has a peak when the delay time τ is equal to 0 and has range sidelobes for the delay times τ being not equal to 0.

The addition values of the autocorrelation calculation results R_AA(τ) and R_BB(τ) have a peak when the delay time τ is equal to 0 and have no range sidelobes (i.e., have values 0) for the delay times τ being not equal to 0. In the following description, a peak occurring when the delay time τ is equal to 0 will be referred to as a main lobe. The above relationships are expressed as Formulae (3):

[Formulae 3]

R _AA(τ)+R_BB(τ)≠0, when τ=0

R _AA(τ)+R_BB(τ)=0, when τ≠0  (3)

Because of the above-described autocorrelation characteristics, the complementary code can reduce the peak sidelobe levels with a shorter code length. Therefore, the use of a complementary code having a short code length can reduce the reception dynamic range in a radar apparatus even in the case where it receives a signal obtained through mixing of reflection waves coming from a nearby target and reflection waves coming from a distant target.

Embodiment 1

First, a radar apparatus according to a first embodiment of the disclosure will be described with reference to the drawings. FIG. 2 illustrates sector radars SR₁ and SR₂ which constitute the radar apparatus 10 according to the first embodiment and measurement ranges of the respective sector radars SR₁ and SR₂. The radar apparatus 10 is configured so as to include plural sector radars, for example, the two sector radars SR₁ and SR₂.

Reception signals received by the sector radar SR₁ include a reflection wave signal that is produced in such a manner that a radar transmission signal transmitted from the sector radar SR₁ is reflected by a target TAR₁ and a radar transmission signal that is an interference wave signal transmitted from the sector radar SR₂. Likewise, reception signals received by the sector radar SR₂ include a reflection wave signal that is produced in such a manner that a radar transmission signal transmitted from the sector radar SR₂ is reflected by a target TAR₂ and a radar transmission signal that is an interference wave signal transmitted from the sector radar SR₂.

The sector radars SR₁ and SR₂ shown in FIG. 2 are installed being opposed to each other so that measurement areas A and B of the respective sector radars SR₁ and SR₂ are arranged so as to form approximately a single straight line and to overlap with each other. Inequalities (4) hold between the distance Rd between the sector radars SR₁ and SR₂ and the maximum measurement distances R₁ and R₂ of the respective sector radars SR₁ and SR₂.

$\begin{array}{cc}\left[\mathrm{Formula}4\right]& \\ \frac{R_d}{2}<R_1,\frac{R_d}{2}<R_2& \left(4\right)\end{array}$

In the following description, it is assumed that the sector radars SR₁ and SR₂ have the same transmission cycle T_r and the same transmission interval T_w but transmit radar transmission signals in an asynchronous manner. In FIG. 2, for the sake of convenience, the two different targets TAR₁ and TAR₂ exist in the respective measurement areas A and B. However, for example, one target TAR₁ may exist in the measurement area A or B or the overlap range of the measurement areas A and B.

How each of the sector radars SR₁ and SR₂ constituting the radar apparatus 10 according to the first embodiment is configured and operates will be described with reference to FIGS. 3-5. FIG. 3 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SR_s (s=1, 2) of the radar apparatus 10 according to the first embodiment. FIG. 4 is a block diagram showing, in detail, the internal configuration of each sector radar SR_s (s=1, 2) of the radar apparatus 10 according to the first embodiment. FIG. 5 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of each sector radar SR_s (s=1, 2) of the radar apparatus 10 according to the first embodiment.

In each of the following embodiments, to simplify the description, operations that are common to the sector radars SR₁ and SR₂ will be described in a generic manner and different operations of the sector radars SR₁ and SR₂ will be described individually. Parameter s takes a value 1 or 2 and represents the ordinal number of each sector radar.

The sector radar SR_s transmits, from a transmission antenna Ant-Tx_s, a radio frequency radar transmission signal generated by a radar transmitter Tx_s. The sector radar SRs receives, by a reception antenna Ant-Rx_s, a reflection wave signal, reflected by a target TAR_s, of the radar transmission signal. The sector radar SRs detects presence/absence of a target TAR_s by performing signal processing on the reflection wave signal received by the reception antenna Ant-Rx_s. The target TAR_s is an object to be detected by the sector radar SR_s and is an automobile, a person, or the like. This also applies to each of the following embodiments.

First, how the individual units of the sector radar SR_s are configured will be described in a simplified manner.

The sector radar SR_s shown in FIG. 3 is configured so as to include a reference signal oscillator Lo_s, the radar transmitter Tx_s, and the radar receiver Rx_s. The radar transmitter Tx_s is configured so as to include a transmission signal generater 2_s and a RF transmitter 3_s which is connected to the transmission antenna Ant-Tx_s. The transmission signal generater 2_s is configured so as to include a pulse transmission controller 21_s, a code generater 22_s, a modulater 23_s and an sth transmission phase shifter 25_s. In each of the embodiments including this embodiment, each transmission antenna or each reception antenna may be configured using a transmission antenna element or a reception antenna element.

The radar transmitter Tx_s and the radar receiver Rx_s are connected to the reference signal oscillator Lo_s and are supplied with a reference signal from the reference signal oscillator Lo_s, whereby processing performed by the radar transmitter Tx_s and processing performed by the radar receiver Rx_s are synchronized with each other.

The radar receiver Rx_s is configured so as to have a RF receiver 4_s, a VGA (variable gain amplifier) unit 5_s, and a signal processer 6_s. The signal processer 6_s is configured so as to include an sth reception phase shifter 62, a correlation value calculator 63_s, a coherent integrator 64_s, and a distance estimator 65_s.

(Radar Transmitter)

Next, how the individual units of the radar transmitter Tx are configured will be described in detail with reference to FIG. 4.

The transmission signal generater 2_s is configured so as to include the pulse transmission controller 21_s, the code generater 22_s, the modulater 23_s, an LPF (lowpass filter) 24_s, the sth transmission phase shifter 25_s, and a D/A (digital to analog) converter 26_s. Although in FIG. 4 the transmission signal generater 2_s includes the LPF 24_s, the LPF 24_s, may be provided in the radar transmitter Tx_s independently of the transmission signal generater 2_s. The RF transmitter 3_s is configured so as to include a quadrature modulater 31_s, frequency converter 32_s, and an amplifier 33_s.

Next, how the individual units of the radar transmitter Tx_s operate will be described in detail.

The transmission signal generater 2_s generates a transmission reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the transmission signal generater 2, operate on the basis of the generated transmission reference clock signal. Let f_TxBB represent the transmission reference clock frequency; then the transmission cycle T_r is expressed as an integer N_r multiple of a discrete time interval 1/f_TxBB which is determined by the transmission reference clock frequency f_TxBB (see Equation (5)). The transmission reference clock frequency f_TxBB is a nominal value and, in actuality, includes a frequency error that varies depending on the radar transmitter Tx_s.

$\begin{array}{cc}\left[\mathrm{Formula}5\right]& \\ T_r=N_r\times \frac{1}{f_{\mathrm{TxBB}}}& \left(5\right)\end{array}$

The transmission signal generater 2_s periodically generates a baseband transmission signal G_s(t_s) (see Equation (6)) by modulating a code sequence C_n having a code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21_s every transmission cycle T_r. Parameter n takes values 1 to L, and parameter L represents the code length of the code sequence C_n. Parameter j is the imaginary number unit which satisfies j²=−1. Parameter t_s represents discrete time.

[Formula 6]

G _s(t_s)=1_s(t_s)+jQ_s(t_s)  (6)

As shown in FIG. 5, the transmission signal G_s(t_s) is such that, for example, each code element of the code sequence C_n is modulated using N_o samples of the transmission reference clock signal in the transmission interval T_w(s) of each transmission cycle T_r. Therefore, in each transmission interval T_w, modulation is done using N_w=(=N_o×L) samples. In the silent interval (T_r−T_w) (s) of each transmission cycle T_r, modulation is done using N_u (=N_r−N_w) samples. Therefore, the transmission signal G_s(t_s) of Equation (6) is expressed as Equation (7):

[Formula 7]

G _s(N_r(m_s−1)+t_s)=I_s(N_r(m_s−1)+t_s)+jQ_s(N_r(m_s−1)+t_s)  (7)

The pulse transmission controller 21_s generates a transmission timing signal for a radio-frequency radar transmission signal every transmission cycle T_r and outputs it to each of the code generater 22_s, the sth transmission phase shifter 25_s, and the sth reception phase shifter 62_s.

The code generater 22_s generates a transmission code of the code sequence C_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21_s every transmission cycle T_r. The code generater 22_s outputs the generated transmission code of the code sequence C_n to the modulater 23_s. That is, the single code generater 22_s generates a single code sequence.

For example, the elements of the code sequence C_n are formed using two values [−1, 1] or four values [1, −1, j, −j]. The transmission code is a code sequence of one of, for example, a Barker code sequence, an M-sequence code, and a Gold code sequence which provides a low range sidelobe characteristic. In the following description, for the sake of convenience, the transmission code of the code sequence C_n will be written as a transmission code C_n.

The modulater 23_s receives the transmission code C_n that is output from the code generater 22_s. The modulater 23_s generates a baseband transmission signal G_s(t_s) of Equation (6) by pulse-modulating the received transmission code C_n. The pulse modulation is amplitude modulation (ASK) or phase modulation (PSK). This also applies to each of the following embodiments.

For example, where the code sequence C_n uses two values [−1, 1], the phase modulation (PSK) becomes BPSK (binary phase shift keying). Where the code sequence C_n uses four values [1, −1, j, −j], the phase modulation (PSK) becomes QPSK (quadrature phase shift keying) or 4-phase PSK. That is, in the phase modulation (PSK), prescribed modulation symbols of a constellation on the IQ plane are assigned.

In the baseband transmission signal G_s(t_s) of Equation (6), I_s(t_s) and Q_s(t_s) represent the in-phase component and the quadrate component of a modulation signal, respectively. The modulater 23_s outputs a transmission signal G_s(t_s), in a preset limited band or lower, of the generated transmission signal G_s(t_s) to the sth transmission phase shifter 25_s via the LPF 24_s. The LPF 24_s may be omitted in the transmission signal generater 2_s. This also applies to each of the following embodiments.

Now, how the sth transmission phase shifter 25_s of the specific sector radar SR_s (s=1) will be described. The sth transmission phase shifter 25_s receives the transmission signal G_s(t_s) that is output from the modulater 23, or the LPF 24_s. The transmission phase shifter 25_s gives a prescribed transmission phase shift to the received transmission signal G_s(t_s) every transmission cycle T_r on the basis of a transmission timing signal that is output from the pulse transmission controller 21_s (see FIG. 5).

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21_s in an m_sth transmission cycle T_r, the sth transmission phase shifter 25_s gives the transmission signal G_s(t_s) a transmission phase shift exp(j(m−1)φ) corresponding to the ordinal number of the transmission cycle T_r (see Equation (8)). Parameter m_s represents the ordinal number of the transmission cycle T_r. Parameter φ_s represents a phase rotation amount (e.g., 90°) given by the sth transmission phase shifter 25_s, and it is preferable that parameter φ_s satisfy the relationship of Inequality (9). The sth transmission phase shifter 25_s outputs a transmission-phase-shift-added transmission signal GP_s(N_r(m_s−1)+t_s) to the D/A converter 26_s. Parameter F_dmax will be described later with reference to FIG. 8.

[Formula 8]

GP ₁(N_r(m₁−1)+t₁)=exp(j(m₁−1)φ₁)G₁(N_r(m₁−1)+t₁)  (8).

[Formula 9]

|φ_s|≧2π×(2F_dmax)×T_w  (9)

The manner of operation of the sth transmission phase shifter 25_s of the sector radar SR_s (s=2) is different from that of the sth transmission phase shifter 25_s of the sector radar SR_s (s=1) in that parameter φ₂ representing the phase rotation amount in Equation (10) is different from parameter φ₁. For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

Furthermore, parameter φ₁ in the transmission phase shift given by the sth transmission phase shifter 25_s of the sector radar SR₁ and parameter φ₂ in the transmission phase shift given by the sth transmission phase shifter 25_s of the sector radar SR₂ are opposite in phase (φ₁=φ₂).

[Formula 10]

GP ₂(N_r(m₂−1)+t₂)=exp(j(m₂−1)φ₂)G₂(m₂−1)+t₂)  (10)

The D/A converter 26_s converts the digital transmission signal GP_s(N_r(m_s−1)+t_s) that is output from the sth transmission phase shifter 25_s into an analog transmission signal. The D/A converter 26, outputs the analog transmission signal to the RF transmitter 3_s.

The RF transmitter 3_s generates a transmission reference clock signal in a carrier frequency band by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the RF transmitter 3_s operate on the basis of the generated transmission reference clock signal.

The quadrature modulater 31_s receives the transmission signal from the D/A converter 26_s and quadrature-modulates it. The quadrature modulater 31_s outputs the quadrature-modulated transmission signal to the frequency converter 32_s.

The frequency converter 32_s receives the transmission signal that is output from the quadrature modulater 31_s, and up-converts the baseband transmission signal using the received transmission signal and the transmission reference clock signal. The frequency converter 32_s thus generates a radio-frequency radar transmission signal. The frequency converter 32_s outputs the generated radar transmission signal to the amplifier 33_s.

The amplifier 33_s receives the radar transmission signal that is output from the frequency converter 32_s, amplifies the level of the received radar transmission signal to a prescribed level, and outputs the amplified signal to the transmission antenna Ant-Tx_s. The amplified radar transmission signal is transmitted, that is, radiated to the space, from the transmission antenna Ant-Tx_s.

The transmission antenna Ant-Tx_s transmits, that is, radiates to the space, the radar transmission signal that is output from the RF transmitter 3_s. As shown in FIG. 5, the radar transmission signal is transmitted during the transmission interval T_w of each transmission cycle T_r and is not transmitted during the silent interval T_r-T_w.

The common reference signal generated by the reference signal oscillator Lo_s is supplied to the RF transmitter 3_s and the RF receiver 4_s. This allows the RF transmitter 3_s and the RF receiver 4_s to operate in synchronism with each other.

(Radar Receiver)

Next, how the individual units of the radar receiver Rx_s are configured will be described in detail with reference to FIG. 4.

As shown in FIG. 4, the radar receiver Rx_s is configured so as to include the RF receiver 4_s to which the reception antenna Ant-Rx_s is connected, the VGA unit 5_s, and the signal processer 6_s. The RF receiver 4_s is configured so as to include an amplifier 41_s, a frequency converter 42_s, and a quadrature detector 43_s. The signal processer 6, is configured so as to include an A/D converter 61_s, the sth reception phase shifter 62_s, the correlation value calculator 63_s, the coherent integrator 64_s, and the distance estimator 65_s. Each unit of the signal processer 6_s performs a calculation periodically with each transmission cycle T_r as a signal processing interval.

Next, how the individual units of the radar receiver Rx_s operate will be described in detail.

The reception antenna Ant-Rx_s receives a reflection wave signal that is a radar transmission signal transmitted from the radar transmitter Tx_s and reflected by a target TAR_s and a radar transmission signal coming from the other sector radar which is installed so as to be opposed to the sector radar SR_s concerned. Each reception signal received by the reception antenna Ant-Rx_s is input to the RF receiver 4_s.

Like the RF transmitter 3_s, the RF receiver 4_s generates a reception reference clock signal in the carrier frequency band by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number.

The amplifier 41_s receives the radio-frequency reception signal received by the reception antenna Ant-Rx_s, amplifies the level of the received reception signal, and outputs the resulting signal to the frequency converter 42_s.

The frequency converter 42_s receives the radio-frequency reception signal that is output from the amplifier 41_s, and down-converts the radio-frequency reception signal into a baseband reception signal using the received radio-frequency reception signal and the reception reference clock signal. The frequency converter 42_s thus generates the baseband reception signal and outputs the generated baseband reception signal to the quadrature detector 43_s.

The quadrature detector 43_s generates a baseband reception signal consisting of an in-phase signal (I signal) and a quadrate signal (Q signal) by quadrature-detecting the baseband reception signal that is output from the frequency converter 42_s. The quadrature detector 43_s outputs the generated reception signal to the VGA unit 5_s.

The VGA unit 5, receives the baseband reception signal that is output from the quadrature detector 43 and includes the I signal and the Q signal, and adjusts the output level of the received baseband reception signal into an input range (dynamic range) of the A/D converter 61_s.

The VGA unit 5_s outputs the output-level-adjusted baseband reception signal including the I signal and the Q signal to the A/D converter 61_s. In the embodiment, to simplify the description, it is assumed that the gain of the VGA unit 5_s is adjusted in advance so that the output level of a reception signal falls within the input range (dynamic range) of the A/D converter 61_s.

Like the RF receiver 4_s, the signal processer 6_s generates a reception reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the signal processer 6_s operate on the basis of the generated reception reference clock signal.

Let f_RxBB represent the reception reference clock frequency; then the transmission cycle T_r is expressed as an integer N_v multiple of a discrete time interval 1/f_RxBB which is determined by the reception reference clock frequency f_RxBB (see Equation (11). It is assumed that the transmission reference clock frequency f_TxBB is equal to an integer N_TR multiple of the reception reference clock frequency f_RxBB (see Equation (12). The reception reference clock frequency f_RxBB is a nominal value and, in actuality, includes a frequency error that varies depending on the radar receiver Rx_s.

$\begin{array}{cc}\left[\mathrm{Formula}11\right]& \\ T_r=N_v\times \frac{1}{f_{\mathrm{RxBB}}}& \left(11\right)\\ \left[\mathrm{Formula}12\right]& \\ f_{\mathrm{TxBB}}=N_{\mathrm{TR}}\times f_{\mathrm{RxBB}}& \left(12\right)\end{array}$

The A/D converter 61_s receives the reception signal that is output from the VGA unit 5_s and includes the I signal and the Q signal, and converts the analog data reception signal into digital data by sampling the received reception signal including the I signal and the Q signal every discrete time 1/f_RxBB on the basis of the reception reference clock frequency f_RxBB.

The A/D converter 61_s outputs the digital data reception signal obtained through the conversion done every discrete time k_s to the sth reception phase shifter 62_s in the form of discrete sample values. A reception signal x_s(k_s) which is a converted, discrete sample value is expressed as a complex number (see Equation (13)) using an I signal Ir_s(k_s) and a Q signal Qr_s(k_s) which are discrete sample values at a discrete time k_s:

[Formula 13]

x _s(k_s(=Ir_s(k_s)+jQr_s(k_s)  (13)

Now, a measurement range of the radar apparatus 10 will be described with reference to FIGS. 6 and 7. FIG. 6 is an explanatory diagram illustrating a measurement period of the sector radar SR₁ in a case that each transmission interval of a radar transmission signal transmitted from the sector radar SR₂ includes the start of the corresponding transmission cycle of a radar transmission signal transmitted from the sector radar SR₁. FIG. 7 is an explanatory diagram illustrating a measurement range of the sth sector radar SR_s, and illustrates transmission intervals T_w, transmission cycles T_r, and measurement ranges T_r-T_w of a radar transmission signal, in a specific manner using discrete times ks.

To simplify the description to be made with reference to FIG. 6, it is assumed that the transmission code (indicated by broken lines) of the radar transmission signal is a complementary code and radar transmission signals of a pair of code sequences a_n and b_n of the complementary code are output in transmission intervals T_w of respective transmission cycles T_r in units of two transmission cycles.

Where each transmission interval of a radar transmission signal (solid line) transmitted from the sector radar SR₂ includes the start of the corresponding transmission cycle T_r of a radar transmission signal transmitted from the sector radar SR₁, transmission phase shifts before and after the start of a transmission cycle T_r are different from each other and reception phase shifts before and after the start of the transmission cycle T_r are different from each other. In the radar apparatus 10, where each transmission interval of a radar transmission signal transmitted from the sector radar SR₂ includes the start of the corresponding transmission cycle T_r of a radar transmission signal transmitted from the sector radar SR₁, the interval from the start of each transmission interval of the radar transmission signal transmitted from the sector radar SR₂ to the start of the corresponding transmission cycle T_r of the sector radar SR₁ is excluded from the measurement range.

That is, in the radar apparatus 10, the interval T_s from the start of each transmission interval of the radar transmission signal transmitted from the sector radar SR₂ to the start of the corresponding transmission cycle T_r of the radar transmission signal transmitted from the sector radar SR₁ is excluded from the measurement range. In FIG. 7, the transmission interval T_w corresponds to discrete times k_s=1 to N_w/N_TR, the measurement range interval T_r-T_w corresponds to discrete times k_s=1 to (N_u−N_w)/N_TR which include the transmission interval T_w, and the interval T_s (=T_w) which is outside the measurement range corresponds to discrete times k_s=(N_u−N_w)/N_TR to N_u/N_TR.

The discrete time k_s represents a sampling time of the A/D converter 61_s; discrete times k_s=1 and k=N_v represent a start time point and an end time point of each transmission cycle T_r, respectively. Although the discrete time k_s can take values 1 to N_v, in substance it takes values 1 to (N_u−N_w)/N_TR because the interval T_s outside the measurement range of the transmission cycle T_r of the radar apparatus 10 is not included in the measurement range.

In an m_sth transmission cycle T_r, the reception signal x_s(k_s) of Equation (13) that is output from the A/D converter 61_s can be given by Equation (14) as a complex baseband signal X_s(N_v(m_s−1)+k_s):

[Formula 14]

X _s(N_s(m_s−1)+k_s)=I_rs(N_s(m_s−1)+k_s)+jQ_rs(N_v(m_s−1)+k_s)  (14)

Now, how the sth reception phase shifter 62_s of the specific sector radar SR_s (s=1) will be described. The sth reception phase shifter 62_s receives a reception signal X_s(N_v(m_s−1)+k_s) that is output from the A/D converter 61_s. The sth reception phase shifter 62_s gives a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifter 25_s to the received reception signal X(N_v(m−1)+k_s) every transmission cycle on the basis of a transmission timing signal that is output from the pulse transmission controller 21_s every transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21_s in an m_sth transmission cycle T_r, the sth reception phase shifter 62_s gives a reception phase shift exp(j(m_s−1)(−φ_s)) corresponding to the ordinal number of the transmission cycle T_r to the reception signal X_s(N_v(m_s−1)+k_s) every transmission cycle (see Equation (15)). The sth reception phase shifter 62_s outputs a reception-phase-shift-added reception signal XP_s(N_v(m_s−1)+k_s) to the correlation value calculator 63_s.

[Formula 15]

XP ₁(N_v(m₁−1)+k₁)=exp(−j(m₁−1)φ₁)X₁(N_v(m₁−1)+k₁)  (15)

The sth reception phase shifter 62_s of the sector radar SR_s (s=2) operates differently from that of the sector radar SR_s (s=1) in that the reception phase shift φ₂ is different from φ₁ (see Equation (16)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

[Formula 16]

XP ₂(N_vc(m₂−1)+k₂)=exp(−j(m₂−1)φ₂)X₂(N_v(m₂−1)+k₂)  (16)

The correlation value calculator 63_s receives the reception signal XP_s(N_v(m_s−1)+k_s) that is output from the sth reception phase shifter 62_s. Based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63_s periodically generates, for discrete times k_s, a transmission code of the code sequence C_n having the code length L transmitted in the m_sth transmission cycle T_r.

The correlation value calculator 63_s calculates sliding correlation values AC_s(k_s, m_s) between the received reception signal XP_s(N_v(m_s−1)+k_s) and the transmission code C_n. Each sliding correlation value AC_s(k m_s) is calculated by performing a sliding correlation operation on the transmission code and the reception signal at each discrete time k_s in the m_sth transmission cycle T_r.

More specifically, the correlation value calculator 63_s calculates sliding correlation values AC_s(k_s, m_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth transmission cycle T_r (each transmission cycle T_r) according to Equation (17). The correlation value calculator 63_s outputs the sliding correlation values AC_s(k_s, m_s) calculated according to Equation (17) to the coherent integrator 64_s. In Equation (17), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}17\right]& \\ {\mathrm{AC}}_s\left(k_s,m_s\right)=\sum _{n=1}^L{\mathrm{XP}}_s\left(N_v\left(m_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)C_n^{*}& \left(17\right)\end{array}$

Although in each of the embodiments including this embodiment the correlation value calculator 63_s performs calculations at discrete times k_s=1 to (N_u−N_w)/N_TR, the measurement range (k_s range) may be narrowed further to, for example, k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR in accordance with the range of presence of a target TAR_s to be measured by the radar apparatus 10. With this measure, in the radar apparatus 10, the amount of calculation of the correlation value calculator 63_s can be reduced further. That is, in the radar apparatus 10, the power consumption can be reduced further as a result of reduction in the calculation amount of the signal processer 6_s.

In the radar apparatus 10, where the correlation value calculator 63_s calculates sliding correlation values AC_s(k_s, m_s) at discrete times k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR, measurement of a reflection wave signal in each transmission interval T_w of a radar transmission signal transmitted from each sector radar SR_s can be omitted.

In the radar apparatus 10, even if a radar transmission signal transmitted from each sector radar SR_s goes around to enter the radar receiver Rx_s directly, a measurement can be performed with its influence eliminated. With the above restriction of the measurement range (the range of discrete times k_s), the coherent integrator 64_s and the distance estimator 65_s also operate in the same restricted measurement range.

The coherent integrator 64_s receives the sliding correlation values AC_s(k m_s) that are output from the correlation value calculator 63_s. The coherent integrator 64_s adds together sliding correlation values AC_s(k m_s) in a prescribed number (NP) of transmission cycles T_r (a period NP×T_r) on the basis of sets of sliding correlation values AC_s(k m_s) that have been calculated in the m_sth transmission cycle T_r for the respective discrete times k_s.

The coherent integrator 64_s calculates a v_sth coherent integration value ACC_s(k_s, v_s) for each discrete time k by adding together, for each discrete time k_s, sliding correlation values AC_s(k_s, m_s) in the prescribed number (NP) of transmission cycles T_r (period NP×T_r) according to Equation (18). Parameter NP represents the number of times of coherent integration performed in the coherent integrator 64_s. Parameter v_s is the ordinal number of each set of NP times of coherent integration. The coherent integrator 64_s outputs the calculated coherent integration values ACC_s(k_s, v_s) to the distance estimator 65_s.

$\begin{array}{cc}\left[\mathrm{Formula}18\right]& \\ {\mathrm{ACC}}_s\left(k_s,v_s\right)=\sum _{m_s=1}^{\mathrm{NP}}{\mathrm{AC}}_s\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(18\right)\end{array}$

By setting the prescribed number NP at an integer multiple of 2π/φ_s in Equation (18), the coherent integrator 64_s can reduce influences of the circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number NP at an integer multiple of 2π/φ_s in the sector radar Sly, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR: signal to noise ratio) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing NP times of coherent integration.

FIG. 8( a) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal of a conventional radar apparatus. FIG. 8(b) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal, before reception phase shifting, of each sector radar ST_s of the radar apparatus 10 of this disclosure. FIG. 8(c) is an explanatory diagram showing a relationship between a DC offset component and a Doppler frequency component which are contained in a reception signal, after the reception phase shifting, of each sector radar ST_s of the radar apparatus 10 of this disclosure.

Detection, by a pulse radar, of a Doppler frequency component contained in reflection waves coming from a target is disclosed in the following Referential non-patent document 2, for example:

• (Referential non-patent document 2) Sang-Dong KIM, Jong-Hun LEE, “A Memory-Efficient Hardware Architecture for a Pulse Doppler Radar Vehicle Detector.” IEICE Trans. Fundamentals, Vol. E94-A, No. 5, pp. 1210-1213, 2011.

FIGS. 8( a)-8(c) are of a case that a reception signal contains a DC offset component stationarily. Assuming a moving target TAR_s, f_a represents the Doppler frequency of a reflection wave signal reflected from the target TAR_s, f_dmax represents a maximum value of f_a in the positive direction, and −F_dmax represents a maximum value of f_d in the negative direction.

In the reception signal of the conventional radar apparatus shown in FIG. 8(a), a DC offset component appears in a Doppler spectrum in a range (2F_dmax) of values the Doppler frequency f_d can take in the reflection wave signal. It is therefore difficult to separate the DC offset component and the Doppler spectrum from each other. Therefore, the conventional radar apparatus is affected by the DC offset component, as a result of which the range sidelobe ratio in a reflection wave signal becomes large and the target ranging characteristic is lowered.

The radar transmitter Tx_s of each sector radar SR_s generates a radio-frequency radar transmission signal by giving a transmission phase shift corresponding to each transmission cycle to a baseband transmission signal which uses prescribed code sequences as a compression code.

Let φ_s and T_r represent the phase rotation amount of the transmission phase shift and the transmission cycle, respectively. Then the Doppler spectrum in the range (2F_dmax) of values the Doppler frequency f_d can take is shifted by φ_s/2πT_r as a result of the transmission phase shifting (see FIG. 8(b)). FIG. 8(b) is of a case that Equation (19) holds between the phase rotation amount φ_s, the transmission cycle T_r, and the maximum Doppler frequency F_dmax.

With this measure, as shown in FIG. 8(b), each sector radar SR_s can separate the DC offset component and the Doppler spectrum in the range (2F_dmax) of values the Doppler frequency f_d can take in a reflection wave signal from each other.

[Formula 19]

φ_s=2π×(2F_dmax)×T_r  (19)

Furthermore, in each sector radar SR_s, the radar transmitter Tx_s converts a radio-frequency reception signal into a baseband reception signal and gives the baseband reception signal a reception phase shift whose phase rotation amount is opposite in direction to that of the transmission phase shift.

That is, as shown in FIG. 8(c), each sector radar SR_s shifts, by −φ_s/2πT_r, the DC offset and the Doppler spectrum in the range (2F_dmax) of values the Doppler frequency f_d can take in the reception signal. As a result, the radar apparatus 10 can separate the DC offset component and the Doppler spectrum from each other and suppress influence of the transmission phase shift on the Doppler spectrum.

In the embodiment, interference between a radar transmission signal transmitted from the sector radar SR₁ and a radar transmission signal transmitted from the sector radar SR₂ can be suppressed effectively by performing coherent integration every two transmission cycles by means of each coherent integrator 64_s. A description will be made of how the interference suppression effect is obtained. For example, assume a case that a radar transmission signal transmitted from the sector radar SR₂ arrives at the sector radar SR₁ as an interference wave signal.

The output of the A/D converter 61_s (s=1) is given by Equation (20) in the case where a reception signal of an m₁th transmission cycle T_r of the sector radar SR₁ and a radar transmission signal (interference wave signal) coming from the sector radar SR₂ are involved.

$\begin{array}{cc}\left[\mathrm{Formula}20\right]& \\ X_1\left(N_v\left(m_1-1\right)+k_1\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{\lfloor {\Delta }_1\left\{N_v\left(m_1-1\right)+k_1\right\}/{\Delta }_2\rfloor -\lfloor {\tau }_{12}/{\Delta }_2\rfloor \right\}\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right\}\right)& \left(20\right)\end{array}$

The first term of Equation (20) represents a desired signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₁ as a radar transmission signal, reflected by a target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁. The second term of Equation (20) represents an interference wave signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₂ as a radar transmission signal, reflected by the same target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁.

Parameter h₁₁ is an amplitude and phase complex attenuation coefficient of a case that a radar transmission signal transmitted from the sector radar SR₁ is received by the sector radar SR₁. Parameter h₁₂ is an amplitude and phase complex attenuation coefficient of a case that a radar transmission signal transmitted from the sector radar SR₂ is received by the sector radar SR₁. Parameters m₂ and N_delay are given by Equations (21) and (22), respectively:

[Formula 21]

m ₂=└{└Δ₁{N_v(m₁−1)+k₁}/Δ₂┘−└τ₁₂/Δ₂┘}/N_v┘−1  (21)

[Formula 22]

N _delay=mod {└Δ₁{N_v(m₁−1)+k₁}/Δ₂┘−└τ₁₂/Δ₂┘},N_v}  (22)

Symbol “└x┘” is an operator of outputting the integer part of a real number x. Parameter τ₁₁ is the delay time that is taken by a radar transmission signal transmitted from the sector radar SR₁ to be reflected by a target TAR_s (s=1) and received by the sector radar SR₁. Parameter T₁₂ is the delay time that is taken by a radar transmission signal transmitted from the sector radar SR₂ to be reflected by a target TAR_s (s=2) or travel directly and be received by the sector radar SR₁. To simplify the description, no filter response characteristics of the radar transmitter TX_s and the radar receiver Rx_s of each sector radar SR_s are taken into consideration.

Furthermore, the output of the A/D converter 61_s of the sector radar SR₁ is given by Equation (23) in the case where a reception signal of the sector radar SR₁ in an (m₁+1)th transmission cycle T_r and a radar transmission signal (interference wave signal) coming from the sector radar SR₂ are involved if it is assumed that the propagation environment is the same as in the m₁th transmission cycle T_r. The phrase “the propagation environment is the same as in the m₁th transmission cycle T_r” means that the complex attenuation coefficients h₁₁ and h₁₂ and the delay times τ₁₁ and τ₁₂ can be regarded as remaining unchanged.

$\begin{array}{cc}\left[\mathrm{Formula}23\right]& \\ X_1\left(N_vm_1+k_1\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_vm_1+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_vm_2+N_{\mathrm{delay}}\right\}\right)& \left(23\right)\end{array}$

The addition value of outputs, that is, sliding correlation values, of the correlation value calculator 63_s of the sector radar SR₁ in two transmission cycles, that is, an m₁th transmission cycle and an (m₁+1)th transmission cycle, is given by Equation (24):

$\begin{array}{cc}\left[\mathrm{Formula}24\right]& \\ {\mathrm{AC}}_1\left(k_1,m_1\right)+{\mathrm{AC}}_1\left(k_1,m_1+1\right)=\sum _{n=1}^L{\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}+\sum _{n=1}^L{\mathrm{XP}}_1\left(N_vm_1+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}& \left(24\right)\end{array}$

The outputs of the sth reception phase shifter 62_s of the sector radar SR₁ in the m₁th transmission cycle T_r and the (m₁+1)th transmission cycle T_r are given by Equations (25) and (26), respectively:

$\begin{array}{cc}\left[\mathrm{Formula}25\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)X_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[h_{11}\exp \left(j\left(m_1-1\right){\phi }_1\right)G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp \left(j\left(m_2-1\right){\phi }_2\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\hspace{1em}{h}_{12}\exp \hspace{1em}\left(j\left[\left(m_2-1\right){\phi }_2-\left(m_1-1\right){\phi }_1\right]\right)\hspace{1em}{G}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right)\right]& \left(25\right)\\ \left[\mathrm{Formula}26\right]& \\ {\mathrm{XP}}_1\left(N_vm_1+k_1\right)=\exp \left(-jm_1{\phi }_1\right)X_1\left(N_vm_1+k_1\right)=\exp \left(-jm_1{\phi }_1\right)\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_vm_1+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_vm_2+N_{\mathrm{delay}}\right\}\right)\right]=\exp \left(-jm_1{\phi }_1\right)\left[h_{11}\exp \left(jm_1{\phi }_1\right)G_1\left(N_{\mathrm{TR}}\left\{N_vm_1+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp \left(jm_2{\phi }_2\right)G_2\left(N_{\mathrm{TR}}\left\{N_vm_2+N_{\mathrm{delay}}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_vm_1+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp (j\left[\left(m_2{\phi }_2-m_1{\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_vm_2+N_{\mathrm{delay}}\right)\right]& \left(26\right)\end{array}$

The first term of each of Equations (25) and (26) represents a desired signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₁ as a radar transmission signal, reflected by a target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁. Therefore, the first terms of the respective Equations (25) and (26) are in phase (see Equation (27)) and hence can provide a coherent integration gain when subjected to the coherent integration according to Equation (24). Symbol ∠[x] is an operator of outputting the phase component of a complex number x.

[Formula 27]

∠[h₁₁G₁(N_TR{N_v(m₁−1)+k₁−└τ₁₁/Δ₁┘})]=∠[h₁₁G₁(N_TR{N_vm₁+k₁−└τ₁₁/Δ₁┘})]  (27)

On the other hand, the second term of each of Equations (25) and (26) represents an interference wave signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₂ as a radar transmission signal, reflected by the target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁.

If the carrier frequency errors of the sector radars SR₁ and SR₂ are approximately equal, that is, if Equation (28) holds, the interference wave signal components in the m₁th transmission cycle and the (m₁+1)th transmission cycle are approximately opposite to each other in phase (see Equation (29)). Therefore, the radar apparatus 10 can suppress the interference wave signal components effectively by performing the coherent integration according to Equation (24).

[Formula 28]

N _vΔ₁≅N_vΔ₂  (28)

[Formula 29]

∠h₁₂exp(j[(m₂φ₂−m₁φ₁)])G₂(N_TR{N_vm₂+N_delay)]−∠h₁₂exp(j[(m₂−1)φ₂−(m₁−1)φ₁])G₂(N_TR{N_v(m₂−1)+N_delay)]=φ₂−φ₁+2πf_devTr≅−π  (29)

Parameter f_dev represents the carrier frequency error between the sector radars SR₁ and SR₂ which is defined by a carrier frequency error due to a frequency error of the transmission reference clock signal and a sampling frequency error due to a frequency error of the reception reference clock signal.

For example, assume that the carrier frequency of the RF transmitter 3_s of the sector radar SR₁ is 76 GHz, the carrier frequency error between the sector radars SR₁ and SR₂ is 0.5 ppm (=0.5×10⁻⁶), and the transmission cycle T_r is 300 ns. Even if the measurable distance of the sector radar SR₁ is equal to 45 m (=C₀×T_r/2; C₀: speed of light), the phase variation due to the carrier frequency error f_dev between the sector radars SR₁ and SR₂ is smaller than 5° (see FIG. 9).

That is, as indicated by Equation (30), the phase variation due to the carrier frequency error f_dev between the sector radars SR₁ and SR₂ is calculated as 4.1°, which is about 2.5% of 180° and hence is negligible. The radar apparatus 10 can thus suppress interference wave components by 20 dB or more. FIG. 9 is a graph showing a relationship between the phase rotation amount due to the frequency error between the sector radars SR₁ and SR₂ and the interference suppression amount of an interference signal coming from the other sector radar in the self sector radar.

[Formula 30]

2π×f_dev×T_r=2π×(76 GHz×0.5 ppm)×300 ns≅0.07[rad]=4.1°  (30)

Although the above description assumes a case that an interference wave signal that originates from the sector radar SR₂ arrives at the sector radar SR₁, the same discussion is likewise applicable to a case that an interference wave signal that originates from the sector radar SR₁ arrives at the sector radar SR₂.

The distance estimator 65_s receives coherent integration values ACC_s(k_s, v_s) at respective discrete times k_s that are output from the coherent integrator 64_s every NP transmission cycles T_r. The distance estimator 65_s estimates a distance to the target TAR on the basis of the received coherent integration values ACC_s(k_s, v) at the respective discrete times k_s. For example, the estimation method disclosed in the following Referential non-patent document 3 can be applied to the distance estimation performed in the distance estimator 65_s:

• (Referential non-patent document 3) Bussgang, J. J., Nesbeda, P., and Safran, H., “A Unified Analysis of Range Performance of CW, Pulse, and Pulse Doppler Radar,” Proceedings of the IRE, Vol. 47, Issue 10, pp. 1753-1762, 1959.

The square of the absolute value of each of coherent integration values that are obtained in the v_sth output cycle (v_s×NP×T_r) and supplied from the coherent integrator 64_s, |ACC_s(k_s, v_s)|², corresponds to a reception level of a reflection wave signal at each discrete time k_s. The distance estimator 65_s estimates a distance Range(k_ps) according to Equation (31) on the basis of a detection time k_p, of a peak reception level that is higher than an environment noise level of the sector radar SR_s by a prescribed value or more. In Equation (31), parameter C₀ represents the speed of light.

$\begin{array}{cc}\left[\mathrm{Formula}31\right]& \\ \mathrm{Range}\left(k_{\mathrm{ps}}\right)=C_0\times \left(k_{\mathrm{ps}}-1\right)\times \frac{1}{2f_{R\times \mathrm{BB}}}& \left(31\right)\end{array}$

Operating in the above-described manner, in the case where plural sector radars are installed being opposed to each other, the radar apparatus 10 according to the first embodiment can suppress interference between the sector radars with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other. Furthermore, the radar apparatus 10 can prevent increase of range sidelobes and suppress degradation of the target ranging performance effectively without incorporating circuit error correction circuits even in the case where circuit errors such as a DC offset and IQ imbalance occur.

In the embodiment, the same code sequence C_n having the code length L is used in the sector radars SR₁ and SR₂. However, the invention is not limited to the case of using the same code sequence C_n and different code sequences C(1)_n and C(2)_n having the code length L may be used. In particular, in the radar apparatus 10, interference between the sector radars SR_s can be suppressed further if the code generaters 21_s of the respective sector radars SR₁ and SR₂ employ code sequences whose cross-correlation is low as the different code sequences C(1)_n and C(2)_n.

The embodiment may be modified so that different code sequences C(1)_n1 and C(2)_n2 having different code lengths L₁ and L₂ are used in the sector radars SR₁ and SR₂, respectively. In particular, in the radar apparatus 10, interference between the sector radars SR_s can be suppressed further if the code generaters 21_s of the respective sector radars SR₁ and SR₂ employ code sequences whose cross-correlation is low as the different code sequences C(1)_n1 and C(2)_n2. Since the sector radars SR₁ and SR₂ transmit radar transmission signals of the different code sequences C(1)_n1 and C(2)_n2, the radar transmission signals transmitted from the respective sector radars SR₁ and SR₂ have different transmission intervals (see FIGS. 10(a) and 10(b)).

FIG. 10( a) is an explanatory diagram illustrating a measurement range of the sector radar SR₁ in a case that the transmission codes used in the respective sector radars SR₁ and SR₂ have different code lengths. FIG. 10(b) is an explanatory diagram illustrating a measurement range of the sector radar SR₂ in a case that the transmission codes used in the respective sector radars SR₁ and SR₂ have different code lengths.

In the radar apparatus 10, where each transmission interval of a radar transmission signal transmitted from the sector radar SR₂ includes the start of the corresponding transmission cycle T_r of a radar transmission signal transmitted from the sector radar SR₁, the interval from the start of each transmission interval of the radar transmission signal transmitted from the sector radar SR₂ to the start of the corresponding transmission cycle T_r of the sector radar SR₁ is excluded from the measurement range.

That is, as shown in FIG. 10(a), in the sector radar SR₁, the transmission interval T_w1 corresponds to discrete times k_s=1 to N_w1/N_TR, the measurement range interval T_r−T_w2 corresponds to discrete times k_s=1 to (N_u−N_w2)/N_TR which include the transmission interval T_w1, and the interval T_s (=T_w1) which is outside the measurement range corresponds to discrete times k_s=(N_u−N_w2)/N_TR to N_u/N_TR.

As shown in FIG. 10(b), in the sector radar SR₂, the transmission interval T_w2 corresponds to discrete times k_s=1 to N_w2/N_TR, the measurement range interval T_r−T_w1 corresponds to discrete times k_s=1 to (N_u−N_w1)/N_TR which include the transmission interval T_w2, and the interval T_s (=T_w2) which is outside the measurement range corresponds to discrete times k_s=(N_u−N_wt)/N_TR to N_u/N_TR.

Furthermore, in the embodiment, the transmission phase shift φ₁ of the sth transmission phase shifter 25_s of the sector radar SR₁ is set at 90° and the reception phase shift φ₂ of the sth transmission phase shifter 25_s of the sector radar SR₂ is set at −90°, φ₁ and φ₂ are not restricted to 90° and −90°, respectively.

The sth transmission phase shifter 25_s of the sector radar SR₁ and the sth transmission phase shifter 25_s of the sector radar SR₂ give the different phase shifts (φ₁, φ₂)=(φ(q, N_i)+α, −φ(q, N_i)+α) (=(qπ/N_i+α, −qπ/N_i+α))). With this measure, each of the sector radars SR₁ and SR₂ which are installed being opposed to each other can suppress an interference wave signal coming from the other sector radar, and can prevent increase of range sidelobes and suppress degradation of the target ranging performance effectively without incorporating circuit error correction circuits even in the case where circuit errors such as a DC offset and IQ imbalance occur.

Parameter q takes values 1 to N_i, parameter N_i is a natural number that is larger than or equal to 2, and parameter α is a fixed phase value. By performing coherent integration every N_i transmission cycles, each coherent integrator 64_s can effectively suppress interference between radar transmission signals coming from the respective sector radars SR₁ and SR₂.

For example, where N_i=3, q=1, and α=0, the phase shifts (φ₁, φ₂)=(φ(1, 3), −φ(1, 3)) become (π/3, −π/3). Where N_i=3, q=2, and α=0, the phase shifts (φ₁, φ₂)=φ(2, 3), −φ(2, 3)) become (2π/3, −2π/3). Performing coherent integration every three transmission cycles, each coherent integrator 64_s can effectively suppress interference between radar transmission signals coming from the respective sector radars SR₁ and SR₂.

How the interference suppression effect is obtained will be described in a general case of N_i transmission cycles instead of three transmission cycles. Assume an example case that the sector radar SR₁ receives a radar transmission signal of the sector radar SR₂ as an interference wave signal.

The output of the A/D converter 61_s (s=1) is given by Equation (20) in the case where a reception signal of an m₁th transmission cycle T_r of the sector radar SR₁ and a radar transmission signal (interference wave signal) coming from the sector radar SR₂ are involved.

Furthermore, the output of the A/D converter 61_s of the sector radar SR₁ is given by Equation (32) in the case where a reception signal of the sector radar SR₁ in each of an (m₁+1)th to (m₁+(N₁−1))th transmission cycles T_r and a radar transmission signal (interference wave signal) coming from the sector radar SR₂ are involved if the propagation environment remains the same as in the m₁th transmission cycle T_r. In Equation (32), parameter w takes values 1 to (N_i−1).

[Formula 32]

X ₁(N_v(m₁+w−1)+k₁)=h₁₁GP₁(N_TR{N_v(m₁+w−1)+k₁−└τ₁₁/Δ₁┘})+h₁₂GP₂(N_TR{N_v(m₂+w−1)+N_delay})  (32)

The output, that is, the addition value of sliding correlation values, of the correlation value calculator 63_s of the sector radar SR₁ in the m₁th to (m₁+(N_i−1))th transmission cycles is given by Equation (33):

$\begin{array}{cc}\left[\mathrm{Formula}33\right]& \\ {\mathrm{AC}}_1\left(k_1,m_1\right)+\sum _{w=1}^{\mathrm{Ni}-1}{\mathrm{AC}}_1\left(k_1,m_1+w-1\right)=\sum _{n=1}^L{\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}+\sum _{w=1}^{\mathrm{Ni}-1}\sum _{n=1}^L{\mathrm{XP}}_1\left(N_v\left(m_1+w-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}& \left(33\right)\end{array}$

The outputs of the sth reception phase shifter 62_s of the sector radar SR₁ in the m₁th transmission cycle T_r and the (m₁+w)th transmission cycle T_r are given by Equations (34) and (35), respectively:

$\begin{array}{cc}\left[\mathrm{Formula}34\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)X_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=\hspace{1em}\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[h_{11}\exp \left(j\left(m_1-1\right){\phi }_1\right)G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp \left(j\left(m_2-1\right){\phi }_2\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp \left(j\left[\left(m_2-1\right){\phi }_2-\left(m_1-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right)\right]& \left(34\right)\\ \left[\mathrm{Formula}35\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1+w-1\right)+k_1\right)=\exp \left(-j\left(m_1+w-1\right){\phi }_1\right)X_1\left(N_v\left(m_1+w-1\right)+k_1\right)=\exp \left(-j\left(m_1+w-1\right){\phi }_1\right)\hspace{1em}\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=\exp \left(-j\left(m_1+w-1\right){\phi }_1\right)\left[h_{11}\exp \left(j\left(m_1+w-1\right){\phi }_1\right)G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp \left(j\left(m_2+w-1\right){\phi }_2\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}\exp (j\left[\left(\left(m_2+w-1\right){\phi }_2-\left(m_1+w-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}\right)\right]& \left(35\right)\end{array}$

The first term of each of Equations (34) and (35) represents a desired signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₁ as a radar transmission signal, reflected by a target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁. Therefore, the first terms of the respective Equations (34) and (35) are in phase (see Equation (36)) and hence can provide a coherent integration gain when subjected to the coherent integration according to Equation (33). Symbol ∠[x] is an operator of outputting the phase component of a complex number x.

[Formula 36]

∠[h₁₁G₁(N_TR{N_s(m₁−1)+k₁−└τ₁₁/Δ₁┘})]=∠[h₁₁G₁(N_TR{N_v(m₁+w−1)+k₁−└τ₁₁/Δ₁┘})]  (36)

On the other hand, the second term of each of Equations (34) and (35) represents an interference wave signal component that is transmitted from the radar transmitter TX_s of the sector radar SR₂ as a radar transmission signal, reflected by the target TAR_s, and received by the radar receiver RX_s of the sector radar SR₁.

If the carrier frequency error between the sector radars SR₁ and SR₂ is within an allowable range, that is, if Equation (28) holds, the interference wave signal components in the m₁th to (m₁+w)th transmission cycles have a phase relationship indicated by Equation (37). Equation (38) represents a result of coherent integration performed on interference wave signal components by the coherent integrator 64_s of the sector radar SR₁. Therefore, in the radar apparatus 10, the interference components have such a phase relationship as to be canceled out each other as is understood from Equation (38) and hence the interference wave signal components can be suppressed effectively. However, the radar apparatus 10 becomes more prone to be affected by phase variations due to the frequency error f_dev as N_i increases. Therefore, N_i has an upper limit that depends on the frequency accuracy of the reference clock signals used in the radar apparatus 10.

$\begin{array}{cc}\left[\mathrm{Formula}37\right]& \\ \angle h_{12}\exp (j\left[\left(m_2+w-1\right){\phi }_2-\left(\left(m_1+w-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}\right)\right]-\angle h_{12}\exp \left(j\left[\left(m_2-1\right){\phi }_2-\left(m_1-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right)\right]=w\left\{{\phi }_2-{\phi }_1\right\}+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r=-2\frac{\pi q}{N_i}w+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r& \left(37\right)\\ \left[\mathrm{Formula}38\right]& \\ {\left\{{\mathrm{AC}}_1\left(k_1,m_1\right)+\sum _{w=1}^{\mathrm{Ni}-1}{\mathrm{AC}}_1\left(k_1,m_1+w-1\right)\right\}}_{\mathrm{INTERFERENCE}}={\left\{{\mathrm{AC}}_1\left(k_1,m_1\right)\right\}}_{\mathrm{INTERFERENCE}}\left\{1+\sum _{w=1}^{\mathrm{Ni}-1}\exp \left(jw\left({\phi }_2-{\phi }_1\right)+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r\right)\right\}={\left\{{\mathrm{AC}}_1\left(k_1,m_1\right)\right\}}_{\mathrm{INTERFERENCE}}\left\{1+\sum _{w=1}^{\mathrm{Ni}-1}\exp \left[-j\left(\frac{2\pi q}{N_i}w+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r\right)\right]\right\}\cong 0& \left(38\right)\end{array}$

Although the above description assumes the case that an interference wave signal that originates from the sector radar SR₂ arrives at the sector radar SR₁, the same discussion is likewise applicable to a case that an interference wave signal that originates from the sector radar SR₁ arrives at the sector radar SR₂.

(Modification of Embodiment 1)

In a modification of the first embodiment, the sth reception phase shifter 62_s of the sector radar SR_s is modified so as to give a reception phase shift to sliding correlation values AC_s(k_s, m_s) that are output from the correlation value calculator 63_s (see FIG. 11).

FIG. 11 is a block diagram showing, in detail, the internal configuration of a sector radar SRa_s of a radar apparatus 10 according to the modification of the first embodiment. Units (blocks) of the sector radar SRa_s having the same (in configuration and operation) units in the sector radar SR_s will be given the same reference symbols as the latter, and their configurations and operations will not be described (only differences will be described).

As shown in FIG. 11, a radar receiver Rxa_s is configured so as to include the RF receiver 4_s, the VGA unit 5_s, and a signal processer 6a_s. The signal processer 6a_s is configured so as to include the A/D converter 61_s, a correlation value calculator 63a_s, an sth reception phase shifter 62a_s, a coherent integrator 64a_s, and the distance estimator 65_s.

The correlation value calculator 63a, receives a reception signal X_s(N_v(m_s−1)+k_s) that is output from the A/D converter 61_s. Based on a reception reference clock signal obtained by multiplying a reference signal by a prescribed number, the correlation value calculator 63a_s periodically generates, for discrete times k_s, a transmission code of a code sequence C_n having a code length L transmitted in an m_sth transmission cycle T_r.

The correlation value calculator 63a, calculates sliding correlation values AC_s(k_s, m_s) between the received reception signal X_s(N_v(m_s−1)+k_s) and the transmission code C_n.

More specifically, the correlation value calculator 63a, calculates sliding correlation values AC_s(k m_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth transmission cycle T_r (each transmission cycle T_r) according to Equation (39). The correlation value calculator 63a_s outputs the sliding correlation values AC_s(k_s, m_s) calculated according to Equation (39) to the sth reception phase shifter 62a_s. In Equation (39), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}39\right]& \\ {\mathrm{AC}}_s\left(k_s,m_s\right)=\sum _{n=1}^LX_s\left(N_v\left(m_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)C_n^{*}& \left(39\right)\end{array}$

Now, how the sth reception phase shifter 62a_s of the specific sector radar SRa_s (s=1) will be described. The sth reception phase shifter 62a_s receives the sliding correlation values AC_s(k_s, m_s) that are output from the correlation value calculator 63a_s. The sth reception phase shifter 62a_s gives a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifter 25_s to the received sliding correlation values AC_s(k_s, m_s) every transmission cycle on the basis of a transmission timing signal that is supplied from the pulse transmission controller 21_s in the m_sth transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the sth pulse transmission controller 21_s in the m_sth transmission cycle T_r, the reception phase shifter 62a_s gives a reception phase shift exp(j(m_s−1)(−φ)) corresponding to the ordinal number of the transmission cycle T_r to the sliding correlation values AC_s(k_s, m_s) every transmission cycle (see Equation (40)). The sth reception phase shifter 62a_s outputs reception-phase-shift-added sliding correlation values ACP_s(k_s, m_s) to the coherent integrator 64a_s.

[Formula 40]

ACP ₁(k₁,m₁)=exp(−j(m₁−1)φ₁)AC₁(k₁,m₁)  (40)

The sth reception phase shifter 62a_s of the sector radar SRa_s (s=2) operates differently from that of the sector radar SRa_s (s=1) in that parameter φ₂ representing the phase rotation amount is different from φ₁ (see Equation (41)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

[Formula 41]

ACP ₂(k₂,m₂)=exp(−j(m₂−1)φ₂)AC₂(k₂,m₂)  (41)

The coherent integrator 64a_s receives the sliding correlation values ACP_s(k_s, m_s) that are output from the sth reception phase shifter 62a_s. The coherent integrator 64a_s adds together, for each discrete time k_s, sliding correlation values ACP_s(k_s, m_s) in a prescribed number (NP) of transmission cycles T_r (a period NP×T_r) on the basis of sets of sliding correlation values ACP_s(k_s, m_s) that have been calculated in the m_yth transmission cycle T_r for the respective discrete times k_s.

The coherent integrator 64a_s calculates a v_sth coherent integration value ACC_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values ACP_s(k_s, m_s) in the prescribed number (NP) of transmission cycles T_r (period NP×T_r) according to Equation (42). Parameter NP represents the number of times of coherent integration performed in the coherent integrator 64a_s. The coherent integrator 64a_s outputs the calculated coherent integration values ACC_s(k_s, v_s) to the distance estimator 65_s.

$\begin{array}{cc}\left[\mathrm{Formula}42\right]& \\ {\mathrm{ACC}}_s\left(k_s,v_s\right)=\sum _{m_s=1}^{\mathrm{NP}}{\mathrm{ACP}}_s\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(42\right)\end{array}$

By setting the prescribed number NP at an integer multiple of 2π/φ_s in Equation (42), the coherent integrator 64a_s can reduce influences of circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number NP at an integer multiple of 2π/φ_s in the sector radar SRa_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing NP times of coherent integration.

As such, the radar apparatus 10 according to the modification of the first embodiment can provide the same advantages as the radar apparatus 10 according to the first embodiment.

Embodiment 2

The first embodiment is directed to the case of using, as the transmission code, one of code sequences capable of providing a low range sidelobe characteristic such as a Barker code sequence, an M-sequence code, and a Gold code sequence. A second embodiment is directed to a case of using a complementary code as the transmission code.

How each of sector radars SRb_s (s=1, 2) constituting a radar apparatus 10 according to the second embodiment is configured and operates will be described with reference to FIGS. 12-14. FIG. 12 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRb_s (s=1, 2) of the radar apparatus 10 according to the second embodiment. FIG. 13 is a block diagram showing, in detail, the internal configuration of each sector radar SRb_s (s=1, 2) of the radar apparatus 10 according to the second embodiment. FIG. 14 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of a radar transmission signal of each sector radar SRb_s (s=1, 2) of the radar apparatus 10 according to the second embodiment.

Units (blocks) of the sector radar SRb_s having the same (in configuration and operation) units in the sector radar SR_s will be given the same reference symbols as the latter, and their configurations and operations will not be described (only differences will be described).

As shown in FIG. 12, the radar apparatus SRb_s is configured so as to include the reference signal oscillator Lo_s, a radar transmitter Txb_s, and a radar receiver Rxb_s. The radar transmitter Txb_s is configured so as to have a transmission signal generater 2b_s and the RF transmitter 3_s which is connected to the transmission antenna Ant-Tx_s. The transmission signal generater 2b_s is configured so as to include a pulse transmission controller 21b_s, a code generater 22b_s, a modulater 23b_s, and an sth transmission phase shifter 25b_s. The code generater 22b_s includes, as at least one code generater, a first code generater 22b1_s and a second code generater 22b2_s and generates at least one code sequence.

The radar transmitter Txb_s and the radar receiver Rxb_s are connected to the reference signal oscillator Lo_s and are supplied with a reference signal from the reference signal oscillator Lo_s, whereby processing performed by the radar transmitter Txb_s and processing performed by the radar receiver Rxb_s are synchronized with each other.

The radar receiver Rxb_s is configured so as to have the RF receiver 4_s, the VGA unit 5_s, and a signal processer 6b_s. The signal processer 6b_s is configured so as to include an sth reception phase shifter 62b_s a correlation value calculator 63b_s a coherent integrator 64b_s, and the distance estimator 65_s.

(Radar Transmitter)

Next, how the individual units of the radar transmitter Txb_s are configured and operate will be described in detail with reference to FIG. 13.

The transmission signal generater 2b_s is configured so as to include the pulse transmission controller 21b_s the code generater 22b_s the modulater 23b_s, the LPF 24_s, the sth transmission phase shifter 25b_s, and the D/A converter 26_s. Although in FIG. 12 the transmission signal generater 2b_s includes the LPF 24_s, the LPF 24_s may be provided in the radar transmitter Txb_s independently of the transmission signal generater 2b_s. The configuration and the manner of operation of the RF transmitter 3_s are the same as those of the RF transmitter 3_s of the sector radar SR_s, and hence descriptions therefor will be omitted.

Next, how the individual units of the radar transmitter Txb_s operate will be described in detail.

The transmission signal generater 2b_s generates a transmission reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the transmission signal generater 2b_s operate on the basis of the generated transmission reference clock signal. Let f_TxBB represent the transmission reference clock frequency; then the transmission cycle T_r is expressed as an integer N_r multiple of a discrete time interval 1/f_TxBB which is determined by the transmission reference clock frequency f_TxBB (see Equation (5)).

The transmission signal generater 2b_s periodically generates a baseband transmission signal G_s(t_s) (see Equation (6)) by modulating a complementary code sequence A_n or B_n having a code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21b_s every transmission cycle T_r. Parameter n takes values 1 to L, and parameter L represents the code length of each of the code sequences A_n and B_n. Parameter j is the imaginary number unit which satisfies j²=−1. Parameter t_s represents discrete time.

For example, as shown in FIG. 14, the transmission signal G_s(n_s) is such that each code element of the complementary code sequence A_n or B_n is modulated using N_o samples of the transmission reference clock signal in the transmission interval T_w (s) of each transmission cycle T_r. Therefore, in each transmission interval T_w, modulation is done using N_w (=N_o×L) samples. In the silent interval (T_r−T_w) (s) of each transmission cycle T_r, modulation is done using N_u (=N_r−N_w) samples. Therefore, the transmission signal G_s(n_s) of Equation (6) is expressed as Equation (7).

The pulse transmission controller 21b_s generates a transmission timing signal for a radio-frequency radar transmission signal every transmission cycle T_r and outputs it to each of the code generater 22b_s, the sth transmission phase shifter 25b_s, and the sth reception phase shifter 62b_s.

The first code generater 22b1_s generates a transmission code of the one complementary code sequence A_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every odd-numbered transmission cycle T_r. The first code generater 22b1_s outputs the generated transmission code of the complementary code sequence A_n to the modulater 23b_s. In the following description, for the sake of convenience, the transmission code of the complementary code sequence A_n will be written as a transmission code A_n.

The second code generater 22b2_s generates a transmission code of the other complementary code sequence B_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every even-numbered transmission cycle T_r. The second code generater 22b2_s outputs the generated transmission code of the complementary code sequence B_n to the modulater 23b_s. In the following description, for the sake of convenience, the transmission code of the complementary code sequence B_n will be written as a transmission code B_n.

It has been described above that in this embodiment the first code generater 22b1_s generates a complementary code sequence A_n having a code length L and the second code generater 22b2_s generates a complementary code sequence B_n having a code length L. However, the first code generater 22b1_s and the second code generater 22b2_s may generate a complementary code sequence B_n having a code length L and a complementary code sequence A_n having a code length L, respectively.

The modulater 23b_s receives the transmission code A_n or B_n that is output from the code generater 22_s. The modulater 23b_s generates a baseband transmission signal G_s(t_s) of Equation (6) by pulse-modulating the received transmission code A_n or B_n. The modulater 23b_s outputs a transmission signal G_s(t_s), in a preset limited band or lower, of the generated transmission signal G_s(n_s) to the sth transmission phase shifter 25b_s via the LPF 24_s.

Now, how the sth transmission phase shifter 25b_s of the specific sector radar SRb_s (s=1) will be described. The sth transmission phase shifter 25b_s receives the transmission signal G_s(t_s) that is output from the modulater 23b_s or the LPF 24_s. The sth transmission phase shifter 25b_s gives a prescribed transmission phase shift to the received transmission signal G_s(t_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every transmission cycle T_r (see FIG. 14).

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21b_s in an m_sth transmission cycles T_r, the sth transmission phase shifter 25b_s gives a transmission phase shift exp(j·floor[(m_s−1)/2]φ_s) corresponding to the ordinal number of the transmission cycle T_r to the transmission signal G_s(t_s) every two transmission cycles (see Equation (43)). Parameter φ_s represents a phase rotation amount (e.g., 90°) that is given in the sth transmission phase shifter 25b_s, and it is preferable that parameter φ_s satisfy the relationship of Inequality (9). The sth transmission phase shifter 25b_s outputs a transmission-phase-shift-added transmission signal GP_s(N_r(m_s−1)+t_s) to the D/A converter 26_s. Symbol floor[x] is an operator of outputting an integer obtained by rounding down a real number x.

$\begin{array}{cc}\left[\mathrm{Formula}43\right]& \\ {\mathrm{GP}}_1\left(N_r\left(m_1-1\right)+t_1\right)=\exp \left(j\mathrm{floor}\left[\frac{m_1-1}{2}\right]{\phi }_1\right)G_1\left(N_r\left(m_1-1\right)+t_1\right)& \left(43\right)\end{array}$

The manner of operation of the sth transmission phase shifter 25b_s of the sector radar SRb_s (s=2) is different from that of the sth transmission phase shifter 25b_s of the sector radar SRb_s (s=1) in that parameter φ_s representing the phase rotation amount in the transmission phase shift exp(j·floor[(m_s−1)/2]φ_s) in Equation (44) is different from parameter φ₁. For example, parameters φ₁ and φ₂ are 90° and −90°, respectively. That is, parameters φ₁ (s=1) and φ₂ (s=2) are opposite in phase (φ₁=φ₂).

$\begin{array}{cc}\left[\mathrm{Formula}44\right]& \\ {\mathrm{GP}}_2\left(N_r\left(m_2-1\right)+t_2\right)=\exp \left(j\mathrm{floor}\left[\frac{m_2-1}{2}\right]{\phi }_2\right)G_2\left(N_r\left(m_2-1\right)+t_2\right)& \left(44\right)\end{array}$

(Radar Receiver)

Next, how the individual units of the radar receiver Rxb_s are configured will be described in detail with reference to FIG. 13.

As shown in FIG. 13, the radar receiver Rxb_s is configured so as to include the RF receiver 4_s to which the reception antenna Ant-Rx_s is connected, the VGA unit 5 and the signal processer 6b_s. The signal processer 6b_s is configured so as to include the A/D converter 61_s, the sth reception phase shifter 62b_s, the correlation value calculator 63b_s, the coherent integrator 64b_s, and the distance estimator 65_s. Each unit of the signal processer 6b_s performs a calculation periodically with each transmission cycle T_r as a signal processing interval.

Next, how the individual units of the radar receiver Rxb_s operate will be described in detail.

Like the RF receiver 4_s, the signal processer 6b_s generates a reception reference clock signal by multiplying a reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the signal processer 6b_s operate on the basis of the generated reception reference clock signal.

Now, how the sth reception phase shifter 62b_s of the specific sector radar SRb_s (s=1) will be described. The sth reception phase shifter 62b_s receives a reception signal X_s(N_v(m_s−1)+k_s) that is output from the A/D converter 61_s. The sth reception phase shifter 62b_s gives a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifter 25b_s to the received reception signal X_s(N_v(m_s−1)+k_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21b_s in an m_sth transmission cycle T_r, the reception phase shifter 62b_s gives a reception phase shift exp(−j·floor[(m_s−1)/2](−φ_s)) corresponding to the ordinal number of the transmission cycle T_r to the reception signal X_s(N_v(m_s−1)+k_s) every two transmission cycles (see Equation (45)). The sth reception phase shifter 62b_s outputs a reception-phase-shift-added reception signal XP_s(N_v(m_s−1)+k) to the correlation value calculator 63b_s.

$\begin{array}{cc}\left[\mathrm{Formula}45\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_1-1}{2}\right]{\phi }_1\right)X_1\left(N_v\left(m_1-1\right)+k_1\right)& \left(45\right)\end{array}$

The sth reception phase shifter 62b_s of the sector radar SR_s (s=2) operates differently from that of the sector radar SR_s (s=1) in that parameter φ₂ representing a reception rotation amount is different from φ₁ (see Equation (46)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

$\begin{array}{cc}\left[\mathrm{Formula}46\right]& \\ {\mathrm{XP}}_2\left(N_v\left(m_2-1\right)+k_2\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_2-1}{2}\right]{\phi }_2\right)X_2\left(N_v\left(m_2-1\right)+k_2\right)& \left(46\right)\end{array}$

The correlation value calculator 63b_s receives the reception signal XP_s(N_v(m_s1)+k_s) that is output from the sth reception phase shifter 62b_s. Based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63b_s periodically generates, for discrete times k_s, a transmission code of the code sequence A_n having the code length L transmitted in an m_sth transmission cycle T_r (m_s (odd number)=2z_s−1 where z_s is a natural number).

Furthermore, based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63b_s periodically generates, for discrete times k_s, a transmission code of the code sequence B_n having the code length L transmitted in an m_sth transmission cycle T_r (m_s (even number)=2z_s).

The correlation value calculator 63b_s calculates sliding correlation values AC_s(k_s, m_s) between the received reception signal XP_s(N_v(m_s−1)+k_s) and the transmission code A_n or B_n. Each sliding correlation value AC_s(k_s, m) is calculated by performing a sliding correlation operation on the transmission code and the reception signal at each discrete time k_s in the m_sth transmission cycle T_r.

More specifically, the correlation value calculator 63b_s calculates sliding correlation values AC_s(k_s, 2z_s−1) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth (m_s (odd number)=2z_s−1) transmission cycle T_r (each transmission cycle T_r) according to Equation (47). The correlation value calculator 63b_s outputs the sliding correlation values AC_s(k_s, 2z_s−1) calculated according to Equation (47) to the coherent integrator 64b_s. In Equation (47), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}47\right]& \\ {\mathrm{AC}}_s\left(k_s,2z_s-1\right)=\sum _{n=1}^L{\mathrm{XP}}_s\left(2N_v\left(z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)A_n^{*}& \left(47\right)\end{array}$

Furthermore, the correlation value calculator 63b_s calculates sliding correlation values AC_s(k_s, 2z_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth (m_s (even number)=2z_s) transmission cycle T_r (each transmission cycle T_r) according to Equation (48). The correlation value calculator 63b_s outputs the sliding correlation values AC_s(k_s, 2z_s) calculated according to Equation (48) to the coherent integrator 64b_s. In Equation (48), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}48\right]& \\ {\mathrm{AC}}_s\left(k_s,2z_s\right)=\sum _{n=1}^L{\mathrm{XP}}_s\left(N_v\left(2z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)B_n^{*}& \left(48\right)\end{array}$

Although in each of the embodiments including this embodiment the correlation value calculator 63b_s performs calculations at discrete times k_s=1 to (N_u−N_w)/N_TR, the measurement range (discrete time k_s range) may be narrowed further to, for example, k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR according to the range of presence of a target TAR_s which is a measurement target of the radar apparatus 10. With this measure, the radar apparatus 10 can further reduce the amount of calculation of the correlation value calculator 63b_s. That is, the radar apparatus 10 can reduce the power consumption further as a result of reduction in the calculation amount of the signal processer 6b_s.

Where the correlation value calculator 63b_s calculates sliding correlation values AC_s(k_s, m_s) at discrete times k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR, the radar apparatus 10 can omit measurement of a reflection wave signal in each transmission interval T_w of the radar transmission signal.

In the radar apparatus 10, even if a radar transmission signal coming from the radar transmitter Txb_s of each sector radar SR_s goes around to enter the radar receiver Rxb_s directly, a measurement can be performed with its influence eliminated. With the above restriction of the measurement range (discrete time k, range), the coherent integrator 64b_s and the distance estimator 65b_s also operate in the same restricted measurement range.

The coherent integrator 64b_s receives the sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s) that are output from the correlation value calculator 63b_s. The coherent integrator 64b_s adds together sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s) in a prescribed number (2NP) of transmission cycles T_r (a period 2NP×T_r) on the basis of sets of sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s) that have been calculated in the two (odd-numbered and even-numbered) transmission cycles T_r for the respective discrete times k_s.

The coherent integrator 64b_s calculates a v_sth coherent integration value ACC_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s) in the prescribed number 2NP of periods (period 2NP×T_r) according to Equation (49). Parameter 2NP represents the number of times of coherent integration performed in the coherent integrator 64b_s. The coherent integrator 64b_s outputs the calculated coherent integration values ACC_s(k_s, v_s) to the distance estimator 65_s.

$\begin{array}{cc}\left[\mathrm{Formula}49\right]& \\ {\mathrm{ACC}}_s\left(k_s,v_s\right)=\sum _{m_s=1}^{2\mathrm{NP}}{\mathrm{AC}}_s\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(49\right)\end{array}$

By setting the prescribed number 2NP at an integer multiple of 2πφ_s in Equation (49), the coherent integrator 64b_s can reduce influences of circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number 2NP at an integer multiple of 2π/φ_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing 2NP times of coherent integration.

As such, the radar apparatus 10 according to the second embodiment can provide advantages that are equivalent to the advantages of the radar apparatus 10 according to the first embodiment even in the case of using a complementary code as a transmission code.

(Modification of Embodiment 2)

In a modification of the second embodiment, as in the modification of the first embodiment, the sth reception phase shifter 62b_s used in the second embodiment is modified so as to give a reception phase shift to sliding correlation values AC_s(k_s, 2z_s) and AC_s(k_s, 2z_s−1) that are output from the correlation value calculator 63b_s (see FIG. 15).

FIG. 15 is a block diagram showing, in detail, the internal configuration of a sector radar SRc_s of a radar apparatus 10 according to the modification of the second embodiment. Units (blocks) of the sector radar SRc_s having the same (in configuration and operation) units in the sector radar SRb_s will be given the same reference symbols as the latter, and their configurations and operations will not be described (only differences will be described).

As shown in FIG. 15, a radar receiver Rxc_s is configured so as to include the RF receiver 4_s, the VGA unit 5_s, and a signal processer 6c_s. The signal processer 6c_s is configured so as to include the A/D converter 61_s, a correlation value calculator 63c_s, an sth reception phase shifter 62c_s, a coherent integrator 64c_s, and the distance estimator 65_s.

The correlation value calculator 63c_s receives a reception signal X_s(N_v(m_s−1)+k_s) that is output from the A/D converter 61_s. Based on a reception reference clock signal obtained by multiplying a reference signal by a prescribed number, the correlation value calculator 63c_s periodically generates, for discrete times k_s, a transmission code of a code sequence A_n having a code length L transmitted in an m_sth transmission cycle T_r (m_s (odd number)=2z_s−1).

Based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63c, periodically generates, for discrete times k_s, a transmission code of a code sequence B_n having the code length L transmitted in an m_sth transmission cycle T_r (m_s (even number)=2z_s). The correlation value calculator 63c_s calculates sliding correlation values AC_s(k_s, m_s) between the received reception signal X_s(N_v(m_s−1)+k_s) and the pulse compression code A_n or B_n.

More specifically, the correlation value calculator 63c_s calculates sliding correlation values AC_s(k_s, 2z_s−1) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth transmission cycle T_r (each transmission cycle T_r; m_s: odd number) according to Equation (50). The correlation value calculator 63c_s outputs the sliding correlation values AC_s(k_s, 2z_s−1) calculated according to Equation (50) to the sth reception phase shifter 62c_s. In Equation (50), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}50\right]& \\ {\mathrm{AC}}_s\left(k_s,2z_s-1\right)=\sum _{n=1}^LX_s\left(2N_v\left(z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)A_n^{*}& \left(50\right)\end{array}$

Furthermore, the correlation value calculator 63c_s calculates sliding correlation values AC_s(k_s, 2z_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth transmission cycle T_r (each transmission cycle T_r; m_s: even number) according to Equation (51). The correlation value calculator 63c_s outputs the sliding correlation values AC_s(k_s, 2z_s) calculated according to Equation (51) to the sth reception phase shifter 62c_s. In Equation (51), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}51\right]& \\ {\mathrm{AC}}_s\left(k_s,2z_s\right)=\sum _{n=1}^LX_s\left(N_v\left(2z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)B_n^{*}& \left(51\right)\end{array}$

Now, how the sth reception phase shifter 62c_s of the specific sector radar SRc_s (s=1) will be described. The sth reception phase shifter 62c_s receives the sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s), that is, the sliding correlation values AC_s(k_s, m_s), that are output from the correlation value calculator 63c_s. The sth reception phase shifter 62c_s gives a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifter 25_s to the received sliding correlation values AC_s(k_s, m_s) every two transmission cycles on the basis of a transmission timing signal that is supplied from the pulse transmission controller 21_s in the m_sth transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21_s in the m_sth transmission cycle T_r, the sth reception phase shifter 62c_s gives a reception phase shift exp(j·floor[(m_s−1)/2](−φ_s)) corresponding to the ordinal number of the transmission cycle T_r to the sliding correlation values AC_s(k_s, m_s) every two transmission cycles (see Equation (52)). The sth reception phase shifter 62c_s outputs reception-phase-shift-added sliding correlation values ACP_s(k_s, m_s) to the coherent integrator 64c_s.

$\begin{array}{cc}\left[\mathrm{Formula}52\right]& \\ {\mathrm{ACP}}_1\left(k_1,m_1\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_1-1}{2}\right]{\phi }_1\right){\mathrm{AC}}_1\left(k_1,m_1\right)& \left(52\right)\end{array}$

The sth reception phase shifter 62c_s of the sector radar SRc_s (s=2) operates differently from that of the sector radar SRc_s (s=1) in that parameter φ₂ representing the phase rotation amount is different from φ₁ (see Equation (53)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

$\begin{array}{cc}\left[\mathrm{Formula}53\right]& \\ {\mathrm{ACP}}_2\left(k_2,m_2\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_2-1}{2}\right]{\phi }_2\right){\mathrm{AC}}_2\left(k_2,m_2\right)& \left(53\right)\end{array}$

The coherent integrator 64c_s receives the sliding correlation values ACP_s(k_s, m_s) that are output from the sth reception phase shifter 62c_s. The coherent integrator 64c_s adds together, for each discrete time k_s, sliding correlation values ACP_s(k_s, m_s) in a prescribed number (2NP) of transmission cycles T_r (a period 2NP×T_r) on the basis of sets of sliding correlation values ACP_s(k_s, m_s) that have been calculated in the m_sth transmission cycle T_r for the respective discrete times k_s.

The coherent integrator 64c_s calculates a v_sth coherent integration value ACC_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values ACP_s(k_s, m_s) in the prescribed number (2NP) or more of transmission cycles T_r (period 2NP×T_r) according to Equation (54). Parameter 2NP represents the number of times of coherent integration performed in the coherent integrator 64c_s. The coherent integrator 64c_s outputs the calculated coherent integration values ACC_s(k_s, v_s) to the distance estimator 65_s.

$\begin{array}{cc}\left[\mathrm{Formula}54\right]& \\ {\mathrm{ACC}}_s\left(k_s,v_s\right)=\sum _{m=1}^{2\mathrm{NP}}{\mathrm{ACP}}_s\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(54\right)\end{array}$

By setting the prescribed number 2NP at an integer multiple of 2π/φ_s in Equation (54), the coherent integrator 64c_s can reduce influences of circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number 2NP at an integer multiple of 2π/φ_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing 2NP times of coherent integration.

As such, the radar apparatus 10 according to the modification of the second embodiment can provide the same advantages as the radar apparatus 10 according to the first embodiment.

Embodiment 3

Each of the above embodiments are directed to the case of suppressing interference between radar transmission signals that are transmitted in an asynchronous manner between the two sector radars SR₁ and SR₂ which are installed being opposed to each other. A third embodiment is directed to a case of suppressing interference between radar transmission signals that are transmitted in an asynchronous manner between N_R sector radars (N_R (natural number)≧3).

In this embodiment, parameter s takes values 1 to N_R and each sector radar is configured in the same manner as the sector radar SRb_s according to the second embodiment or the sector radar SRc_s according to the modification of the second embodiment. In this embodiment, only differences from, for example, the sector radar STb_s according to the second embodiment will be described.

Although this embodiment is directed to a case of using a complementary code as in the second embodiment, the concept of this embodiment is likewise applicable to a case of using the same kind of transmission code as used in the first embodiment. In this case, each sector radar is configured in the same manner as the sector radar Sly according to the first embodiment or the sector radar SRa_s according to the modification of the first embodiment.

(Radar Transmitter)

The sth transmission phase shifter 25b_s of the sector radar SRb_s receives a transmission signal G_s(t_s) that is output from the modulater 23b_s or the LPF 24_s. The sth transmission phase shifter 25b_s gives a prescribed transmission phase shift to the received transmission signal G_s(t_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every transmission cycle T_r (see FIG. 14).

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21b_s in an m_sth transmission cycles T_r, the sth transmission phase shifter 25b_s gives a transmission phase shift exp(j·floor[(m_s−1)/2]φ_s) corresponding to the ordinal number m_s of the transmission cycles T_r to the transmission signal G_s(t_s) every two transmission cycles (see Equation (43)). Parameter φ_s represents a phase rotation amount (e.g., 90°) that is given in the sth transmission phase shifter 25b_s, and it is preferable that parameter φ_s satisfy the relationship of Inequality (9). The sth transmission phase shifter 25b_s outputs a transmission-phase-shift-added transmission signal GP_s(N_r(m_s−1)+t_s) to the D/A converter 26_s.

(Radar Receiver)

The sth reception phase shifter 62b_s receives a reception signal X_s(N_v(m_s−1)+k_s) that is output from the A/D converter 61_s. The sth reception phase shifter 62b_s gives a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifter 25b_s to the received reception signal X_s(N_v(m_s−1)+k_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21b_s every transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21b_s in an m_sth transmission cycle T_r, the reception phase shifter 62b_s gives a reception phase shift exp(−j·floor[(m_s−1)/2](−φ_s)) corresponding to the ordinal number m_s of the transmission cycle T_r to the reception signal X_s(N_v(m_s−1)+k_s) every two transmission cycles (see Equation (45)). The sth reception phase shifter 62b_s outputs a reception-phase-shift-added reception signal XP_s(N_v(m_s−1)+k_s) to the correlation value calculator 63b_s.

The coherent integrator 64b_s receives sliding correlation values AC_s(k_s, 2z_s−1) and AC_s(k_s, 2z_s) that are output from the correlation value calculator 63b_s. The coherent integrator 64b_s adds together sliding correlation values AC_s(k_s, 2z_s) and AC_s(k_s, 2z_s) in a prescribed number (2NP) of transmission cycles T_r (a period 2NP×T_r) on the basis of sets of sliding correlation values AC_s(k_s, 2z_s) and AC_s(k_s, 2z_s) that have been calculated in the two (odd-numbered and even-numbered) transmission cycles T_r for the respective discrete times k_s.

The coherent integrator 64b_s calculates a v_sth coherent integration value ACC_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values AC(k_s, 2z_s−1) and AC_s(k_s, 2z_s) in the prescribed number 2NP of periods (period 2NP×T_r) according to Equation (49). Parameter 2NP represents the number of times of coherent integration performed in the coherent integrator 64b_s. The coherent integrator 64b_s outputs the calculated coherent integration values ACC_s(k_s, v_s) to the distance estimator 65_s.

By setting the prescribed number 2NP at an integer multiple of 2π/φ_s in Equation (49), the coherent integrator 64b_s can reduce influences of circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number 2NP at an integer multiple of 2π/φ_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing 2NP times of coherent integration.

The sth transmission phase shifters 25_s of the sector radars SRb_s (s=1 to N_R) give phase shifts φ_s=φ(q_s, N_R)+α(=((2q_s−1)m/N_R)+α) that are different from each other in phase rotation direction. With this measure, the sector radars SRb_s can suppress an interference wave signal coming from the other sector radar in similar manners, and can prevent increase of range sidelobes and suppress degradation of the target ranging performance effectively without incorporating circuit error correction circuits even in the case where circuit errors such as a DC offset and IQ imbalance occur.

Parameter q_s (=s−1) takes values 0 to N_R−1, and parameter a is a fixed phase value. By performing coherent integration every 2N_R transmission cycles, the sth coherent integrator 64_s can effectively suppress interference between a radar transmission signal of the sector radar it belongs to and a radar transmission signal coming from another sector radar.

For example, where N_R=3 and α=0, phase shifts (φ₁, φ₂, φ₃)=(φ(−1, 3), φ(1, 3), φ(2, 3)) are set at (π/3, −π/3, π). Performing coherent integration every 2N_R transmission cycles, the sth coherent integrator 64b_s can effectively suppress interference between a radar transmission signal of the sector radar it belongs to and a radar transmission signal coming from another sector radar.

How the interference suppression effect is obtained will be described for a general case of 2N_R transmission cycles. Assume an example case that the sector radar SR₁ receives a radar transmission signal of a zth sector radar as an interference wave signal. Parameter z takes values 2 to N_R.

The output of the A/D converter 61_s (s=1) is given by Equation (55) in the case where a reception signal of an m₁th transmission cycle T_r of the sector radar SR₁ and a radar transmission signal (interference wave signal) coming from the zth sector radar are involved. Parameters m_z and N_delay(z) are given by Equations (56) and (57), respectively.

Furthermore, the output of the A/D converter 61_s of the sector radar SR₁ is given by Equation (58) in the case where a reception signal of the sector radar SR₁ in each of an (m₁+1)th to (m₁+(2N_R−1))th transmission cycles T_r and a radar transmission signal (interference wave signal) coming from the zth sector radar SR_z are involved if the propagation environment remains the same as in the m₁th transmission cycle T_r. In Equation (58), parameter w takes values 1 to (2N_R−1).

$\begin{array}{cc}\left[\mathrm{Formula}55\right]& \\ X_1\left(N_v\left(m_1-1\right)+k_1\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\sum _{z=1}^{N_R}h_{1z}{\mathrm{GP}}_z\left(N_{\mathrm{TR}}\left\{\lfloor {\Delta }_1\left\{N_v\left(m_1-1\right)+k_1\right\}/{\Delta }_z\rfloor -\lfloor {\tau }_{1z}/{\Delta }_z\rfloor \right\}\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{1z}{\mathrm{GP}}_z\left(N_{\mathrm{TR}}\left\{N_v\left(m_z-1\right)+N_{\mathrm{delay}\left(z\right)}\right\}\right)& \left(55\right)\\ \left[\mathrm{Formula}56\right]& \\ m_z=\lfloor \left\{\lfloor {\Delta }_1\left\{N_v\left(m_1-1\right)+k_1\right\}/{\Delta }_z\rfloor -\lfloor {\tau }_{1z}/{\Delta }_z\rfloor \right\}/N_v\rfloor -1& \left(56\right)\\ \left[\mathrm{Formula}57\right]& \\ N_{\mathrm{delay}\left(z\right)}=\mathrm{mod}\left\{\lfloor {\Delta }_1\left\{N_v\left(m_1-1\right)+k_1\right\}/{\Delta }_z\rfloor -\lfloor {\tau }_{1z}/{\Delta }_z\rfloor \right\},N_v\}& \left(57\right)\\ \left[\mathrm{Formula}58\right]& \\ X_1\left(N_v\left(m_1+w-1\right)+k_1\right)=h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\sum _{z=1}^{N_R}h_{1z}{\mathrm{GP}}_z\left(N_{\mathrm{TR}}\left\{N_v\left(m_z+w-1\right)+N_{\mathrm{delay}\left(z\right)}\right\}\right)]& \left(58\right)\end{array}$

The addition value of outputs, that is, sliding correlation values, of the correlation value calculator 63b_s of the sector radar SR₁ in the m₁th to (m₁+(2N_R−1))th transmission cycles is given by Equation (59). In Equation (59), the code sequence C_n is one of the complementary code sequences A_n and B_n.

$\begin{array}{cc}\left[\mathrm{Formula}59\right]& \\ {\mathrm{AC}}_1\left(k_1,m_1\right)+\sum _{w=1}^{2N_R-1}{\mathrm{AC}}_1\left(k_1,m_1+w-1\right)=\sum _{n=1}^L{\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}+\sum _{w=1}^{2N_R-1}\sum _{n=1}^L{\mathrm{XP}}_1\left(N_v\left(m_1+w-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right)C_n^{*}& \left(59\right)\end{array}$

The outputs of the sth reception phase shifter 62_s of the sector radar SR₁ in the m₁th transmission cycle T_r and the (m₁+w)th transmission cycle T_r are given by Equations (60) and (61), respectively:

$\begin{array}{cc}\left[\mathrm{Formula}60\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)X_1\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\sum _{z=1}^{N_R}h_{1z}{\mathrm{GP}}_z\left(N_{\mathrm{TR}}\left\{N_v\left(m_z-1\right)+N_{\mathrm{delay}\left(z\right)}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\sum _{z=1}^{N_R}h_{1z}\exp \left(j\left[\left(m_z-1\right){\phi }_z-\left(m_1-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_z-1\right)+N_{\mathrm{delay}\left(z\right)}\right)\right]& \left(60\right)\\ \left[\mathrm{Formula}61\right]& \\ {\mathrm{XP}}_1\left(N_v\left(m_1+w-1\right)+k_1\right)=\exp \left(-j\left(m_1+w-1\right){\phi }_1\right)X_1\left(N_v\left(m_1+w-1\right)+k_1\right)=\exp \left(-j\left(m_1+w-1\right){\phi }_1\right)\hspace{1em}\left[h_{11}{\mathrm{GP}}_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+h_{12}{\mathrm{GP}}_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}\right\}\right)\right]=h_{11}G_1\left(N_{\mathrm{TR}}\left\{N_v\left(m_1+w-1\right)+k_1-\lfloor {\tau }_{11}/{\Delta }_1\rfloor \right\}\right)+\sum _{z=1}^{N_R}h_{1z}\exp (j\left[\left(\left(m_z+w-1\right){\phi }_z-\left(m_1+w-1\right){\phi }_1\right]\right)G_2\left(N_{\mathrm{TR}}\left\{N_v\left(m_z+w-1\right)+N_{\mathrm{delay}\left(z\right)}\right)\right]& \left(61\right)\end{array}$

The first term of each of Equations (60) and (61) represents a desired signal component that is transmitted from the radar transmitter TXb_s of the sector radar SR₁ as a radar transmission signal, reflected by a target TAR_s, and received by the radar receiver RXb_s of the sector radar SR₁. Therefore, the first terms of the respective Equations (60) and (61) are in phase (see Equation (62)) and hence can provide a coherent integration gain when subjected to the coherent integration according to Equation (59). Symbol ∠[x] is an operator of outputting the phase component of a complex number x.

[Formula 62]

∠[h₁₁G(N_TR{N_v(m₁−1)+k₁−└τ₁₁/Δ₁┘})]=∠[h₁₁G(N_TR{N_v(m₁+w−1)+k₁−└τ₁₁/Δ₁┘})]  62)

On the other hand, the second term of each of Equations (60) and (61) represents an interference wave signal component that is transmitted from the radar transmitter of the zth sector radar as a radar transmission signal, reflected by the target, and received by the radar receiver RXTb_s of the sector radar SR₁.

If the carrier frequency error between the sector radar SR₁ and the zth sector radar is within an allowable range, that is, if Inequalities (63) hold, the interference wave signal components in the m₁th to (m₁+w)th transmission cycles have a phase relationship indicated by Equation (64). Equation (65) represents a result of coherent integration of the interference wave signal components by the coherent integrator 64b_s. Therefore, in the radar apparatus 10, the interference signal components have such a relationship that their signal components are canceled out each other by the coherent integration according to Equation (59) and hence the interference wave signal components can be suppressed effectively as is understood from Equation (65). However, the radar apparatus 10 becomes more prone to be affected by phase variations due to the frequency error f_dev as N_R increases. Therefore, N_R has an upper limit that depends on the frequency accuracy of the reference clock signals used in the radar apparatus 10.

$\begin{array}{cc}\left[\mathrm{Formulae}63\right]& \\ N_v{\Delta }_1\cong N_v{\Delta }_z,2\pi f_{\mathrm{dev}\left(z\right)}{\mathrm{wT}}_r<<0.1\pi & \left(63\right)\\ \left[\mathrm{Formula}64\right]& \\ \angle h_{1z}\exp (j\left[\left(\left(m_z+w-1\right){\phi }_z-\left(m_1+w-1\right){\phi }_1\right]\right)G\left(N_{\mathrm{TR}}\left\{N_v\left(m_z+w-1\right)+N_{\mathrm{delay}\left(z\right)}\right)\right]-\angle h_{1z}\exp \left(j\left[\left(m_z-1\right){\phi }_z-\left(m_1-1\right){\phi }_1\right]\right)G\left(N_{\mathrm{TR}}\left\{N_v\left(m_z-1\right)+N_{\mathrm{delay}\left(z\right)}\right\}\right)=w\left\{{\phi }_z-{\phi }_1\right\}+2\pi f_{\mathrm{dev}}\mathrm{wTr}=\frac{2\pi }{N_R}\mathrm{zw}+2\pi f_{\mathrm{dev}\left(z\right)}\mathrm{wTr}\cong \frac{2\pi }{N_R}\mathrm{zw}& \left(64\right)\\ \left[\mathrm{Formula}65\right]& \\ {\left\{{\mathrm{AC}}_1\left(k_1,m_1\right)+\sum _{w=1}^{N_{R-1}}{\mathrm{AC}}_1\left(k_1,m_1+w-1\right)\right\}}_{\mathrm{INTERFERENCE}\left(z\right)}=\sum _{z=1}^{N_R}{{\mathrm{AC}}_1\left(k_1,m_1\right)}_{\mathrm{INTERFERENCE}\left(z\right)}\left\{1+\sum _{w=1}^{N_R-1}\exp \left(jw\left({\phi }_z-{\phi }_1\right)+2\pi f_{\mathrm{dev}}\mathrm{wTr}\right)\right\}=\sum _{z=1}^{N_R}{{\mathrm{AC}}_1\left(k_1,m_1\right)}_{\mathrm{INTERFERENCE}\left(z\right)}\left\{1+\sum _{w=1}^{N_R-1}\exp \left(j\frac{2\pi }{N_R}\mathrm{zw}+2\pi f_{\mathrm{dev}}\mathrm{wTr}\right)\right\}\cong \sum _{z=1}^{N_R}{{\mathrm{AC}}_1\left(k_1,m_1\right)}_{\mathrm{INTERFERENCE}\left(z\right)}\left\{1+\sum _{w=1}^{N_R-1}\exp \left(j\frac{2\pi }{N_R}\mathrm{zw}\right)\right\}=0& \left(65\right)\end{array}$

Although the above description assumes the case that an interference wave signal that originates from the zth sector radar arrives at the sector radar SR₁, the same discussion is likewise applicable to a case that an interference wave signal that originates from the sector radar SR₁ arrives at the zth sector radar.

Embodiment 4

Each of the above embodiments are directed to the case of suppressing interference between radar transmission signals that are transmitted in an asynchronous manner between plural sector radars that are installed being opposed to each other. A fourth embodiment is directed to a case of suppressing interference between radar transmission signals that are transmitted in an asynchronous manner between plural sector radars that are installed being opposed to each other and each of which has plural radar transmitters that transmit radar transmission signals in a synchronous manner and plural radar receivers. The plural sector radars constituting a radar apparatus 10 according to this embodiment installed being opposed to each other as shown in FIG. 2 or 16, for example. In the following description, it is assumed that two sector radars constitute the radar apparatus 10 and hence parameter s takes values 1 and 2.

FIG. 17 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRd_s (s=1, 2) of the radar apparatus 10 according to the fourth embodiment. FIG. 18 is a block diagram showing, in detail, the internal configuration of each sector radar SRd_s (s=1, 2) of the radar apparatus 10 according to the fourth embodiment. FIG. 19 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission codes of radar transmission signals of a first radar transmitter and a second radar transmitter of each sector radar SRd_s (s=1, 2) of the radar apparatus 10 according to the fourth embodiment.

First, how the individual units of the sector radar SRd_s are configured will be described in a simplified manner. In the following description, operations that are common to the plural radar transmitters or the plural radar receivers of the same sector radar SRd_s will be described in a generic manner using parameter y and different operations of the plural radar transmitters or the plural radar receivers will be described individually. Parameter y takes a value 1 or 2 and represents the ordinal number of each of the radar transmitters and each of the radar receivers of the same sector radar SRd_s.

The sector radar SRd_s shown in FIG. 17 is configured so as to include a reference signal oscillator Lo_s, a pulse transmission controller 2d1_s, a first radar transmitter Txd1_s, a second radar transmitter Txd2_s, a first radar receiver Rxd1_s, and a second radar receiver Rxd2_s. The first radar transmitter Txd1_s is configured so as to have a transmission signal generater 2d1_s and a RF transmitter 31_s which is connected to a transmission antenna Ant-Tx1_s. The transmission signal generater 2d1_s is configured so as to include a code generater 221_s, a modulater 231, and an sth transmission phase shifter 25d_s. The transmission antenna Ant-Tx1_s or a reception antenna Ant-Rx1_s may be configured using a transmission antenna element or a reception antenna element. The configuration of the second radar transmitter Txd2_s is the same as that of the first radar transmitter Txd1_s and hence a description therefor will be omitted.

The first radar transmitter Txd1_s, the second radar transmitter Txd2_s, the first radar receiver Rxd1_s, and the second radar receiver Rxd2_s are connected to the reference signal oscillator Lo_s and are supplied with a reference signal from the reference signal oscillator Lo_s, whereby pieces of processing performed by the first radar transmitter Txd1_s, the second radar transmitter Txd2_s, the first radar receiver Rxd1_s, and the second radar receiver Rxd2_s are synchronized with each other.

The first radar receiver Rxd1_s is configured so as to have a RF receiver 41_s, a VGA unit 51_s, and a signal processer 6d1_s. The signal processer 6d1_s is configured so as to include an sth reception phase shifter 62d1_s, a correlation value calculator 63d1_s, a coherent integrator 64d1_s, and a distance estimator 651_s. The configuration of the second radar receiver Rxd2_s is the same as that of the first radar receiver Rxd1_s and hence a description therefor will be omitted.

(Yth Radar Transmitter (y=1 or 2))

Next, how the individual units of the yth first radar transmitter Txd1 (y=1) of the sector radar SRd_s are configured will be described in detail with reference to FIG. 18.

The transmission signal generater 2d1_s is configured so as to include the code generater 221_s, the modulater 231_s, an LPF 241_s, the sth transmission phase shifter 25d1_s, and a D/A converter 261_s. Although in FIG. 18 the transmission signal generater 2d1_s includes the LPF 241_s, the LPF 241_s may be provided in the first radar transmitter Txd1_s independently of the transmission signal generater 2d1_s. The configuration and the manner of operation of the RF transmitter 31_s are the same as those of the RF transmitter 3_s used in each of the above embodiments, and hence descriptions therefor will be omitted.

Next, how the individual units of each radar transmitter operate will be described in detail for an example case that y is equal to 1 (first radar transmitter Txd1_s). However, the following description is likewise applicable to the other case that y is equal to 2 (second radar transmitter Txd2_s). In each of the following embodiments, operations that are common to the plural radar transmitters of the same sector radar SRd_s will be described in a generic manner using parameter y and different operations of the plural radar transmitters will be described individually.

The pulse transmission controller 21d_s generates a transmission timing signal for a radio-frequency radar transmission signal every transmission cycle T_r. The pulse transmission controller 21d_s outputs the generated transmission timing signal to the code generater and the sth transmission phase shifter of each of the first radar transmitter Txd1_s and the second radar transmitter Txd2_s and the sth reception phase shifter of each of the first radar receiver Rxd1_s and the second radar receiver Rxd2_s.

The transmission signal generater 2d1_s generates a transmission reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the transmission signal generater 2d1_s operate on the basis of the generated transmission reference clock signal. Let f_TxBB represent the transmission reference clock frequency; then the transmission cycle T_r is expressed as an integer N_r multiple of a discrete time interval 1/f_TxBB which is determined by the transmission reference clock frequency f_TxBB (see Equation (66)). The transmission reference clock frequency f_TxBB is a nominal value and, in actuality, includes a frequency error that varies depending on the radar transmitters Tx_s.

$\begin{array}{cc}\left[\mathrm{Formula}66\right]& \\ T_r=N_r\times \frac{1}{f_{T\times \mathrm{BB}}}& \left(66\right)\end{array}$

The transmission signal generater 2d1_s periodically generates a baseband transmission signal G^y_s(t_s) (see Equation (67)) by modulating a code sequence C(1)_n having a code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21d_s every transmission cycle T_r. Parameter n takes values 1 to L, and parameter L represents the code length of the code sequence C(1)_n. Parameter j is the imaginary number unit which satisfies j²=−1. Parameter t_s represents discrete time.

[Formula 67]

G _s ^y(t_s)=I_s^y(t_s)+jQ_s^y(t_s)  (67)

The transmission signal generater of the second radar transmitter Txd2_s periodically generates a baseband transmission signal G^y_s(t_s) (see Equation (67)) by modulating a code sequence C(2)_n having the code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21d_s every transmission cycle T_r. The code sequences C(1)_n and C(2)_n are different code sequences which are orthogonal or low in correlation.

As shown in FIG. 19, the transmission signal G^y_s(t_s) is such that, for example, each code element of the code sequence C(1)_n or C(2)_n is modulated using N_o samples of the transmission reference clock signal in the transmission interval T_w (s) of each transmission cycle T_r. Therefore, in each transmission interval T_w, modulation is done using N_w (=N_o×L) samples. In the silent interval (T_r−T_w) (s) of each transmission cycle T_r, modulation is done using N_u (=N_r−N_w) samples. Therefore, the transmission signal G^y_s(t_s) of Equation (6) is expressed as Equation (68):

[Formula 68]

G _s ^y(N_r(m_s−1)+t_s)=I_x^y(N_r(m_s−1)+t_s)+jQ_s^y(N_r(m_s−1)+t_s)  (68)

The code generater 221_s generates a transmission code of the code sequence C(1)_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21d_s every transmission cycle T_r. The code generater 221_s outputs the generated transmission code of the code sequence C(1)_n to the modulater 231_s. That is, the single code generater 221_s generates a single code sequence.

The code generater of the second radar transmitter Txd2_s generates a transmission code of the code sequence C(2)_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21d_s every transmission cycle T_r. The code generater outputs the generated transmission code of the code sequence C(2)_n to the modulater. That is, the single code generater generates a single code sequence.

For example, the elements of each of the code sequences C(1)_n and C(2)_n are formed using two values [−1, 1] or four values [1, −1, j, −j]. The transmission code is a code sequence of one of, for example, a Barker code sequence, an M-sequence code, and a Gold code sequence which provides a low range sidelobe characteristic.

The modulater 231_s receives the transmission code C(1)_n or C(2)_n that is output from the code generater 221_s. The modulater 231_s generates a baseband transmission signal G^y_s(t_s) of Equation (67) by pulse-modulating the received transmission code C(1)_n or C(2)_n.

In the baseband transmission signal G_s(t_s) of Equation (67), I^y_s(t) and Q^y_s(t_s) represent the in-phase component and the quadrate component of a modulation signal, respectively. The modulater 231_s outputs a transmission signal G^y_s(t_s), in a preset limited band or lower, of the generated transmission signal G^y_s(t_s) to the sth transmission phase shifter 25d1_s via the LPF 241_s. The LPF 241_s may be omitted in the transmission signal generater 2d1_s. This also applies to each of the following embodiments.

Now, how the sth transmission phase shifters of the specific sector radar SRd_s (s=1) operate will be described. The sth transmission phase shifters of the first radar transmitter Txd1_s and the second radar transmitter Txd1_s receive the transmission signals G^y_s(t_s) that are output from the modulators or the LPFs, respectively. The transmission phase shifters give a common, prescribed transmission phase shift to the received transmission signals G^y_s(t_s) every transmission cycle T_r on the basis of a transmission timing signal that is output from the pulse transmission controller 21d_s (see FIG. 19).

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21d_s in an m_sth transmission cycle T_r, the sth transmission phase shifters of the first radar transmitter Txd1_s and the second radar transmitter Txd1_s give the transmission signals G^y_s(t_s) a common transmission phase shift exp(j(m_s−1)φ_s) corresponding to the ordinal number of the transmission cycle T_r every transmission cycle (see Equation (69)). Parameter m_s is a natural number and represents the ordinal number of the transmission cycle T_r. Parameter φ_s represents a phase rotation amount (e.g., 90°) given by the sth transmission phase shifters, and it is preferable that parameter φ_s satisfy the relationship of Inequality (9). The sth transmission phase shifters output transmission-phase-shift-added transmission signals GP^y_s(N_r(m_s−1)+t_s) to the D/A converters 261_s respectively.

[Formula 69]

GP _t ^y(N_r(m₁−1)+t₁)=exp(j(m₁−1)φ₁)G_t^y(N_r(m₁−1)+t_z)  (69)

The manner of operation of the sth transmission phase shifters of the sector radar SRd_s (s=2) is different from that of the sth transmission phase shifters of the sector radar SRd_s (s=1) in that parameter φ₂ representing the phase rotation amount in the transmission phase shift exp(j(m₂−1)φ₂) in Equation (70) is different from parameter (N. For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

Furthermore, parameter φ₁ in the transmission phase shift given by the sth transmission phase shifters of the first radar transmitter Txd1_s and the second radar transmitter Txd2_s of the sector radar SRd_s (s=1) and parameter φ₂ in the transmission phase shift given by the sth transmission phase shifters of the first radar transmitter and the second radar transmitter of the sector radar SRd_s (s=2) are opposite in phase (φ₁=φ₂).

[Formula 70]

GP ₂ ^y(N_r(m₂−1)+t₂)=exp(j(m₂−1)φ₂)G₂^y(N_r(m₂−1)+t₂)  (70)

The D/A converter 261_s converts the digital transmission signal GP^y_s(N_r(m_s−1)+t_s) that is output from the sth transmission phase shifter 25d1_s into an analog transmission signal. The D/A converter 261_s outputs the analog transmission signal to the RF transmitter 31_s.

(Yth Radar Receiver (y=1 or 2))

Next, how the individual units of the yth first radar receiver Rxd1_s (y=1) of the sector radar SRd_s are configured will be described in detail with reference to FIG. 18.

The first radar receiver Rxd1_s is configured so as to include the RF receiver 41_s to which the reception antenna Ant-Rx1_s is connected, the VGA unit 51_s, and the signal processer 6d1_s. The configuration and the manner of operation of the RF receiver 41_s are the same as those of the RF receiver 4_s used in each of the above embodiments, and hence descriptions therefor will be omitted. The signal processer 6d1_s is configured so as to include an A/D converter 611_s, the sth reception phase shifter 62d1_s, the correlation value calculator 63d1_s, the coherent integrator 64d1_s, and the distance estimator 651_s. Each unit of the signal processer 6d1_s performs a calculation periodically with each transmission cycle T_r as a signal processing interval.

Next, how the individual units of each yth radar receiver operate will be described in detail for an example case that y is equal to 1 (first radar receiver Rxd1_s). However, the following description is likewise applicable to the other case that y is equal to 2 (second radar transmitter Rxd2_s).

The reception antenna Ant-Rx1_s receives a reflection wave signal that is a radar transmission signal transmitted from the first radar transmitter Txd1_s or the second radar transmitter Txd2_s and reflected by a target TAR_s and a radar transmission signal coming from the other sector radar which is installed so as to be opposed to the sector radar SR_s concerned. Each reception signal received by the reception antenna Ant-Rx1_s is input to the RF receiver 41_s.

Like the RF transmitter 31_s, the RF receiver 41_s generates a reception reference signal in the carrier frequency band by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number.

The VGA unit 51_s receives a baseband reception signal that is output from the RF receiver 41_s and includes an I signal and a Q signal, and adjusts the output level of the received baseband reception signal into an input range (dynamic range) of the A/D converter 611_s.

The VGA unit 51_s outputs the output-level-adjusted baseband reception signal including the I signal and the Q signal to the A/D converter 611_s. In the embodiment, to simplify the description, it is assumed that the gain of the VGA unit 51_s is adjusted in advance so that the output level of a reception signal falls within the input range (dynamic range) of the A/D converter 611_s.

Like the RF receiver 41_s, the signal processer 6d1_s generates a reception reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the signal processer 61_s operate on the basis of the generated reception reference clock signal.

Let f_RxBB represent the reception reference clock frequency; then the transmission cycle T_r is expressed as an integer N_v multiple of a discrete time interval 1/f_RxBB which is determined by the reception reference clock frequency f_RxBB (see Equation (71). It is assumed that the transmission reference clock frequency f_TxBB is equal to an integer N_TR multiple of the reception reference clock frequency f_RxBB (see Equation (72)).

$\begin{array}{cc}\left[\mathrm{Formula}71\right]& \\ T_r=N_v\times \frac{1}{f_{R\times \mathrm{BB}}}& \left(71\right)\\ \left[\mathrm{Formula}72\right]& \\ f_{T\times \mathrm{BB}}=N_{\mathrm{TR}}\times f_{R\times \mathrm{BB}}& \left(72\right)\end{array}$

The A/D converter 611, receives the reception signal that is output from the VGA unit 51_s and includes the I signal and the Q signal, and converts the analog data reception signal into digital data by sampling the received reception signal including the I signal and the Q signal every discrete time 1/f_RxBB on the basis of the reception reference clock frequency f_RxBB.

The A/D converter 611, outputs the digital data reception signal obtained through the conversion done every discrete time k, to the sth reception phase shifter 621, in the form of discrete sample values. A reception signal x_s(k_s) which is a converted, discrete sample value is expressed as a complex number (see Equation (73)) using an I signal Ir^y_s(k_s) and a Q signal Q^yr_s(k_s) which are discrete sample values at a discrete time k_s:

[Formula 73]

x _s ^y(k_s)=Ir_s^y(k_s)+jQr_s^y(k_s)  (73)

In an m_sth transmission cycle T_r, the reception signal x_s(k_s) of Equation (73) which is output from the A/D converter 611_s is expressed as a complex baseband signal X^y_s(N_v(m_s−1)+k_s) (see Equation (74)).

[Formula 74]

X _s ^y(N_v(m_s−1)+k_s)=Ir_s^y(N_v(m_s−1)+k_s)+jQr_j^y(N_v(m_s−1)+k_s)  (74)

Now, how the sth reception phase shifters of the specific sector radar SRd_s (s=1) operate will be described. The sth reception phase shifters of the first radar receiver Rx1_s and the second radar receiver Rx2_s receive the reception signals X^y_s(N_v(m_s−1)+k_s) that are output from the A/D converters, respectively. The sth reception phase shifters give a common reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifters to the received reception signal X^y_s(N_v(m_s−1)+k_s) every transmission cycle on the basis of a transmission timing signal that is output from the pulse transmission controller 21d_s every transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21d_s in an m_sth transmission cycle T_r, the sth reception phase shifters give a common reception phase shift exp(j(m_s−1)(−φ_s)) corresponding to the ordinal number of the transmission cycle T_r to the reception signals X^y_s(N_v(M_s−1)+k_s) every transmission cycle (see Equation (75)), respectively. Parameter φ_s represents the common phase rotation amount (e.g., φ₁=−90° given by the sth reception phase shifters, and it is preferable that parameter φ_s satisfy Inequality (9). The sth reception phase shifters output reception-phase-shift-added reception signals XP^y_s(N_v(m_s−1)+k_s) to the correlation value calculators, respectively.

[Formula 75]

XP _t ^y(N_v(m₁−1)+k₁)=exp(−j(m₁−1)φ₁)X_j^y(N_v(m₁−1)+k₁)  (75)

The sth reception phase shifters of the sector radar SRd_s (s=2) operate differently from those of the sector radar SRd_s (s=1) in that the phase shift φ₂ representing a phase rotation amount is different from φ₁ (see Equation (76)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

[Formula 76]

XP ₂ ^y(N_v(m₂−1)+k₂)=exp(−j(m₂−1)φ₂)X₂^y(N_v(m₂−1)+k₂)  (76)

The correlation value calculator 63d1_s receives the reception signal XP^y_s(N_v(m_s−1)+k_s) that is output from the sth reception phase shifter 62d1_s. Based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63d1_s periodically generates, for discrete times k_s, a transmission code of the code sequence C(y)_n having the code length L transmitted in the m_sth transmission cycle T_r.

The correlation value calculator 63d1_s calculates sliding correlation values AC^y_s(k_s, m_s) between the received reception signal XP^y_s(N_v(m_s−1)+k_s) and the transmission code C(y)_n. Each sliding correlation value AC^y_s(k_s, m_s) is calculated by performing a sliding correlation operation on the transmission code and the reception signal at each discrete time k_s in the m_sth transmission cycle T_r.

More specifically, the correlation value calculator 63d1_s calculates sliding correlation values AC^y_s(k_s, m_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth transmission cycle T_r (each transmission cycle T_r) according to Equation (77). The correlation value calculator 63d1_s outputs the sliding correlation values AC^y_s(k_s, m_s) calculated according to Equation (77) to the coherent integrator 64d1_s. In Equation (77), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}77\right]& \\ {\mathrm{AC}}_s^y\left(k_s,m_s\right)=\sum _{n=1}^L{\mathrm{XP}}_s^y\left(N_v\left(m_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right){C\left(y\right)}_n^{*}& \left(77\right)\end{array}$

Although in each of the embodiments including this embodiment the correlation value calculator 63d1, performs calculations at discrete times k_s=1 to (N_u−N_w)/N_TR, the measurement range (the range of discrete times k_s) may be narrowed further to, for example, k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR in accordance with the range of presence of a target TAR_s to be measured by the radar apparatus 10. With this measure, in the radar apparatus 10, the amount of calculation of the correlation value calculator 63d1_s can be reduced further. That is, in the radar apparatus 10, the power consumption can be reduced further as a result of reduction in the calculation amount of the signal processer 6d1_s.

In the radar apparatus 10, where the correlation value calculator 63d1_s calculates sliding correlation values AC^y_s(k_s, m_s) at discrete times k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR, measurement of a reflection wave signal in each transmission interval T_w of a radar transmission signal can be omitted.

In the radar apparatus 10, even if a radar transmission signal transmitted goes around to enter the first radar receiver Rx1_s or the second radar receiver Rx2_s directly, a measurement can be performed with its influence eliminated. With the above restriction of the measurement range (the range of discrete times k_s), the coherent integrator 64d1_s and the distance estimator 651_s also operate in the same restricted measurement range.

The coherent integrator 64d1_s receives the sliding correlation values AC^y_s(k_s, m_s) that are output from the correlation value calculator 63d1_s. The coherent integrator 64_s adds together sliding correlation values AC^y_s(k_s, m_s) in a prescribed number (NP) of transmission cycles T_r (a period NP×T_r) on the basis of sets of sliding correlation values AC^y_s(k_s, m_s) that have been calculated in the m_sth transmission cycle T_r for the respective discrete times k_s.

The coherent integrator 64d1_s calculates a v_sth coherent integration value ACC^y_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values AC_s(1_s, m_s) in the prescribed number (NP) of transmission cycles T_r (period NP×T_r) according to Equation (78). Parameter NP represents the number of times of coherent integration performed in the coherent integrator 64d1_s. The coherent integrator 64d1_s outputs the calculated coherent integration values ACC^y_s(k_s, v_s) to the distance estimator 651_s.

$\begin{array}{cc}\left[\mathrm{Formula}78\right]& \\ {\mathrm{ACC}}_s^y\left(k_s,v_s\right)=\sum _{m_s=1}^{\mathrm{NP}}{\mathrm{AC}}_s^y\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(78\right)\end{array}$

By setting the prescribed number NP at an integer multiple of 2π/φ_s in Equation (78), the coherent integrator 64d1_s can reduce influences of the circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number NP at an integer multiple of 2π/φ_s in the sector radar SRd_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing NP times of coherent integration.

The sth transmission phase shifters of the first radar receiver Rx1_s and the second radar receiver Rx2_s of each sector radar SRd_s give phase shifts φ_s=φ(q_s, N_i)+α (=((2q_s−1)π/N_i)+α) that are different from each other in phase rotation direction. With this measure, the sector radars SRd_s can suppress an interference wave signal coming from the other sector radar in similar manners, and can prevent increase of range sidelobes and suppress degradation of the target ranging performance effectively without incorporating circuit error correction circuits even in the case where circuit errors such as a DC offset and IQ imbalance occur.

Parameter q_s (=s−1) takes values 0 to N_i−1, and parameter a is a fixed phase value. By performing coherent integration every N_i transmission cycles, each sth coherent integrator can effectively suppress interference between a radar transmission signal of the sector radar it belongs to and a radar transmission signal coming from the other sector radar.

For example, where N_i=2, q_s=1, and α=0, phase shifts (φ₁, φ₂) are set at (π/2, −π/2). Performing coherent integration every N_i (two) transmission cycles, each sth coherent integrator 64_s can effectively suppress interference between a radar transmission signal of the sector radar it belongs to and a radar transmission signal coming from the other sector radar.

For example, where N_i=3, q_s=1, and α=0, phase shifts (φ₁, φ₂, φ₃)=(φ(−1, 3), φ(1, 3), φ(2, 3)) are set at (π/3, −π/3, π). Performing coherent integration every N_i (three) transmission cycles, each sth coherent integrator 64_s can effectively suppress interference between a radar transmission signal of the sector radar it belongs to and a radar transmission signal coming from the other sector radar.

In this embodiment, each of the coherent integrators 64d1_s of the first radar receiver Rx1_s and the second radar receiver Rx2_s of the sector radar SRd_s (s=1) performs coherent integration every two transmission cycles. With this measure, each of the coherent integrator 64d1_s of the first radar receiver Rx1_s and the second radar receiver Rx2_s of the sector radar SRd_s (s=1) can effectively suppress interference between a radar transmission signal of the sector radar SRd_s (s=1) and a radar transmission signal coming from the other sector radar SRd_s (s=2). How the interference suppression effect is obtained will be described. Assume an example case that the sector radar SRd_s (s=1) receives a radar transmission signal of the sector radar SRd_s (s=2) as an interference wave signal.

The output of the A/D converter 611_s (s=1) is given by Equation (79) in the case where a reception signal of an m₁th transmission cycle T_r of the sector radar SRd_s (s=1) and a radar transmission signal (interference wave signal) coming from the sector radar SRd_s (s=2) are involved.

$\begin{array}{cc}\left[\mathrm{Formula}79\right]& \\ X_1^y\left(N_v\left(m_1-1\right)+k_1\right)=\sum _{y=1}^2h_{11}^y{\mathrm{GP}}_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\sum _{y=1}^2h_{12}^y{\mathrm{GP}}_2^y\left(N_{\mathrm{TR}}\left\{\lfloor {\Delta }_1\left\{N_v\left(m_1-1\right)+k_1\right\}/{\Delta }_2\rfloor -\lfloor {\tau }_{12}^y/{\Delta }_2\rfloor \right\}\right)=\sum _{y=1}^2h_{11}^y{\mathrm{GP}}_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\sum _{y=1}^2h_{12}^y{\mathrm{GP}}_2^y\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}^y\right\}\right)& \left(79\right)\end{array}$

The first term of Equation (79) represents a desired signal component that is transmitted from the respective radar transmitters of the sector radar SRd_s (s=1) as radar transmission signals, reflected by a target TAR_s, and received by each radar receiver of the sector radar SRd_s (s=1). The second term of Equation (79) represents an interference wave signal component that is transmitted from the respective radar transmitters of the sector radar SRd_s (s=2) as radar transmission signals, reflected by the same target TAR_s, and received by each radar receiver of the sector radar SRd_s (s=1).

In Equation (79), parameter h₁₁^y represents an amplitude and phase attenuation coefficient of a case that a radar transmission signal transmitted from a yth radar transmitter of the sector radar SRd_s (s=1) is received by a yth radar receiver of the sector radar SRd_s (s=1). Parameter h₁₂^y represents an amplitude and phase attenuation coefficient of a case that a radar transmission signal transmitted from a yth radar transmitter of the sector radar SRd_s (s=2) is received by the yth radar receiver of the sector radar SRd_s (s=1). Parameters m₂ and N_delay are given by Equations (80) and (81), respectively:

[Formula 80]

m ₂=└{└Δ₁{N_c(m₁−1)+k₁}/Δ₂┘−└τ₁₂^y/Δ₂┘}/N_v┘−1  (80)

[Formula 81]

N _delay ^y=mod {└Δ₁{N_r(m₁−1)+k₁}/Δ₂┘−└τ₁₂^y/Δ₂┘},N_v}  (81)

Symbol “└x┘” is an operator of outputting the integer part of a real number x. Parameter τ₁₁^y is the delay time that is taken by a radar transmission signal transmitted from the sector radar SRd_s (s=1) to be reflected by a target TAR_s (s=1) and received by the sector radar SRd_s (s=1). It is assumed that parameter τ₁₁^y of the case of y=1 is in the same transmission cycle T_r as parameter τ₁₁^y of the case of y=2.

Parameter τ₁₂^y is the delay time that is taken by a radar transmission signal transmitted from the sector radar SRd_s (s=2) to be reflected by a target TAR_s (s=2) or travel directly and be received by the sector radar SRd_s (s=1). It is assumed that parameter τ₁₂^y of the case of y=1 is in the same transmission cycle T_r as parameter τ₁₂^y of the case of y=2.

To simplify the description, no filter response characteristics of the radar transmitters TX_s and the radar receivers Rx_s of each sector radar SRd_s are taken into consideration.

Furthermore, the output of the A/D converter 611_s of the sector radar SRd_s (s=1) is given by Equation (82) in the case where a reception signal of a yth radar receiver of the sector radar SRd_s (s=1) in an (m₁+1)th transmission cycle T_r and a radar transmission signal (interference wave signal) coming from the sector radar SRd_s (s=2) are involved if it is assumed that the propagation environment is the same as in the m₁th transmission cycle T_r. The phrase “the propagation environment is the same as in the m₁th transmission cycle T_r” means that the complex attenuation coefficients h₁₁^y and h₁₂^y and the delay times τ₁₁^y and τ₁₂^y can be regarded as remaining unchanged.

$\begin{array}{cc}\left[\mathrm{Formula}82\right]& \\ X_1^y\left(N_v\left(m_1+w-1\right)+k_1\right)=\sum _{y=1}^2h_{11}^y{\mathrm{GP}}_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1+w-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\sum _{y=1}^2h_{12}^y{\mathrm{GP}}_2^y\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}^y\right\}\right)& \left(82\right)\end{array}$

The addition value of outputs, that is, sliding correlation values, of the correlation value calculator of a yth radar receiver of the sector radar SRd_s (s=1) in N_i transmission cycles, that is, an m₁th transmission cycle and an (m₁+(N_i−1))th transmission cycle, is given by Equation (83). In Equation (83), the code sequence C_n is one of code sequences A_n and B_n.

$\begin{array}{cc}\left[\mathrm{Formula}83\right]& \\ {\mathrm{AC}}_1^y\left(k_1,m_1\right)+\sum _{w=1}^{\mathrm{Ni}-1}{\mathrm{AC}}_1^y\left(k_1,m_1+w-1\right)=\sum _{n=1}^L{\mathrm{XP}}_1^y\left(N_v\left(m_1-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right){C\left(y\right)}_n^{*}+\sum _{w=1}^{\mathrm{Ni}-1}\sum _{n=1}^L{\mathrm{XP}}_1^y\left(N_v\left(m_1+w-1\right)+\frac{N_o}{N_{\mathrm{TR}}}\left(n-1\right)+k_1\right){C\left(y\right)}_n^{*}& \left(83\right)\end{array}$

The outputs of each sth reception phase shifter of the sector radar SRd_s (s=1) in the m₁th transmission cycle T_r and the (m₁+w)th transmission cycle T_r are given by Equations (84) and (85), respectively:

$\begin{array}{cc}\left[\mathrm{Formula}84\right]& \\ \begin{array}{c}{\mathrm{XP}}_1^y\left(\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1\end{array}\right)=\exp \left(-j\left(m_1-1\right){\phi }_1\right)X_1^y\left(N_v\left(m_1-1\right)+k_1\right)\\ =\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[\begin{array}{c}\sum _{y=1}^2h_{11}^y{\mathrm{GP}}_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{y=1}^2h_{12}^y{\mathrm{GP}}_2^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_2-1\right)+\\ N_{\mathrm{delay}}^y\end{array}\right\}\right)\end{array}\right]\\ =\exp \left(-j\left(m_1-1\right){\phi }_1\right)\left[\begin{array}{c}\sum _{y=1}^2h_{11}^y\exp \left(j\left(m_1-1\right){\phi }_1\right)G_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{y=1}^2h_{12}^y\exp \left(j\left(m_2-1\right){\phi }_2\right)G_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_2-1\right)+\\ N_{\mathrm{delay}}^y\end{array}\right\}\right)\end{array}\right]+\\ \sum _{y=1}^2h_{11}^yG_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(m_1-1\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{d=1}^2h_{12}^y\exp \left(j\left[\begin{array}{c}\left(m_2-1\right){\phi }_2-\\ \left(m_1-1\right){\phi }_1\end{array}\right]\right)G_1^y\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}^y\right)\right]\end{array}& \left(84\right)\\ \left[\mathrm{Formula}85\right]& \\ \begin{array}{c}{\mathrm{XP}}_1^y\left(\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right)+\\ k_1\end{array}\right)=\exp \left(-j\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right){\phi }_1\right)X_1^y\left(N_v\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right)+k_1\right)\\ =\exp \left(-j\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right){\phi }_1\right)\left[\begin{array}{c}\sum _{y=1}^2h_{11}^y{\mathrm{GP}}_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{y=1}^2h_{12}^y{\mathrm{GP}}_2^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_2+\\ w-1\end{array}\right)+\\ N_{\mathrm{delay}}^y\end{array}\right\}\right)\end{array}\right]\\ =\exp \left(-j\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right){\phi }_1\right)\left[\begin{array}{c}\sum _{y=1}^2h_{11}^y\exp \left(j\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right){\phi }_1\right)G_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{y=1}^2h_{12}^y\exp \left(j\left(\begin{array}{c}{m}_2+\\ w-1\end{array}\right){\phi }_2\right)G_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_2+\\ w-1\end{array}\right)+\\ N_{\mathrm{delay}}^y\end{array}\right\}\right)\end{array}\right]+\\ \sum _{y=1}^2h_{11}^yG_1^y\left(N_{\mathrm{TR}}\left\{\begin{array}{c}{N}_v\left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right)+\\ k_1-\lfloor {\tau }_{11}^y/{\Delta }_1\rfloor \end{array}\right\}\right)+\\ \sum _{y=1}^2h_{12}^y\exp \left(j\left[\begin{array}{c}\left(\begin{array}{c}{m}_2+\\ w-1\end{array}\right){\phi }_2-\\ \left(\begin{array}{c}{m}_1+\\ w-1\end{array}\right){\phi }_1\end{array}\right]\right)G_1^y\left(N_{\mathrm{TR}}\left\{N_v\left(\begin{array}{c}{m}_2+\\ w-1\end{array}\right)+N_{\mathrm{delay}}^y\right)\right]\end{array}& \left(85\right)\end{array}$

The first term of each of Equations (84) and (85) represents a desired signal component that is transmitted from the respective radar transmitters TXd1_s of the sector radar SRd_s (s=1) as radar transmission signals, reflected by a target TAR_s, and received by the radar receiver RXd1_s. Therefore, the first terms of the respective Equations (84) and (85) are in phase (see Equation (86)) and hence can provide a coherent integration gain when subjected to the coherent integration according to Equation (83). Symbol ∠[x] is an operator of outputting the phase component of a complex number x.

[Formula 86]

∠[h₁₁^yG₁^y(N_TR{N_v(m₁−1)+k₁−└τ₁₁^y/Δ₁┘})]=∠[h₁₁^yG₁^y(N_TR{N_v(m₁+w−1)+k₁−└τ₁₁^y/Δ₁┘})]  (86)

On the other hand, the second term of each of Equations (84) and (85) represents an interference wave signal component that is transmitted from the radar transmitters of the sector radar SRd_s (s=2) as radar transmission signals, reflected by the target TAR_s, and received by each radar receiver RXd1_s of the sector radar SRd_s (s=1).

If the carrier frequency error between the sector radar SRd_s (s=1) and the sector radar SRd_s (s=2) is within an allowable range, that is, if Inequalities (63) hold, interference wave signal components in the m₁th transmission cycle and the (m₁+w)th transmission cycle are in a phase relationship indicated by Equation (87).

$\begin{array}{cc}\left[\mathrm{Formula}87\right]& \\ \angle h_{12}^y\exp (j\left[\left(m_2+w-1\right){\phi }_2-\left(\left(m_1+w-1\right){\phi }_1\right]\right)G_1^y\left(N_{\mathrm{TR}}\left\{N_v\left(m_2+w-1\right)+N_{\mathrm{delay}}^y\right)\right]-\angle h_{12}^y\exp \left(j\left[\left(m_2-1\right){\phi }_2-\left(m_1-1\right){\phi }_1\right]\right)G_1^y\left(N_{\mathrm{TR}}\left\{N_v\left(m_2-1\right)+N_{\mathrm{delay}}\right)\right]=w\left\{{\phi }_2-{\phi }_1\right\}+2\pi f_{\mathrm{dev}}\mathrm{wTr}=-2\frac{\pi q}{N_p}w+2\pi f_{\mathrm{dev}}\mathrm{wTr}& \left(87\right)\end{array}$

In Equation (87), parameter f_dev represents the carrier frequency error between the sector radar SRd_s (s=1) and the sector radar SRd_s (s=2) which is defined by a carrier frequency error due to a frequency error of the transmission reference clock signal and a sampling frequency error due to a frequency error of the reception reference clock signal.

If the carrier frequency error between the sector radars SRd_i and the sector radar SRd₂ is within an allowable range, that is, if Inequalities (63) hold, interference wave signal components in the m₁th transmission cycle to the (m₁+w)th transmission cycle have a phase relationship indicated by Equation (87). Equation (88) represents a result of coherent integration performed on interference signal components by each coherent integrator 64d_s. Therefore, in the radar apparatus 10, the interference components have such a phase relationship as to be canceled out each other as is understood from Equation (88) and hence the interference wave signal components can be suppressed effectively. However, the radar apparatus 10 becomes more prone to be affected by phase variations due to the frequency error f_dev as N_i increases. Therefore, N_i has an upper limit that depends on the frequency accuracy of the reference clock signals used in the radar apparatus 10.

$\begin{array}{cc}\left[\mathrm{Formula}88\right]& \\ {\left\{{\mathrm{AC}}_1^y\left(k_1,m_1\right)+\sum _{w=1}^{\mathrm{Ni}-1}{\mathrm{AC}}_1^y\left(k_1,m_1+w-1\right)\right\}}_{\mathrm{INTERFERENCE}\left(y\right)}={\left\{{\mathrm{AC}}_1^y\left(k_1,m_1\right)\right\}}_{\mathrm{INTERFERENCE}\left(y\right)}\left\{1+\sum _{w=1}^{\mathrm{Ni}-1}\exp \left(jw\left({\phi }_2-{\phi }_1\right)+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r\right)\right\}={\left\{{\mathrm{AC}}_1^y\left(k_1,m_1\right)\right\}}_{\mathrm{INTERFERENCE}\left(y\right)}\left\{1+\sum _{w=1}^{\mathrm{Ni}-1}\exp \left[-j\left(\frac{2\pi q}{N_i}w+2\pi f_{\mathrm{dev}}{\mathrm{wT}}_r\right)\right]\right\}\cong 0& \left(88\right)\end{array}$

Although the above description assumes a case that an interference wave signal that originates from the sector radar SRd_s (s=2) arrives at the sector radar SRd_s (s=1), the same discussion is likewise applicable to a case that an interference wave signal that originates from the sector radar SRd_s (s=1) arrives at the sector radar SRd_s (s=2).

The distance estimator 651_s receives coherent integration values ACC_s^y (k_s, v_s) at respective discrete times k_s that are output from the coherent integrator 641_s every NP transmission cycles T_r. The distance estimator 651_s estimates a distance to the target TAR on the basis of the received coherent integration values ACC^y_s(k_s, v) at the respective discrete times k_s. For example, the estimation method disclosed in the above-mentioned Referential non-patent document 3 can be applied to the distance estimation performed in the distance estimator 651_s.

The square of the absolute value of each of coherent integration values that are obtained in the v_sth output cycle (v_s×NP×T_r) and supplied from the coherent integrator 641_s, |ACC^y_s(k_s, v_s)|², corresponds to a reception level of a reflection wave signal at each discrete time k_s. The distance estimator 651_s estimates a distance Range(k_ps) according to Equation (31) on the basis of a detection time k_ps of a peak reception level that is higher than an environment noise level of the sector radar SRd_s by a prescribed value or more. In Equation (31), parameter C₀ represents the speed of light.

Operating in the above-described manner, in the case where plural sector radars are installed being opposed to each other, the radar apparatus 10 according to the fourth embodiment can suppress interference between the sector radars with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other. Furthermore, the radar apparatus 10 can prevent increase of range sidelobes and suppress degradation of the target ranging performance effectively without incorporating circuit error correction circuits even in the case where circuit errors such as a DC offset and IQ imbalance occur.

Embodiment 5

The radar apparatus 10 according to the fifth embodiment is different from that according to the fourth embodiment in that a complementary code is used as the transmission code.

How each of sector radars SRe_s (s=1, 2) constituting the radar apparatus 10 according to the fifth embodiment is configured and operates will be described with reference to FIGS. 20-22. FIG. 20 is a block diagram showing, in a simplified manner, the internal configuration of each sector radar SRe_s (s=1, 2) of the radar apparatus 10 according to the fifth embodiment. FIG. 21 is a block diagram showing, in detail, the internal configuration of each sector radar SRe_s (s=1, 2) of the radar apparatus 10 according to the fifth embodiment. FIG. 22 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission codes of radar transmission signals of a first radar transmitter and a second radar transmitter of each sector radar SRe_s (s=1, 2) of the radar apparatus 10 according to the fifth embodiment.

Units (blocks) of the sector radar SRe_s having the same (in configuration and operation) units in the sector radar SRd_s will be given the same reference symbols as the latter, and their configurations and operations will not be described (only differences will be described).

As shown in FIG. 20, the radar apparatus SRe_s is configured so as to include the reference signal oscillator Lo_s, a pulse transmission controller 21e_s, a first radar transmitter Txe1_s, a second radar transmitter Txe2_s, a first radar receiver Rxe1_s, and a second radar receiver Rxe2_s. The first radar transmitter Txe1_s is configured so as to have a transmission signal generater 2e l_s and the RF transmitter 31_s which is connected to the transmission antenna Ant-Tx1_s. The transmission signal generater 2e1_s is configured so as to include a code generater 22e1, a modulater 231, and an sth transmission phase shifter 25e1_s. The code generater 22e1_s includes, as at least one code generater, a first code generater 22e11_s and a second code generater 22e12_s and generates at least one code sequence.

The first radar transmitter Txe1_s, the second radar transmitter Txe2_s, the first radar receiver Rxe1_s, and the second radar receiver Rxe2_s are connected to the reference signal oscillator Lo_s and are supplied with a reference signal from the reference signal oscillator Lo_s, whereby pieces of processing performed by the first radar transmitter Txe1_s, the second radar transmitter Txe2_s, the first radar receiver Rxe1_s, and the second radar receiver Rxe2_s are synchronized with each other.

The first radar receiver Rxe1_s is configured so as to have the RF receiver 41_s, the VGA unit 51_s, and a signal processer 6e1_s. The signal processer 6e1_s is configured so as to include an sth reception phase shifter 62e1_s, a correlation value calculator 63e1_s, a coherent integrator 64e1_s, and the distance estimator 651_s. The configuration of the second radar receiver Rxe2_s is the same as that of the first radar receiver Rxe1_s and hence a description therefor will be omitted.

(Yth Radar Transmitter (y=1 or 2))

Next, how the individual units of the yth first radar transmitter Txe1_s (y=1) of the sector radar SRe_s are configured will be described in detail with reference to FIG. 21.

The transmission signal generater 2e1_s is configured so as to include the code generater 22e1_s, the modulater 231_s, the LPF 241_s, the sth transmission phase shifter 25e1_s, and the D/A converter 261_s. Although in FIG. 21 the transmission signal generater 2e1_s includes the LPF 241_s, the LPF 241_s may be provided in the first radar transmitter Txe1_s independently of the transmission signal generater 2e1_s. The configuration and the manner of operation of the RF transmitter 31_s are the same as those of the RF transmitter 3_s used in each of the above embodiments, and hence descriptions therefor will be omitted.

Next, how the individual units of each yth radar transmitter operate will be described in detail for an example case that y is equal to 1 (first radar transmitter Txe1_s). However, the following description is likewise applicable to the other case that y is equal to 2 (second radar transmitter Txe2_s).

The pulse transmission controller 21e, generates a transmission timing signal for a radio-frequency radar transmission signal every transmission cycle T_r. The pulse transmission controller 21e_s outputs the generated transmission timing signal to the code generater and the sth transmission phase shifter of each of the first radar transmitter Txe1_s and the second radar transmitter Txe2_s and the sth reception phase shifter of each of the first radar receiver Rxe1_s and the second radar receiver Rxe2_s.

The transmission signal generater 2e1_s generates a transmission reference clock signal by multiplying the reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the transmission signal generater 2e1_s operate on the basis of the generated transmission reference clock signal. Let f_TxBB represent the transmission reference clock frequency; then the transmission cycle T_r is expressed as an integer N_r multiple of a discrete time interval 1/f_TxBB which is determined by the transmission reference clock frequency f_TxBB (see Equation (66)). The transmission reference clock frequency f_TxBB is a nominal value and, in actuality, includes a frequency error that varies depending on the radar transmitter Tx_s.

The transmission signal generater 2e1_s periodically generates a baseband transmission signal G_s(t_s) (see Equation (67)) by modulating a complementary code sequence A_n having a code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21e_s every transmission cycle T_r. Parameter n takes values 1 to L, and parameter L represents the code length of the code sequence A_n. Parameter j is the imaginary number unit which satisfies j²=−1. Parameter t_s represents discrete time.

The transmission signal generater of the second radar transmitter Txe2_s periodically generates a baseband transmission signal G^y_s(t_s) (see Equation (67)) by modulating a complementary code sequence B_n having the code length L on the basis of a transmission timing signal (for a radar transmission signal) which is output from the pulse transmission controller 21e_s every transmission cycle T_r. Parameter n takes values 1 to L, and parameter L represents the code length of the code sequence B_n.

For example, as shown in FIG. 22, the transmission signal G^y_s(n_s) is such that each code element of the complementary code sequence A_n or B_n is modulated using N_o samples of the transmission reference clock signal in the transmission interval T_w (s) of each transmission cycle T_r. Therefore, in each transmission interval T_w, modulation is done using N_w (=N_o×L) samples. In the silent interval (T_r−T_w) (s) of each transmission cycle T_r, modulation is done using N_u (=N_r−N_w) samples. Therefore, the transmission signal G^y_s(n_s) of Equation (67) is expressed as Equation (68).

The first code generater 22e11_s generates a transmission code of the one complementary code sequence A_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21e_s every odd-numbered transmission cycle T_r. The first code generater 22e11_s outputs the generated transmission code of the complementary code sequence A_n to the modulater 231_s.

The second code generater 22e12_s generates a transmission code of the other complementary code sequence B_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21e_s every even-numbered transmission cycle T_r. The second code generater 22e12_s outputs the generated transmission code of the complementary code sequence B_n to the modulater 231_s.

The first code generater of the second radar transmitter Txe2_e generates a transmission code of the one complementary code sequence B_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21e_s every odd-numbered transmission cycle T_r. The first code generater outputs the generated transmission code of the complementary code sequence B_n to the modulater.

Furthermore, the second code generater of the second radar transmitter Txe2_e generates a transmission code of the other complementary code sequence A_n of the complementary code sequences A_n and B_n having the code length L on the basis of a transmission timing signal that is output from the pulse transmission controller 21e, every even-numbered transmission cycle T_r. The second code generater outputs the generated transmission code of the complementary code sequence A_n to the modulater.

The modulater 231_s receives the transmission code A_n or B_n that is output from the code generater 22e1_s. The modulater 231, generates a baseband transmission signal G^y_s(t_s) of Equation (67) by pulse-modulating the received transmission code A_n or B_n. The modulater 231_s outputs a transmission signal G^y_s(t_s), in a preset limited band or lower, of the generated transmission signal G^y_s(n_s) to the sth transmission phase shifter 25e1_s via the LPF 241_s.

Now, how the sth transmission phase shifters of the specific sector radar SRe_s (s=1) operate will be described. The sth transmission phase shifters of the first radar transmitter Txe1_s and the second radar receiver Rxe2_s receive the transmission signals G^y_s(t_s) that are output from the modulators or the LPFs, respectively. The sth transmission phase shifters give a common, prescribed transmission phase shift to the received transmission signal G^y_s(t_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21e_s every transmission cycle T_r (see FIG. 22).

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21e_s in an m_sth transmission cycles T_r, the sth transmission phase shifters of the first radar transmitter Txe1_s and the second radar receiver Rxe2_s give a common transmission phase shift exp(j·floor[(m_s−1)/2]φ_s) corresponding to the ordinal number of the transmission cycle T_r to the transmission signal G^y_s(t_s) every two transmission cycles (see Equation (89)). Parameter φ_s represents a phase rotation amount (e.g., 90°) that is given in the sth transmission phase shifters 25e1_s, and it is preferable that parameter φ_s satisfy the relationship of Inequality (9). The sth transmission phase shifters output transmission-phase-shift-added transmission signals GP^y_s(N_r(m_s−1)+t_s) to the D/A converters 261_s, respectively. Symbol floor[x] is an operator of outputting an integer obtained by rounding down a real number x.

$\begin{array}{cc}\left[\mathrm{Equation}89\right]& \\ {\mathrm{GP}}_1^y\left(N_r\left(m_1-1\right)+t_1\right)=\exp \left(j\mathrm{floor}\left[\frac{m_1-1}{2}\right]{\phi }_1\right)G_1^y\left(N_r\left(m_1-1\right)+t_1\right)& \left(89\right)\end{array}$

The manner of operation of the sth transmission phase shifters of the sector radar SRe_s (s=2) is different from that of the sth transmission phase shifters of the sector radar SRe_s (s=1) in that parameter φ₂ representing the phase rotation amount in the transmission phase shift exp(j·floor[(m_s−1)/2]φ_s) in Equation (90) is different from parameter φ₁ and has a value −90°, for example.

Furthermore, parameter φ₁ in the transmission phase shift given by the sth transmission phase shifters of the first radar transmitter Txe1_s and the second radar transmitter Txe2_s of the sector radar SRe_s (s=1) and parameter φ₂ in the transmission phase shift given by the sth transmission phase shifters of the first radar transmitter and the second radar transmitter of the sector radar SRe_s (s=2) are opposite in phase (φ₁=−φ₂).

$\begin{array}{cc}\left[\mathrm{Equation}\left(90\right)\right]& \\ {\mathrm{GP}}_2^y\left(N_r\left(m_2-1\right)+t_2\right)=\exp \left(j\mathrm{floor}\left[\frac{m_2-1}{2}\right]{\phi }_2\right)G_2^y\left(N_r\left(m_2-1\right)+t_2\right)& \left(90\right)\end{array}$

(Yth Radar Receiver (y=1 or 2))

Next, how the individual units of the yth first radar receiver Rxe_s (y=1) of the sector radar SRe_s are configured will be described in detail with reference to FIG. 21.

The radar receiver Rxe1_s is configured so as to include the RF receiver 41_s to which the reception antenna Ant-Rx1_s is connected, the VGA unit 51_s, and the signal processer 6e1_s. The configuration and the manner of operation of the RF receiver 41_s are the same as those of the RF receiver 4_s used in each of the above embodiments, and hence descriptions therefor will be omitted. The signal processer 6e1_s is configured so as to include the A/D converter 611_s, the sth reception phase shifter 62e1_s, the correlation value calculator 63e1, the coherent integrator 64e1, and the distance estimator 651_s. Each unit of the signal processer 6e1_s performs a calculation periodically with each transmission cycle T_r as a signal processing interval.

Next, how the individual units of each yth radar receiver operate will be described in detail for an example case that y is equal to 1 (first radar receiver Rxe1_s). However, the following description is likewise applicable to the other case that y is equal to 2 (second radar receiver Rxe2_s).

The reception antenna Ant-Rx1_s receives a reflection wave signal that is a radar transmission signal transmitted from the first radar transmitter Txe1_s or the second radar transmitter Txe2_s and reflected by a target TAR_s and a radar transmission signal coming from the other sector radar which is installed so as to be opposed to the sector radar SR_s concerned. Each reception signal received by the reception antenna Ant-Tx1_s is input to the RF receiver 41_s.

The VGA unit 51_s receives a baseband reception signal that is output from the RF receiver 41_s and includes an I signal and a Q signal, and adjusts the output level of the received baseband reception signal into an input range (dynamic range) of the A/D converter 611_s.

The VGA unit 51_s outputs the output-level-adjusted baseband reception signal including the I signal and the Q signal to the A/D converter 611_s. In the embodiment, to simplify the description, it is assumed that the gain of the VGA unit 51_s is adjusted in advance so that the output level of a reception signal falls within the input range (dynamic range) of the A/D converter 611_s.

Like the RF receiver 41_s, the signal processer 6e 1_s generates a reception reference clock signal by multiplying a reference signal generated by the reference signal oscillator Lo_s by a prescribed number. The individual units of the signal processer 6e1_s operate on the basis of the generated reception reference clock signal.

Now, how the sth reception phase shifters of the specific sector radar SRe_s (s=1) operate will be described. The sth reception phase shifters of the first radar receiver Rxe1_s and the second radar receiver Rxe2_s receive reception signals X^y_s(N_v(m_s−1)+k_s) that are output from the A/D converters, respectively. The sth reception phase shifting units give a reception phase shift that is opposite in direction to the phase shift component that was given by the sth transmission phase shifters to the received reception signals X^y_s(N_v(m_s−1)+k_s) every two transmission cycles on the basis of a transmission timing signal that is output from the pulse transmission controller 21e_s every transmission cycle T_r.

More specifically, based on a transmission timing signal that is supplied from the pulse transmission controller 21e_s in an m_sth transmission cycle T_r, the reception phase shifters of the first radar receiver Rxe1_s and the second radar receiver Rxe2_s give a common reception phase shift exp(−j·floor[(m_s−1)/2](−φ_s)) corresponding to the ordinal number of the transmission cycle T_r to the reception signals X^y_s(N_v(m_s−1)+k_s) every two transmission cycles (see Equation (91)), respectively. The sth reception phase shifters output reception-phase-shift-added reception signals XP^y_s(N_v(m_s−1)+k_s) to the correlation value calculator, respectively.

$\begin{array}{cc}\left[\mathrm{Formula}91\right]& \\ {\mathrm{XP}}_1^y\left(N_v\left(m_1-1\right)+k_1\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_1-1}{2}\right]{\phi }_1\right)X_1^y\left(N_v\left(m_1-1\right)+k_1\right)& \left(91\right)\end{array}$

The sth reception phase shifters of the sector radar SR_s (s=2) operate differently from those of the sector radar SR_s (s=2) in that parameter φ₂ representing a rotation amount is different from φ₁ (see Equation (92)). For example, parameters φ₁ and φ₂ are 90° and −90°, respectively.

$\begin{array}{cc}\left[\mathrm{Formula}92\right]& \\ {\mathrm{XP}}_2^y\left(N_v\left(M_2-1\right)+k_2\right)=\exp \left(-j\mathrm{floor}\left[\frac{m_2-1}{2}\right]{\phi }_2\right)X_2^y\left(N_v\left(m_2-1\right)+k_2\right)& \left(92\right)\end{array}$

The correlation value calculator 63e1_s receives the reception signal XP^Y_s(N_v(m_s−1)+k_s) that is output from the sth reception phase shifter 62e1_s. Based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63e1_s periodically generates, for discrete times k_s, a transmission code of the code sequence A_n having the code length L transmitted in an m_sth transmission cycle T_r (m_s (odd number)=2z_s−1 where z_s is a natural number). Furthermore, based on the reception reference clock signal obtained by multiplying the reference signal by the prescribed number, the correlation value calculator 63e1_s periodically generates, for discrete times k_s, a transmission code of the code sequence B_n having the code length L transmitted in an m_sth transmission cycle T_r (m_s (even number)=2z_s).

The correlation value calculator 63e1_s calculates sliding correlation values AC^y_s(k_s, m_s) between the received reception signal XP^y_s(N_v(m_s−1)+k_s) and the transmission code A_n or B_n. Each sliding correlation value AC^y_s(k_s, m_s) is calculated by performing a sliding correlation operation on the transmission code and the reception signal at each discrete time k_s in the m_sth transmission cycle T_r.

More specifically, the correlation value calculator 63e1_s calculates sliding correlation values AC^y_s(k_s, 2z_s−1) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth (m_s (odd number)=2z_s−1) transmission cycle T_r (each transmission cycle T_r) according to Equation (93). The correlation value calculator 63e1_s outputs the sliding correlation values AC^y_s(k_s, 2z_s−1) calculated according to Equation (93) to the coherent integrator 64e1_s. In Equation (93), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}93\right]& \\ {\mathrm{AC}}_s^y\left(k_s,2z_s-1\right)=\sum _{n=1}^L{\mathrm{XP}}_s^y\left(2N_v\left(z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)A_n^{*}& \left(93\right)\end{array}$

Furthermore, the correlation value calculator 63e1, calculates sliding correlation values AC^y_s(k_s, 2z_s) at discrete times k_s (=1 to (N_u−N_w)/N_TR) in an m_sth (m_s (even number)=2z_s) transmission cycle T_r (each transmission cycle T_r) according to Equation (94). The correlation value calculator 63e1_s outputs the sliding correlation values AC^y_s(k_s, 2z_s) calculated according to Equation (94) to the coherent integrator 64e1_s. In Equation (94), the asterisk “*” is the complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}94\right]& \\ {\mathrm{AC}}_s^y=\left(k_s,2z_s\right)=\sum _{n=1}^L{\mathrm{XP}}_s^y\left(N_v\left(2z_s-1\right)+\frac{N_o\left(n-1\right)}{N_{\mathrm{TR}}}+k_s\right)B_n^{*}& \left(94\right)\end{array}$

Although in each of the embodiments including this embodiment the correlation value calculator 63e1_s performs calculations at discrete times k_s=1 to (N_u−N_w)/N_TR, the measurement range (discrete time k_s range) may be narrowed further to, for example, k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR according to the range of presence of a target TAR_s which is a measurement target of the radar apparatus 10. With this measure, the radar apparatus 10 can further reduce the amount of calculation of the correlation value calculator 63e1_s. That is, the radar apparatus 10 can reduce the power consumption further as a result of reduction in the calculation amount of the signal processer 6e1_s.

Where the correlation value calculator 63e1_s calculates sliding correlation values AC_s(k_s, m_s) at discrete times k_s=N_w/N_TR+1 to (N_u−N_w)/N_TR, the radar apparatus 10 can omit measurement of a reflection wave signal in each transmission interval T_w of the radar transmission signal.

In the radar apparatus 10, even if a radar transmission signal coming from each radar transmitter goes around to enter the radar receiver directly, a measurement can be performed with its influence eliminated. With the above restriction of the measurement range (discrete time k_s range), the coherent integrator 64e1_s and the distance estimator 65e1_s also operate in the same restricted measurement range.

The coherent integrator 64e1_s receives the sliding correlation values AC^y_s(k_s, 2z_s−1) and AC^y_s(k_s, 2z_s) that are output from the correlation value calculator 63e1_s. The coherent integrator 64e1_s adds together sliding correlation values AC^y_s(k_s, 2z_s−1) and AC^y_s(k_s, 2z_s) in a prescribed number (2NP) of transmission cycles T_r (a period 2NP×T_r) on the basis of sets of sliding correlation values AC^y_s(k_s, 2z_s−1) and AC^y_s(k_s, 2z_s) that have been calculated in the two (odd-numbered and even-numbered) transmission cycles T_r for the respective discrete times k_s.

The coherent integrator 64e1_s calculates a v_sth coherent integration value ACC^y_s(k_s, v_s) for each discrete time k_s by adding together, for each discrete time k_s, sliding correlation values AC^y_s(k_s, 2z_s−1) and AC^y_s(k_s, 2z_s) in the prescribed number 2NP of periods (period NP×T_r) according to Equation (95). Parameter 2NP represents the number of times of coherent integration performed in the coherent integrator 64e1_s. The coherent integrator 64e1_s outputs the calculated coherent integration values ACC^y_s(k_s, v_s) to the distance estimator 651_s.

$\begin{array}{cc}\left[\mathrm{Formula}95\right]& \\ {\mathrm{ACC}}_s^y\left(k_s,v_s\right)=\sum _{m_s=1}^{2\mathrm{NP}}{\mathrm{AC}}_s^y\left(k_s,\mathrm{NP}\left(v_s-1\right)+m_s\right)& \left(95\right)\end{array}$

By setting the prescribed number 2NP at an integer multiple of 2π/φ_s in Equation (95), the coherent integrator 64e1_s can reduce influences of circuit errors even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. That is, by setting the prescribed number 2NP at an integer multiple of 2π/φ_s, the radar apparatus 10 can prevent degradation of the target ranging performance even if a reflection wave signal contains circuit errors such as a DC offset component and IQ imbalance. Furthermore, the radar apparatus 10 can improve the reception quality (SNR) of a reflection wave signal by suppressing noise components contained in the reflection wave signal by performing 2NP times of coherent integration.

As such, the radar apparatus 10 according to the fifth embodiment can provide advantages that are equivalent to the advantages of the radar apparatus 10 according to the fourth embodiment even in the case of using a complementary code as a transmission code.

Although the various embodiments have been described above with reference to the drawings, it goes without saying that this disclosure is not limited to those examples. It is apparent that those skilled in the art would conceive various changes or modifications within the confines of the claims. And such changes or modifications should naturally be construed as being included in the technical scope of the invention.

In the above-described first embodiment, the pulse transmission controller 21_s is provided in each radar transmitter Tx_s of each sector radar R_s (s=1, 2). However, the pulse transmission control unit 21_s may be provided outside each of the two sector radars SR_s (s=1, 2) or a single pulse transmission control unit may be shaped by the two sector radars SR_s (s=1, 2). That a single pulse transmission control unit may be shaped by the two sector radars SR_s (s=1, 2) also applies to the second and third embodiments.

In the above-described second embodiment, the first code generater 22b1_s generates a complementary code sequence A_n having a code length L and the second code generater 22b2_s generates a complementary code sequence B_n having the code length L. However, the invention is not limited to such a case. The same advantages can be obtained even by a configuration in which the first code generater 22b1_s and the second code generater 22b2_s (s=1) generate complementary code sequences A_n and B_n having a code length L, respectively, and the first code generater 22b1_s and the second code generater 22b2_s (s=2) generate complementary code sequences U_n and V_n having the code length L, respectively, in which the code U_n is different from the code A_n and the code V_n is different from the code B_n (see FIG. 24).

FIG. 24 is an explanatory diagram showing relationships between individual transmission intervals, individual transmission cycles, and individual transmission phase shift components of transmission signals in a case that different pairs of complementary code sequences are used in the sector radars SRb_s (s=1, 2) which constitute the radar apparatus according to the second embodiment. In FIG. 24, the complementary code sequences U_n and V_n are in a complementary code relationship.

Furthermore, in the radar apparatus 10, the interference between the sector radars SRb_s (s=1, 2) can be suppressed further by employing, as the code sequences U_n and A_n, code sequences having small cross-correlation values.

Since the cross-correlation values between the code sequences U_n and A_n determine the amount of interference between the codes, it is the best that they have cross-correlation values being equal to zero. However, it is preferable to employ code sequences having cross-correlation values being smaller than or equal to 0.1 because it is appropriate to at least make the amount of interference between the codes smaller than or equal to 20 dB.

Still further, in the radar apparatus 10, the interference between the sector radars SRb_s (s=1, 2) can be suppressed further by employing, as the code sequences V_n and B_n code sequences having small cross-correlation values.

What is more, in the radar apparatus 10, the interference between the sector radars SRb_s (s=1, 2) can be suppressed even further by employing, as the code sequences U_n, V_n, A_n, and B_n, such code sequences that the sums of the cross-correlation values between the code sequences U_n and A_n and the cross-correlation values between the code sequences V_n and B_n are equal to zero.

This will be explained below. A cross-correlation result (cross-correlation values) R_AU(τ) between the one code sequence A_n of the former of the complementary code sequences (A_n, B_n) and the complementary code sequences (U_n, V_n) and the one code sequence U_n of the latter complementary code sequence is calculated according to Equation (96).

A cross-correlation calculation result (cross-correlation values) R_BV(τ) between the other code sequence B_n of the former complementary code sequence and the other code sequence V_n of the latter complementary code sequence is calculated according to Equation (97). Symbol R represents a cross-correlation value calculation result (cross-correlation values). However, it is assumed that each of the complementary code sequences A_n and B_n is zero when n>L or n<1 (i.e., A_n=0, B_n=0, U_n=0, and V_n=0 when n>L or n<1). The asterisk “*” is a complex conjugate operator.

$\begin{array}{cc}\left[\mathrm{Formula}96\right]& \\ R_{\mathrm{AU}}\left(\tau \right)=\sum _{n=1}^LA_nU_{n+\tau }& \left(96\right)\\ \left[\mathrm{Formula}97\right]& \\ R_{\mathrm{BV}}\left(\tau \right)=\sum _{n=1}^LB_nV_{n+\tau }^{*}& \left(97\right)\end{array}$

The cross-correlation value calculation result R_AU(τ) calculated according to Equation (96) has a peak when the delay time (or shift time) τ is equal to 0 and has range sidelobes for the delay times τ being not equal to 0. Likewise, the cross-correlation calculation result R_BV(τ) calculated according to Equation (97) has a peak when the delay time τ is equal to 0 and has range sidelobes for the delay times τ being not equal to 0.

In the radar apparatus 10, the interference between the sector radars SRb_s (s=1, 2) can be suppressed even further by employing, as the (A_n, B_n) and the complementary code sequences (U_n, V_n), such code sequences that the sums of values, at the same delay times τ, of cross-correlation value calculation results (R_AU(τ) and R_BV(τ)) are equal to zero irrespective of the delay time τ (see Equation (98)).

[Formula 98]

R _AU(τ)+R_BV(τ)=0  (98)

The present application is based on Japanese Patent Application No. 2011-252100 filed on Nov. 17, 2011, the disclosure of which is incorporated herein by reference.

INDUSTRIAL APPLICABILITY

This disclosure is useful when applied to a radar apparatus which, in the case where plural sector radars are installed being opposed to each other, suppresses interference between the sector radars with a simple configuration by making it unnecessary to synchronize transmission cycles between the sector radars opposed to each other.

DESCRIPTION OF SYMBOLS

• 10: Radar apparatus
• 2 _s, 2b_s, 2d1_s, 2e1_s: Transmission signal generater
• 3 _s, 31_s: RF transmitter
• 4 _s, 41_s: RF receiver
• 5 _s, 51_s: VGA unit
• 6 _s, 6a_s, 6b_s, 6c_s, 6d1_s, 6e1_s: Signal processer
• 21 _s, 21b_s, 21d_s, 21e_s: Pulse transmission controller
• 22 _s, 22b_s, 221_s, 22d1_s, 22e1_s: Code generater
• 22 b 1 _s, 22e11_s: First code generater
• 22 b 2 _s, 22e12_s: Second code generater
• 23 _s, 23b_s, 231_s: Modulater
• 24 _s, 24b_s, 241_s: LPF
• 25, 25b_s, 25d1_s, 25e1_s: Sth transmission phase shifter
• 26 _s, 26b_s, 261_s: D/A converter
• 31 _s: Quadrature modulater
• 32 _s, 42_s: Frequency converter
• 33 _s, 41_s: Amplifier
• 43 _s: Quadrature detector
• 61 _s, 611_s: A/D converter
• 62 _s, 62a_s, 62b_s, 62c_s, 62d1_s, 62e1_s: Sth reception phase shifter
• 63 _s, 63a_s, 63b_s, 63c_s, 63d1_s, 63e1_s: Correlation value calculator
• 64 _s, 64a_s, 64b_s, 64c_s, 64d1_s, 64e1_s: Coherent integrator
• 65 _s, 651_s: Distance estimator
• Rx_s, Rxa_s, Rxb_s, Rxc_s: Radar receiver
• SR_s, SRa_s, SRb_s, SRc_s, SRd_s, SRe_s: Sector radar
• Tx_s, Txb_s: Radar transmitter

Claims

1. A radar apparatus comprising: a first radar transmitter that transmits a first radar transmission signal generated using a first code sequence having a prescribed code length from a first transmission antenna as a first radio-frequency signal; anda second radar transmitter that transmits a second radar transmission signal generated using a second code sequence having a prescribed code length from a second transmission antenna as a second radio-frequency signal, wherein:the first radar transmission signal is a signal generated by modulating the first code sequence that has been phase-shifted on the basis of a first transmission timing signal;the second radar transmission signal is a signal generated by modulating the second code sequence that has been phase-shifted on the basis of a second transmission timing signal; anda phase shift given to the first code sequence is opposite to a phase shift given to the second code sequence.

2. The radar apparatus according to claim 1, wherein the first radar transmitter comprises: a first pulse transmission controller that generates the first transmission timing signal every transmission cycle;a first radar code generater that generates the first code sequence;a first modulater that generates a first baseband signal by modulating the first code sequence on the basis of the first transmission timing signal;a first transmission phase shifter that shifts the phase of the first baseband signal on the basis of the first transmission timing signal; anda first RF transmitter that converts the phase-shifted first baseband signal into the radio-frequency first radar transmission signal and transmits the radio-frequency first radar transmission signal from the first transmission antenna.

3. The radar apparatus according to claim 2, wherein: the first radar code generater comprises: a first code generater that generates a transmission code of a third code sequence that is a complementary code sequence having a prescribed code length on the basis of the first transmission timing signal generated every odd-numbered transmission cycle; anda second code generater that generates a transmission code of a fourth code sequence that is a complementary code sequence having the prescribed code length on the basis of the first transmission timing signal generated every even-numbered transmission cycle;the first code sequence consists of the third code sequence and the fourth code sequence; andthe first transmission phase shifter gives a phase shift corresponding to two transmission cycles.

4. The radar apparatus according to claim 1, wherein the second radar transmitter comprises: a second pulse transmission controller that generates the second transmission timing signal every transmission cycle;a second radar code generater that generates the second code sequence;a second modulater that generates a second baseband signal by modulating the second code sequence on the basis of the second transmission timing signal;a second transmission phase shifter that shifts the phase of the second baseband signal on the basis of the second transmission timing signal; anda second RF transmitter that converts the phase-shifted second baseband signal into the radio-frequency second radar transmission signal and transmitting the radio-frequency second radar transmission signal from the second transmission antenna.

5. The radar apparatus according to claim 4, wherein: the second radar code generater comprises: a third code generater that generates a transmission code of a fifth code sequence that is a complementary code sequence having a prescribed code length on the basis of the second transmission timing signal generated every odd-numbered transmission cycle; anda fourth code generater that generates a transmission code of a sixth code sequence that is a complementary code sequence having the prescribed code length on the basis of the second transmission timing signal generated every even-numbered transmission cycle;the second code sequence consists of the fifth code sequence and the sixth code sequence; andthe second transmission phase shifter gives a phase shift corresponding to two transmission cycles.

6. The radar apparatus according to claim 1, further comprising: a first radar receiver that detects presence/absence of a target on the basis of a reflection wave signal that is produced through reflection of the radio-frequency first radar transmission signal by the target, the first radar receiver comprising:a first RF receiver that receives the reflection wave signal by a first reception antenna and converts the received reflection wave signal into a baseband first reception signal;a first reception phase shifter that gives an opposite phase shift that is opposite to the first phase shift given by the first transmission phase shifter to the first reception signal supplied from the first RF receiver on the basis of the first transmission timing signal;a first correlation value calculator that calculates correlation values between the opposite-phase-shift-given first reception signal and the first transmission code; anda first coherent integrator that adds together correlation values calculated by the first correlation value calculator in a prescribed number of transmission cycles.

7. The radar apparatus according to claim 6, further comprising: a second radar receiver that detects presence/absence of the target on the basis of a reflection wave signal that is produced through reflection of the radio-frequency second radar transmission signal by the target, the second radar receiver comprising:a second RF receiver that receives the reflection wave signal by a second reception antenna and converts the received reflection wave signal into a baseband second reception signal;a second reception phase shifter that gives an opposite phase shift that is opposite to the second phase shift given by the second transmission phase shifter to the second reception signal supplied from the second RF receiver on the basis of the second transmission timing signal;a second correlation value calculator that calculates correlation values between the opposite-phase-shift-given second reception signal and the second transmission code; anda second coherent integrator that adds together correlation values calculated by the second correlation value calculator in a prescribed number of transmission cycles.

8. The radar apparatus according to claim 7, wherein: the first radar receiver further comprises: a first VGA unit that adjusts an output level of the first reception signal supplied from the first RF receiver; anda first A/D converter that converts an analog first reception signal whose output level has been adjusted by the first VGA unit into a digital first reception signal; andthe second radar receiver further comprises: a second VGA unit that adjusts an output level of the second reception signal supplied from the second RF receiver; anda second A/D converter that converts an analog first reception signal whose output level has been adjusted by the second VGA unit into a digital second reception signal.

9. The radar apparatus according to claim 6, wherein: the first radar receiver further comprises a first distance estimator that estimates a distance to the target on the basis of correlation value addition results supplied from the first coherent integrator; andthe second receiver further comprises a second distance estimator that estimates a distance to the target on the basis of correlation value addition results supplied from the second coherent integrator.

10. The radar apparatus according to claim 6, wherein the third, fourth, fifth, and sixth code sequences are the same code sequence and have the same code length.

11. The radar apparatus according to claim 6, wherein the third, fourth, fifth, and sixth code sequences are different code sequences and have the same code length.

12. The radar apparatus according to claim 6, wherein the third code sequence or the fourth code sequence has a different code length than the fifth code sequence or the sixth code sequence.

13. The radar apparatus according to claim 1, further comprising: a third radar transmitter that generates a third baseband signal using a seventh code sequence having the prescribed code length on the basis of the first transmission timing signal, shifting the phase of the third baseband signal on the basis of the first transmission timing signal, converting the phase-shifted third baseband signal into a radio-frequency third radar transmission signal, and transmitting the latter from a third transmission antenna; anda fourth radar transmitter that generates a fourth baseband signal using an eighth code sequence having the prescribed code length on the basis of the second transmission timing signal, shifting the phase of the fourth baseband signal on the basis of the second transmission timing signal, converting the phase-shifted fourth baseband signal into a radio-frequency fourth radar transmission signal, and transmitting the latter from a fourth transmission antenna, wherein:the phase shift given to the third baseband signal is opposite to the phase shift given to the fourth baseband signal;the phase shift given to the first baseband signal is opposite to the phase shift given to the third baseband signal; andthe phase shift given to the second baseband signal is opposite to the phase shift given to the fourth baseband signal.

14. The radar apparatus according to claim 1, wherein: a phase shift of 90° is given in the first radar transmitter; anda phase shift of −90° is given in the second radar transmitter.

15. The radar apparatus according to claim 11, wherein sums of cross-correlation values between the third code sequence and the fifth code sequence and cross-correlation values between the fourth code sequence and the sixth code sequence are equal to zero.