Patent Application
20080117099
The present invention relates to an antenna radar system which may be used in automobile technology, as well as a method for its operation.
In motor vehicle technology, up to now, almost exclusively long range radar (LRR) systems have been used for the long-range recording of detection targets. However, in that field, too, there is an increasing requirement for using short range radar (SRR) systems having short range detection, for instance, for carrying out clearance measurements in vehicle columns (automatic drive off in bumper to bumper driving or the like) or for use as a parking aid.
In general, the detection field for short-range applications has a substantially greater opening angle in comparison to long-range applications. But because of smaller so-called EIRP values in the short-range applications, the latter also have a shorter range. The EIRP (equivalent isotropic radiated power) value mentioned represents a pure operand, and tells how great a transmission power one would have to supply an antenna that radiates in all spatial directions isotropically, in order to achieve the same power flow density in the long-range field as would be the case using a bundling directional antenna in its main transmit direction. For these reasons, it is almost impossible to provide a common antenna aperture for the LRR and the SRR function.
In the antenna radar systems known from the related art, that are suitable for the automotive field, the incoming signal is frequency-translated downwards (“down converted”) in homodyne fashion, using an unbalanced one-diode converter (mixer). This causes noise in the amplitude-modulated output signal, whereby, as a result, the sensitivity of the radar system to objects at a short distance are considerably limited.
Besides that, antenna radar systems already being used outside of automotive technology, that are optimized towards short-range recording or detection, only achieve a minimum measuring distance of 0.5 m, at this time. However, in the above-mentioned driving situations (bumper to bumper traffic, etc.), as short as possible a minimum measuring distance, in the range of a few decimeters, is desirable.
Therefore, an object of the exemplary embodiment and/or exemplary method of the present invention is based on further developing the antenna radar system of the type described at the outset in such a way that the short-range weakness described for the known systems is removed. However, this further development should be oriented, if possible, towards existing antenna radar systems, in order to keep development costs and manufacturing costs as low as possible.
The exemplary embodiment and/or exemplary method of the present invention involves providing a push-pull mixer, in an antenna radar system discussed here, in a short-range antenna path, which utilizes the same or at least a very similar intermediate frequency as is provided, as is well known, for a phase lock loop (PLL).
The antenna radar system proposed according to the exemplary embodiment and/or exemplary method of the present invention can be operated, using the push-pull mixer in the short range using heterodyne mixing, the frequency offsets of potential detection targets coming to lie far outside the phase noise of a local oscillator (LO) situated in the short-range antenna path, and the amplitude modulated noise (AM noise) in the LO path is suppressed.
Thus, the push-pull mixer suppresses the amplitude modulation noise that is mostly present in the LO path, which is automatically mixed together with the carrier frequency of the transmit signal in sidebands situated around the intermediate frequency. As is well known, the carrier frequency itself does not vary in its amplitude in response to amplitude-modulated signals. Rather, the modulation occurs in the form of signal components having frequencies somewhat above and below the carrier frequency, which signal components are generally designated as “sidebands”.
As a result, using the exemplary embodiment and/or exemplary method of the present invention, short-range measurements having a resolution of a few decimeters are made possible.
In one embodiment, the LO of the push-pull mixer is fed with the fourth (4th) harmonic of a reference oscillator; however, instead of using the fourth harmonic, two frequency doublers may also be provided, which has the additional advantage that a maximum LO power can be held in reserve for the push-pull mixer.
In the automotive field, in the radar systems discussed here, monostatic antennas are mostly used in which the irradiated and radiated signals (so-called “RX/TX feeds”) use a common aerial lens. The polarization axes of these two signals mostly have an angle of 45° in the radar systems mentioned, in order to ensure that the signals, coming from an approaching vehicle equipped with the same radar, are received cross-polarized with respect to one's own receive signal. Based on this measure, disturbing interferences between the signals of the two vehicles are effectively suppressed. The antenna radar system according to the exemplary embodiment and/or exemplary method of the present invention can be designed for this purpose in such a way that the apertures of the long-range and the short-range functions are operated cross-polarized, a timed multiplexer of long-range and short-range mode being implemented using switchable transmit preamplifiers in the transmit path of the long-range and short-range radar functions. For, based on the antenna characteristics known per se of radar antennas, that is, the predefined primary and secondary lobes in the radiation and irradiation characteristics, without the measures that were mentioned, there would be a crossfeed (coupling) between these two functions. Based on the cross-polarization for the short-range and the long-range functions, an extremely effective decoupling is achieved between these two functions, so that these functions are able to be integrated, without trouble, into a single antenna radar system.
Using the proposed heterodyne frequency translation (mixing), an existing, predominantly long-range antenna radar system (LRR) can be upgraded by a high-resolution short-range detection, in order, for instance, to make available a combination system for both the short range and the long range.
The antenna radar according to the exemplary embodiment and/or exemplary method of the present invention, having the advantages mentioned, can also be used in bistatic antennas which, as is well known, have separate transmit and receive paths. Even based on this path separation, a cross feed of the transmit signal into the receiver is minimized.
a and 2b show an overview representation of frequencies occurring basically in response to down converters and up converters using a converter shown in
a and 4b show “typical” signal patterns in the time range of a one diode converter (a) and a push-pull mixer (b) in comparison.
As in available high frequency (HF) transmit/receive technology, for instance, messaging technology, the antenna radar systems included here, in order to make possible the reception of very short wavelengths, which, in turn, has connected with it relatively high location resolution, have a phase lock loop “PLL” 70, which is modulated by a digital divider N, and which has available to it an integrated voltage-controlled oscillator (VCO) which is used for generating a carrier signal. As can be seen in
In the signal receiving detectors mentioned, one distinguishes, as is well known, between a direct detection and a heterodyne detection. In the direct detection, an incoming receive signal is directly processed further, whereas in heterodyne detection an additional signal f_LO, fed in by a local oscillator (LO) 30, is overlaid on receive signal f_E. These two frequencies are mixed in order to obtain the intermediate frequency (f_ZF) that was mentioned. Intermediate frequency f_ZF is in a frequency range which is easy to amplify and which makes possible the use of frequency selection circuits having in each case the desired bandwidth. Such a heterodyne receiver is shown schematically in
Normally, the intermediate frequency is predefined in a fixed manner, and only oscillator 30 and input circuit 10 are tuned to each other. The crux of the circuit is mixer stage 20, in which receive signal f_E is overlaid with oscillator frequency f_LO. If one were to fit in a frequency-independent resistor at the output of mixer stage 20, the four following different frequencies would result:
Generally, only intermediate frequency f_ZF is of interest, and therefore one connects a bandfilter 40, that is tuned to f_ZF, to the output line of mixer stage 20. From there, it goes for further amplification and selection to a post-connected intermediate frequency amplifier 50 and a detector 60, that is, in turn, post-connected to the latter, used for the final demodulation of the amplitude-modulated input signal.
Accordingly, using the heterodyne receiver shown schematically in
The above-described frequency conversion either takes place using “up converters” or using “down converters”, depending on whether the desired output signal is to lie above or below the input signal. Such a converter basically represents a three-port junction having inputs for input frequency f_E and local oscillator frequency f_LO, as well as an output for intermediate frequency f_ZF, the mixing representing a nonlinear process in which at least two of the variables named are multiplied by each other. An ideal converter, between ports ‘E’ and ‘ZF’ behaves like an adapted, lossy two-port, which simply additionally undertakes a frequency shift. At input ‘LO’ the local oscillator signal is supplied having the frequency f_LO, which determines the difference between f_E and f_ZF and which, as a rule, is substantially stronger than the two other signals. For the connection between the three frequencies mentioned, F_ZF=±(f_E−f_LO) applies.
In a down converter, the input signal, having a frequency f_E, has a higher frequency than that of the desired output signal, f_ZF. Depending on whether f_E is greater or smaller an f_LO, the positive or negative sign applies in the above equation. The connection between these three frequencies are shown in
Besides the desired output frequency f1, there also especially occurs the so-called “image frequency” f_SP, which has a frequency of f_SP=f_LO−f_ZF (see
The meaning of the image frequency (in
The antenna radar system according to the exemplary embodiment and/or exemplary method of the present invention shown in
The synchronous operation mentioned is only possible, without interference of the two functions with each other, because between the SRR functions and the LRR functions in this exemplary embodiment, a cross polarization occurs, which has the effect of a sufficient signal technology insulation between the two functions. In the cross polarization, the polarized signals of the SRR function and the LRR function are operated in a manner known per se, polarized perpendicularly to each other, whereby it is prevented in many situations that the signals are able to be superposed at all, constructively or destructively.
The feeds indicated only schematically in
With regard to the patch arrays, one should note that their technical details are not important in the present connection. Such a patch array for a high frequency antenna is described in detail, for example, in the simultaneously filed Patent Application (having Application file number R. 307998), to which we make full reference in this connection. The feeds mentioned, 290-305, 237 and 365 are situated spatially separated from one another for the reasons named.
The circuit shown in
An input signal 200 supplied by a phase lock loop (=PLL) that is not shown is first of all used to operate a voltage-controlled oscillator (VCO) 205. The oscillating signal generated by the VCO (here a transmit VCO) 205 is fed to short-range transmit antenna 237 using a power splitter 210, 215. This input signal 200 is supplied to a converter 320, which may use capacitive coupling element 310, and its input signal, in turn, originates with a source 340. To do this, the fourth harmonic of a stable reference oscillator 340, which, in the current exemplary embodiment, oscillates at a frequency of 4*18.65 GHz=74.6 GHz, and which, in the present instance, oscillates at a frequency of 76.5 GHz±125 MHz, is mixed at a diode 320 that is situated serially in SRR path 310-365. The fourth harmonic mentioned is generated in the present exemplary embodiment from the signal supplied by reference oscillator 340, using two frequency doublers 330, 335 that are connected in series.
As a result, the intermediate frequency yielded in response to the down conversion is in this example at 76.5 GHz−74.6 GHz=1900 MHz. The image frequency that was mentioned, which has a critical effect, as mentioned above, especially in the case of interference signals when these are mixed to the same intermediate frequency, is at 72.7 GHz in the present example.
The frequency generated in 330, 335 and 340 is supplied to a push-pull mixer 345-360. The exact method of functioning of push-pull mixer 345-360 will be described below in greater detail in light of
Based on the heterodyne down conversion of the intermediate frequency signal, that will also be described below, and the antenna signal supplied via SRR feed 365, the frequency offsets of potential detection targets occur widely outside the phase noise of LO 330, 335 and 340. The phase noise is, for example, mixed by reflection at RX feed 365 in a DC-near frequency range.
The AM noise of LO 330, 335 and 340, on the other hand, is mixed directly by demodulation in the mixer in the DC-near frequency range. The AM suppression takes place in the push-pull mixer by destructive interference based on the different polarity of the two diodes.
In the technical implementation of the exemplary embodiment and/or exemplary method of the present invention, the basic modulation form of known systems can be maintained, whereby one may extensively use existing electronic systems (VCO, PLL, reference StaLO, etc.).
It should further be noted that, in order to achieve the above-mentioned properties in an alternatively possible pulse radar (instead of the present continuous wave radar), in contrast to the design attempt described above, an implementation of rapid switches, their drivers and a highly precise, variable delay electronic system would be required. However, the components mentioned are very costly.
For the down conversion of the signals supplied by VCO 205, long-range antenna path (LRR path) 210-305 has four unbalanced one diode mixers 270-285. Mixer diodes 270-285 lie in each case separately in the path of each Tx/Rx feed 290-305. Mixer diodes 270-285 functionally correspond to switches, in this context, which are opened and closed in the clock pulse of oscillator 205. Via four patch antennas 290-305 also designed as patch arrays, the Tx signals reach a focussing unit (such as a lens) and are radiated from there. The reflected components reach patch antennas 290-305 via the focussing unit, and are mixed into the baseband, using mixer diodes 270-285. The lower frequency ZF signal, yielded by the down conversion using mixer diodes 270-285, is then supplied, in turn, via a TP structure 240-265, that is post-connected to patch antennas 290-305 and mixer diodes 270-285 and brings together the entire received power, to a second preamplifier 220 that is provided with a bias voltage 225.
a and 4b illustrate the method of functioning of a one diode mixer (
As may be seen in
As may be seen in
The right half of
| Number | Date | Country | Kind |
|---|---|---|---|
| 102004046632.7 | Mar 2004 | DE | national |
| Filing Document | Filing Date | Country | Kind | 371c Date |
|---|---|---|---|---|
| PCT/EP05/53816 | 8/3/2005 | WO | 00 | 3/22/2007 |
