Patent Grant
11380481
The present disclosure generally relates to radio transmitter with transmit signal strength indicator, and more particularly to a radio transmitter and method that accurately estimates a transmit power under a variety of load conditions.
A radio transmitter comprises a power amplifier configured to output a first RF (radio frequency) signal to a local antenna, which converts the RF signal into an electromagnetic wave that can propagate in the air and received by a remote antenna, which converts the electromagnetic wave into a second RF signal that is received by a radio receiver. It is desirable that a power of electromagnetic wave is sufficiently high that the second RF signal is detectable by the radio receiver but also not too high lest the power amplifier might consume more energy than needed and the electromagnetic wave might cause a strong interference to other radio receivers nearby that are not supposed to receive the first RF signal. The power of the electromagnetic wave is approximately equal to a power of first RF signal, which depends on a power of an input signal to the power amplifier, a voltage gain of the power amplifier, and an impedance matching between the power amplifier and the local antenna. In general, the power of the first RF signal is unknown without using a TSSI (transmit signal strength indicator). The TSSI is configured to estimate a power of an output of a power amplifier.
Zolfaghari discloses a TSSI in U.S. Pat. No. 7,620,371. The TSSI disclosed therein detects an envelope of a RF signal and assumes a certain load resistance from the antenna that receives the RF signal. The assumption is no always valid and thus the method disclosed therein may not be applicable in a general application scenario.
What is desired is a method to accurately estimate a transmit power under various load conditions.
In an embodiment, a radio transmitter comprises: a power amplifier configured to receive an input voltage signal and output an output voltage signal; a transformer configured to receive the output voltage signal and output a load voltage signal to a load; a sensing inductor configured to output a sensed current signal in accordance with a magnetic coupling with the transformer; a digitally controlled phase shifter configured to receive the output voltage signal and output a phase-shifted voltage signal in accordance with a phase control code; a mixer configured to output a mixed current signal in accordance with a mixing of the sensed current signal and the phase-shifted voltage signal; and a transimpedance amplifier with of a low-pass response configured to convert the mixed current signal into a mean voltage signal.
In an embodiment, a method comprises: receiving an input voltage signal; converting the input voltage signal into an output voltage signal using a power amplifier; transforming the output voltage signal into a load voltage signal at a load using a transformer; outputting a sensed current signal using a sensing inductor in accordance with a magnetic coupling with the transformer; converting the output voltage signal into a phase-shifted voltage signal in accordance with a phase control code using a digitally control phase shifter; mixing the sensed current signal with the phase-shifted voltage signal into a mixed current signal using a mixer; converting the mixed current signal into a mean voltage signal using a transimpedance amplifier with a low-pass response; and obtaining an estimate of a power delivered into the load by multiplying a value of the mean voltage signal with a power scaling factor.
The present disclosure is directed to radio transmitter and transmit signal strength indicator. While the specification describes several example embodiments of the disclosure considered favorable modes of practicing the invention, it should be understood that the invention can be implemented in many ways and is not limited to the particular examples described below or to the particular manner in which any features of such examples are implemented. In other instances, well-known details are not shown or described to avoid obscuring aspects of the disclosure.
Persons of ordinary skill in the art understand terms and basic concepts related to microelectronics that are used in this disclosure, such as “voltage,” “current,” “power,” “CMOS (complementary metal oxide semiconductor),” “impedance,” “inductor,” “magnetic coupling,” “resistor,” “capacitor,” “resistance,” “capacitance,” “resistive impedance,” “operational amplifier,” “power amplifier,” “transimpedance amplifier,” “mixer (or frequency mixer),” “transformer,” “load,” “frequency,” “phase,” “phase shift,” “differential signal,” and “negative feedback.” Terms like these are used in a context of microelectronics, and the associated concepts are apparent to those of ordinary skills in the art and thus will not be explained in detail here.
Persons of ordinary skills in the art understand basic trigonometry and do not need an explanation of, for instance,
and a time average of a periodic sinusoidal signal is zero.
Throughout this disclosure, a “signal” is either a voltage or a current carrying a certain information.
Throughout this disclosure, a differential signaling scheme is used extensively. A differential voltage signal comprises two single-ended voltage signals denoted with suffixes “+” and “−,” respectively, attached in subscript, and a value of the differential voltage signal is represented by a difference between said two single-ended voltages. For instance, VI (VO, VS, VM) comprises VI+ (VO+, VS+, VM+) and VI− (VO−, VS−, VM−) and a value of VI (VO, VS, VM) is represented by a difference between VI+ (VO+, VS+, VM+) and VI− (VO−, VS−, VM−). Likewise, a differential current signal comprises two currents denoted with suffixes “+” and “−,” respectively, attached in subscript. For instance, IS (IM) comprises IS+ (IM+) and IS− (IM−) and a value of IS (IM) is represented by a difference between IS+ (IM+) and IS− (IM−).
This present disclosure is described in an engineering sense. For instance, “X is equal to Y” means “a difference between X and Y is smaller than a specified engineering tolerance”; “X is much smaller than Y” means “X divided by Y is smaller than an engineering tolerance”; and “X is zero” means “X is smaller than a specified engineering tolerance.”
A functional block diagram of a transmitter 100 in accordance with an embodiment of the present disclosure is shown in
The output voltage signal VO can be modeled by the following equation:
VO=AO cos(ωt), (1)
where, AO denotes an amplitude of the output voltage signal VO, ω denotes an angular frequency, and t denotes a time variable.
Along with the output voltage signal VO, an output current signal IO is also received by the transformer 120. The output current signal IO can be modeled by the following equation:
IO=BO cos(ωt−θ1), (2)
where, BO denotes an amplitude of the output current signal IO, and θ1 denotes a phase delay between VO and IO. Assume the transformer 120 introduces a loss factor G0. Then, a power PL delivered to the load 180 is equal to an average of a product of the output voltage signal VO and the output current signal IO times the loss factor G0, i.e.
PL=G0<VOIO> (3)
Here, <⋅> denotes a time-averaging function. Substituting equations (1) and (2) into equation (3), we can obtain:
Note that VOIO contains a time-varying term pertaining to cos(2ωt−θ1) that vanishes after time-averaging.
The sensing inductor 130 performs a sensing function so that the sensed current signal IS is proportional to the output current signal IO with a phase delay. Mathematically, that can be modeled by the following equation:
IS=BOG1 cos(ωt−θ1−θ2), (5)
where, G1 and θ2 denote a proportional constant and a phase delay, respectively, pertaining to the magnetic coupling between the sensing inductor 130 and the transformer 120.
The digitally controlled phase shifter 140 performs a phase shifting function so that the shifted voltage signal VS is proportional to the output voltage signal VO with a phase shift controlled by the phase control code CP. Mathematically, that can be modeled by the following equation:
VS=AOG2 cos(ωt−θ3), (6)
where, G2 and θ3 denote a proportional constant and a phase shift, respectively, pertaining to the digitally controlled phase shifter 140.
Mixer 160 performs a multiplication function that can be modeled by the following equation
IM=G3VSIS, (7)
where, G3 denotes a gain factor of the mixer 160.
TIA 170 performs a time-averaging function that can modeled by the following equation:
VM=G4<IM>, (8)
where, G4 denotes a gain factor of the TIA 170.
From equations (5), (6), (7), and (8), we can obtain
VM=G3G4<BOG1 cos(ωt−θ1−θ2)AOG2 cos(ωt−θ3)>, (9)
which can be simplified to
From equations (4) and (10), we obtain
PL=GVM (11)
where
is a power scaling factor, provided θ2=θ3, i.e. a phase shift amount of the digitally controlled phase shifter 140 equals a phase delay of the sensing inductor 130. In other words, VM times the power scaling factor G is equal to PL, provided θ2=θ3.
The phase control code CP is set such that the condition θ2=θ3 is met. During an initial calibration, a proper setting of CP that leads to the condition θ2=θ3 is found, and so is the scaling factor G. Then, G·VM is an accurate estimate of the power delivered to the load 180.
Power amplifiers are well known in the prior art. The power amplifier 110 of
In an embodiment, transmitter 100 is embodied by an integrated circuit fabricated on a silicon substrate using a CMOS (complementary metal oxide semiconductor) process with a multi-layer structure including a first metal layer, a second metal layer, and a via layer configured to allow an electrical connection between a first metal on the first metal layer and a second metal on a second metal layer. A top view of a layout of the transformer 120 and the sensing inductor 130 in accordance with an embodiment of the present disclosure is shown in
The digitally controlled phase shifter 140 comprises a poly-phase filter 141 configured to receive the output voltage signal VO and output a four-phase voltage signal comprising a 0-degree phase V0, a 90-degree phase V90, a 180-degree phase V180, and a 270-degree phase V270, and a digitally controlled phase rotator 142 configured to receive the four-phase voltage signal and output the phase-shifted voltage signal VS in accordance with the phase control code CP.
A poly-phase filter receives a two-phase signal and output a four-phase signal. The output voltage signal VO, which in differential embodiment comprises VO+ and VO−, is a two-phase signal wherein VO+ is a first phase, VO− is a second phase, and VO+ and VO− are 180-degrees out of phase. A schematic diagram of an embodiment of the poly-phase filter 141 is shown in
A digitally controlled phase rotator receives a four-phase voltage signal and output a two-phase voltage signal by means of a phase interpolation in accordance with a control code. For digitally controlled phase rotator 142, VO, V90, V180, and V270 embody the four-phase voltage signal, VS+ and VS− embody the two-phase voltage signal, and CP embodies the control code. In an embodiment, CP is a N-bit binary code, where N is an integer greater than 2, and a phase of VS+ is equal to one of 2N phases uniformly spaced between 0-degrees and 360-degrees. In an embodiment, N is 6. A digitally controlled phase rotor that can be used to embody the digitally controlled phase rotator 142 can be found in “A Wideband CMOS Linear Digital Phase Rotor” authored by Wang and Hajimiri and published in 2007 IEEE Custom Integrated Conference, San Jose, Calif.
By using the poly-phase filter 141 followed by the digitally controlled phase rotator, VS is proportional to VO with a phase delay that can be controlled by the phase control code CP.
A schematic diagram of an embodiment of the mixer 160 is shown in
A transimpedance amplifier (TIA) converts a current signal into a voltage signal. A schematic diagram of an embodiment of TIA 170 is shown in
In an embodiment, an analog-to-digital converter (not shown in figure) is used to digitize the mean voltage signal VM. This way, equation (11) can be evaluated numerically.
In an embodiment, a calibration process is performed to ensure the condition θ2=θ3 is met. A calibration process in accordance with an embodiment of the present disclosure is stated as follows. Set the amplitude and the frequency of the input voltage signal V to a target value (for instance, the amplitude is 100 mV, the frequency is 5 GHz). Use a power measuring instrument (for instance a power meter) of a purely resistive impedance (for instance, 50-Ohm) as a load (i.e. replacing the load 180 with the power measuring instrument). Read the power shown by the power measurement instrument. Sweep all the settings of the phase control code CP and identify a setting that leads to a largest value of VM. Find the power scaling factor G by calculating the ratio between the reading of the power from the power measuring instrument and the largest value of VM. The setting of the phase control code CP that leads to the largest value of VM and the power scaling factor G are recorded. This concludes the calibration at said target value of the input voltage signal VI. Note that with a load of purely resistive impedance (such as the power measuring instrument), the output voltage signal VO and the output current signal IO are in phase (i.e. θ1=0), and the largest value of VM occurs when θ3 equals θ2, per equation (10). Therefore, the setting of the phase control code CP that leads to the largest value of VM leads to the condition of θ2=θ3, which is the proper setting we need for using equation (11). Then, we can place any circuit as load 180, and use the value of VM times the power scaling factor G to find the power delivered to the circuit per equation (11).
In an embodiment, the calibration is performed at a plurality of frequencies (for instance, 5 GHz, 5.4 GHz, and 5.8 GHz). At each frequency, the calibration is performed at a plurality of amplitudes (for instance, 10 mV and 100 mV). For each combination of frequency and amplitude, the setting of CP that leads to the largest value of VM and the power scaling factor G are found and recorded. This way, a look up table can be established. When using the look up table, an entry that is closest to the amplitude and frequency of the input voltage signal V is used. For instance, if the amplitude is 80 mV and the frequency is 5.13 GHz, the entry pertaining to amplitude of 100 mV and frequency of 5 GHz in the lookup table will be used (because 80 mV is closer to 100 mV than to 10 mV, while 5.13 GHz is closer to 5 GHz than to 5.4 GHz and to 5.8 GHz). In an embodiment, a gain of inverting amplifiers 331, 332, 333, and 334 is adjusted in accordance with the amplitude of the input voltage signal V to be calibrated: the gain is adjusted to be higher for a lower amplitude of the input voltage signal VI. This way, an impairment due to a noise in the mixer 160 and the TIA 170 can be mitigated.
As illustrated by a flow diagram shown in
Those skilled in the art will readily observe that numerous modifications and alterations of the device and method may be made while retaining the teachings of the disclosure. Accordingly, the above disclosure should not be construed as limited only by the metes and bounds of the appended claims.
| Number | Name | Date | Kind |
|---|---|---|---|
| 7620371 | Zolfaghar | Nov 2009 | B2 |
| 20220018882 | Bellaouar | Jan 2022 | A1 |
| Number | Date | Country | |
|---|---|---|---|
| 20210217557 A1 | Jul 2021 | US |
