Patent Application
20120068677
The present invention relates in general to a power converter suitable for driving an inductive load at high frequencies. The present invention relates particularly to a driver for an inductively coupled gas-discharge lamp.
For driving an inductive load, a class-E amplifier has a suitable design, basically capable of operating at a high operating frequency with a high efficiency. Its basic design is shown in
The device has a resonance frequency determined by the series capacitor Cs1 and series inductor Ls1. In operation, the switch Q1 is switched ON and OFF with a duty cycle of 50%, wherein the switching frequency is close to the resonance frequency. With a good design of the components and their layout, the amplifier can have very low losses and thus a very high efficiency, up to over 80%. It is noted that a proper design at least includes Zero Voltage Switching and Zero Derivative Switching, as should be known to a person skilled in this art.
When high power is desirable, it is preferred to use a symmetrical push-pull design, as illustrated in
Q2; it typically includes the non-linear parasitic drain-source capacitance of the MOSFET Q2 as well as a parallel arranged external capacitor, having capacitance much higher than the said parasitic drain-source capacitance in order to reduce the influence of the non-linear parasitic drain-source capacitance as much as possible. The load 10 has its second terminal coupled to second node B with, for reasons of symmetry, a second series arrangement of a second series capacitance Cs2 and a second series inductance Ls2 connected between second node B and load 10.
In this standard design, the second switch Q2 is controlled with a 180° phase difference with respect to the first switch Q1.
In commercial lamp drivers, it is desirable to have cheap components. In normal, commercially available switches, the gate capacitance and resistance have a substantial value, which leads to a loss of power, which loss is proportional to the switching frequency. There is a general desire to increase the overall efficiency of electric and electronic apparatus as much as possible, in other words to reduce power losses as much as possible. Therefore, it is an objective of the present invention to improve the above-described class-E power amplifier design so as to provide a cost-effective converter potentially capable of having increased efficiency.
According to an important aspect, the present invention deviates from the design rule that the switching frequency should be close to the output frequency. In contrast, the present invention proposes that the output frequency is three times the switching frequency. As a result, the switching frequency is reduced when the output frequency is maintained, resulting in a reduction of the power loss in the switching stage.
Further advantageous elaborations are mentioned in the dependent claims.
These and other aspects, features and advantages of the present invention will be further explained by the following description of one or more preferred embodiments with reference to the drawings, in which same reference numerals indicate same or similar parts, and in which:
For driving an inductively coupled gas discharge lamp, a suitable output frequency with respect to on the one hand discharge efficiency and on the other hand EMI/EMC restrictions is 13.56 MHz.
Typically, in the case of a gas discharge lamp, the inductance of the load is high enough, so that Ls1 and Ls2 can be omitted. The resulting circuit 1 is shown in
A controller for controlling the switches Q1 and Q2 is indicated at 30. The controller 30 generates control signals for the switches Q1 and Q2 such that each switch is switched ON and OFF at a switching frequency that is a factor three lower than the desired output frequency, i.e. the switching frequency is equal to 4.52 MHz. This ON/OFF switching results in a current signal that can be approximated by a square wave signal, or, if the finite rise time and fall time of the signal edges for avoiding switching losses are taken into account, a trapezium wave signal, which involves a frequency spectrum having a large content of the third harmonic frequency.
It is noted that the circuit comprises two resonance circuits with two different resonance frequencies, assuming that the circuit is symmetrical. One resonance is the series resonance provided by the load 10 in conjunction with the series capacitors Cs1 and Cs2. The other resonance is the parallel resonance provided by the main inductor L1 and the drain capacitance Cds1 of the corresponding switch Q1. It is noted that a similar resonance is the parallel resonance provided by the second main inductor L2 and the drain capacitance Cds2 at the second switch Q2, but in the case of a good, symmetrical design of the circuit this second parallel resonance has the same parallel resonance frequency. It is possible, of course, to calculate the series resonance frequency and the parallel resonance frequency of the separate circuits. However, said resonance circuits influence each other, and it is difficult and not particularly enlightening, to give a general formula expressing the overall resonance frequency of the entire circuit 1 as a function of said two resonance frequencies, as should be clear to a person skilled in the art.
For obtaining an output frequency of 13.56 MHz, an intuitive approach might be to tune the series resonance circuit and the parallel resonance circuit to this frequency. However, the parallel resonance circuit has a very high (theoretically infinite) impedance at its resonance frequency, and if a series resonance circuit, tuned for the same frequency, is connected to this parallel resonance circuit, it does not filter anything. This is undesirable as it is intended that these circuits form a band-pass filter for the third harmonic of the switching frequency.
Inventor has found that a proper design of the circuit 1 involves a series resonance frequency close to (but not equal to) the desired output frequency of 13.56 MHz, and a parallel resonance frequency in the order of about 8 MHz.
Cs1 and Cs2 are adapted to set the series resonance frequency substantially equal to the desired value.
The inductances of the main inductor L1 [and L2] are chosen such that the parallel resonance frequency of the circuit L1/Cds1 [and L2/Cds2] is substantially equal to the desired value.
Inventor has found, by simulation and calculation, that, for the case of a 300 W inductive lamp having an inductance of 2.2 pH, optimum values for the several components are as follows:
L1=L2=550 nH
Cds1=Cds2=600 pF
Cs1=Cs2=66 pF
The inventor has found that a careful design of the main inductors L1 and L2 is important in achieving an efficiency as high as possible. This relates to the design of the individual inductors as well as to the physical arrangement of the inductors with respect to each other in the actual device.
When designing an individual inductor, important boundary conditions are the size of the inductor, low loss, and low external magnetic field, apart from the inductance value. In a test design suitable for a 300 W converter, it was calculated that the inductor should have an inductance of 550 nH. The inductor was designed to have 7 windings of thin copper strip in a toroid shape. The strip had a width of 6.35 mm and a thickness of 127 μm; this minimizes the losses due to the skin effect, considering that the skin depth was calculated to be about 31 μm. The core body of the coil, made from a non-ferro material so that the core can be considered to be an air core, had a cylindrical shape with an outer diameter of 50 mm, an inner diameter of 20 mm, and a height of 42 mm.
It is noted that small variations are possible; nevertheless, the described design is considered to be an optimal compromise. If the inner diameter is reduced, the distance between two adjacent windings in the inner space of the toroid is reduced so that the influence of proximity losses increases. If the inner diameter is increased, the surface area surrounded by each winding is reduced, which should be compensated by increasing the height and/or the outer diameter. Likewise, if the number of windings is increased, the distance between two adjacent windings in the inner space of the toroid is reduced, whereas, if the number of windings is decreased, the surface area should be increased. Likewise, if the width of the copper strip is increased, the influence of proximity losses increases, whereas, if the width of the copper strip is decreased, the Ohmic resistance of the inductor increases. Further, it is possible to use a magnetic material for the core body, but it was found that such coil designs would have higher internal losses.
Each inductor unavoidably has some stray magnetic field. For optimum operation, it is desirable that the magnetic coupling to the other circuit components is minimized as much as possible. In a practical device, however, the distance between the coils and the other components cannot be selected higher at will: the length of the connecting wires and the overall bulkiness of the device must remain within limits. In order to minimize the magnetic coupling to the other circuit components in the case of a push-pull converter, having two coils, the invention proposes that, when the two coils are arranged close together, special measures are taken.
The inventor has found that, for optimum operation, the exact inductance values of the coils are important. When the two coils are arranged close together, they have a mutual magnetic coupling, which will effectively change their inductance values.
In order to reduce these problems, the invention proposes that the layout of the circuit components in the practical realization is made as symmetric as possible.
Furthermore, the two coils are electrically connected in mutually opposite manner, as indicated by dots in
In the case of a single-ended design (see
It is noted that the improvement offered by this aspect of the present invention is not only offered at a phase difference of exactly 60°. The local maximum at 60° is relatively broad, so that for a certain range of phase differences around 60° the efficiency is actually higher than at 180°; for instance, at a phase difference of 54° the efficiency was still found to be 92.16%. Further, it is noted that the exact maximum is achieved at a phase difference slightly lower than 60°, but this is considered to be due to imperfections in the test circuit.
An embodiment within the scope of the present invention involves a phase difference equal to 180° or in a range close to 180°. From
Unexpectedly, the inventor has found that an embodiment with a phase difference equal to 60° or in a range close to 60° even has a better efficiency. For a range from about 45° to about 70°, the efficiency is at least higher than 90%, in other words the efficiency drops by about 2% if the phase difference is increased/decreased by about 20%.
The inventor has investigated the sensitivity of the circuit to changing component values, in other words the tolerance of the components.
For instance, in
Graphs B relate to the inductance of the load (i.e. lamp), the nominal optimal value being 2.2 μH. It can be seen that, for the phase difference being set equal to 180°, the available range for variation is wider as compared to the phase difference being set equal to 60°, as far as the limiting factor of the voltage drop over the switch is concerned: the load inductance may vary from about −8% to about +8% at 180° and may only vary from about −6% to about +6% at 60°. However, when the load inductance is varied from about −6% to about +6%, the efficiency at 60° is always higher than the efficiency at 180°.
Graphs C relate to the drain capacitances Cds1 and Cds2, the nominal optimal value being 600 pF. It can be seen that, for the phase difference being set equal to 180°, the voltage drop over the switch does not seem to provide any restrictions, whereas for the case of the phase difference being set equal to 60°, the capacitance should not vary by more than −10% or more than +30%. However, it can also be seen that the efficiency drops much faster in the case of the phase difference being set equal to 180° as compared to 60°. For the phase difference being set equal to 60°, the drain capacitances can be varied by about −10% to about +15% before the efficiency drops below 92%, whereas if, in the case of the phase difference being set equal to 180°, the drain capacitances are varied by about −10% or about +15%, the efficiency has dropped below 91.6%. For the entire range from about −10% to about +15%, the efficiency at 60° is always higher than the efficiency at 180°.
Graphs D relate to the series capacitances Cs1 and Cs2, the nominal optimal value being 66 pF. It can be seen that, for the phase difference being set equal to 180°, the available range for variation is wider as compared to the phase difference being set equal to 60°, as far as the limiting factor of the voltage drop over the switch is concerned: the load inductance may vary from about −8% to about +8% at 180° and may only vary from about −6% to about +6% at 60°. However, when the load inductance is varied from about −6% to about +6%, the efficiency at 60° is always higher than the efficiency at 180°.
Thus, the inventor has proven that, for a wide range of component tolerances, a phase difference of 60° provides for a better efficiency as compared to a phase difference of 60°.
Thus, the present invention provides a resonant power converter 1 for driving an inductive load, designed for operation at an operational frequency Fop of 13.56 MHz and comprising:
a series arrangement of a first inductor L1 and a first controllable switch Q1 connected to a DC voltage source DC;
a series arrangement of a second inductor L2 and a second controllable switch Q2 connected to said DC voltage source DC;
a first parallel capacitance Cds1 associated with the first controllable switch Q1;
a second parallel capacitance Cds2 associated with the second controllable switch Q2;
a controller 30 for driving the switches Q1, Q2;
the load is coupled between said nodes A, B;
the switches alternate between a conductive state and a non-conductive state at a duty cycle of 50%;
the switching frequency Fsw is one-third of said operational frequency Fop.
While the invention has been illustrated and described in detail in the drawings and foregoing description, it should be clear to a person skilled in the art that such illustration and description are to be considered illustrative or exemplary and not restrictive. The invention is not limited to the disclosed embodiments; rather, several variations and modifications are possible within the protective scope of the invention as defined in the appending claims.
Other variations to the disclosed embodiments can be understood and effected by those skilled in the art in practicing the claimed invention, from a study of the drawings, the disclosure, and the appended claims. In the claims, the word “comprising” does not exclude other elements or steps, and the indefinite article “a” or “an” does not exclude a plurality. A single processor or other unit may fulfill the functions of several items recited in the claims. The mere fact that certain measures are recited in mutually different dependent claims does not indicate that a combination of these measures cannot be used to advantage. Any reference signs in the claims should not be construed as limiting the scope.
In the above, the present invention has been explained with reference to block diagrams, which illustrate functional blocks of the device according to the present invention. It is to be understood that one or more of these functional blocks may be implemented in hardware, where the function of such a functional block is performed by individual hardware components, but it is also possible that one or more of these functional blocks are implemented in software, so that the function of such a functional block is performed by one or more program lines of a computer program or a programmable device such as a microprocessor, microcontroller, digital signal processor, etc.
| Number | Date | Country | Kind |
|---|---|---|---|
| 09160722.6 | May 2009 | EP | regional |
| Filing Document | Filing Date | Country | Kind | 371c Date |
|---|---|---|---|---|
| PCT/IB2010/052201 | 5/18/2010 | WO | 00 | 11/18/2011 |
