Patent Application
20060006957
The present invention relates to a radio frequency (RF) circuit component with multiple resonators. Such an RF circuit component can be used effectively as a filter or branch device for an RF signal processor in a communications system.
An RF circuit component, including a resonator as its basic element, is an essential component of an RF communications system. For example, a mobile communications system needs an RF circuit component functioning as a narrow band filter to utilize its frequency range effectively. Also, in the base station of a mobile communications system and a communication satellite, the development of a narrow-band, low-loss and small-sized filter that can withstand huge power has been long awaited.
Also, milliwave or quasi milliwave band wireless communications systems, developed remarkably these days, have used waveguide filters but desperately need such small-sized and low-loss filters, too.
Some of currently used RF circuit components such as resonator filters adopt a transmission line structure. An RF circuit component with a transmission line structure is small-sized and applicable up to radio frequencies falling within the microwave and milliwave ranges. Also, such an RF circuit component has a two-dimensional structure defined on a substrate, can be easily combined with other circuits and components, and is used extensively today.
As a typical example of a planar transmission line structure, an RF circuit component, which makes a disklike resonator exhibit a filtering characteristic by coupling a dipole mode thereto with protrusions provided for portions of its outer periphery, was reported in U.S. Pat. No. 5,172,084, for example.
The present inventor invented a multi-stage filter such as that shown in FIG. 7 and disclosed it in Japanese Laid-Open Patent Application Publication No. 2000-77905. This filter includes three elliptical conductors 2a, 2b and 2c, which are arranged in line, and two coupling terminals 6a and 6b coupled to the elliptical conductor 2a.
This filter can create an attenuation pole in a curve representing a filter characteristic. However, it is still difficult even for this filter to create the attenuation pole at a desired frequency and with a desired quantity of attenuation. This is because the frequency and quantity of attenuation of the attenuation pole need to be adjusted according to the specific combination of the degree of coupling between the elliptical conductors 2a, 2b and 2c, the filter characteristic and the quantity of filter loss.
Japanese Laid-Open Patent Application Publications No. 8-46413 and No. 10-308611 disclose RF circuit components including a disklike or elliptical conductor as its resonator. However, it is difficult for each of these RF circuit components to control the transmission characteristic with high precision, which is a common problem for them.
In order to overcome the problems described above, an object of the present invention is to provide an RF circuit component that achieves a desired frequency and a desired quantity of attenuation with a simple configuration.
An RF circuit component according to the present invention includes a substrate with a principal surface and a plurality of resonators, including a first resonator, a second resonator and a third resonator, which are arranged on the principal surface of the substrate so as to be coupled in series together. Each of the first, second and third resonators is made of a conductor supported on the substrate. The resonant modes of each of the first, second and third resonators include two fundamental resonant modes that oscillate perpendicularly to each other within a plane that is defined parallel to the principal surface of the substrate. The second resonator is arranged between the first and third resonators, and the oscillation direction of one of the fundamental resonant modes of the second resonator defines an angle greater than 0 degrees but smaller than 90 degrees with respect to that of its associated fundamental resonant mode of the first resonator and/or the third resonator.
In one preferred embodiment, the second resonator is made of a conductor, which has an elliptical cross section when taken parallel to the principal surface, and the oscillation directions of the two fundamental resonant modes of the second resonator are respectively parallel to the major axis and minor axis of the elliptical cross section.
In another preferred embodiment, each of the first and third resonators is made of a conductor, which has an elliptical cross section when taken parallel to the principal surface, and the oscillation directions of the two fundamental resonant modes in each of the first and third resonators are respectively parallel to the major axis and minor axis of the elliptical cross section.
In another preferred embodiment, the RF circuit component further includes an input coupling terminal for inputting an RF signal to one of the resonators and an output coupling terminal for outputting the RF signal from another one of the resonators.
In another preferred embodiment, each of the two resonators coupled to the input and output coupling terminals, respectively, is made of a conductor, which has the shape of an ellipse when taken parallel to the principal surface. The input coupling terminal is coupled to the resonator at a point away from an intersection between the major or minor axis of the ellipse and the ellipse itself, and the output coupling terminal is coupled to the resonator at a point away from an intersection between the major or minor axis of the ellipse and the ellipse itself.
In another preferred embodiment, the first resonator and the input coupling terminal are directly connected together and the third resonator and the output coupling terminal are directly connected together.
In another preferred embodiment, the RF circuit component further includes a metallic housing that is arranged so as to surround the substrate, and a screw is provided so as to go through the metallic housing.
In another preferred embodiment, the conductor is made of a superconductor material.
FIGS. 5(a), 5(b) and 5(c) are plan views showing various arrangements of resonators in RF circuit components according to the present invention.
Hereinafter, an RF circuit component according to a first preferred embodiment of the present invention will be described with reference to FIGS. 1(a) through 1(d).
As shown in FIGS. 1(a) and 1(b), the RF circuit component of this preferred embodiment includes a, substrate 1 with a principal surface and a first resonator 21, a second resonator 22, a third resonator 23 and a fourth resonator 24, which are arranged on the principal surface of the substrate 1 so as to be coupled in series together.
Each of these resonators 21, 22, 23 and 24 is an elliptical conductor pattern provided on the principal surface of the substrate 1. The resonant modes of each of these resonators 21, 22, 23 and 24 include two fundamental resonant modes (dipole modes), which oscillate perpendicularly to each other within a plane that is defined parallel to the principal surface of the substrate 1. In a circular or elliptical planar resonator, two of the fundamental resonant modes thereof, which have the lowest resonant frequency, will be referred to herein as “dipole modes”. The resonant modes of a circular planar resonator are sometimes identified in association with an electric field distribution in a propagation mode of a cylindrical waveguide (see J. Watkins, “Circular Resonant Structures in Microstrip”, Electron. Lett., 5, 21, p. 524 (1969)). According to such an association, the “dipole modes” in this specification may be called “TM11 modes”.
In the resonators 21, 22, 23 and 24 shown in
In a disklike resonator with a completely round shape, the two independent dipole modes have degeneracy and therefore have the same resonant frequency. In an elliptical resonator on the other hand, the two dipole modes no longer have the degeneracy and have two mutually different resonant frequencies that are defined by the major and minor axes of the ellipse, respectively. Accordingly, the elliptical resonator can utilize the two modes separately from each other, thereby functioning as two resonators with different resonant frequencies by itself.
In this preferred embodiment, the oscillation direction of the fundamental resonant mode of the first resonator 21 (as indicated by the arrow 50) is parallel to that of the fundamental resonant mode of the fourth resonator 24. However, the oscillation direction of one of the fundamental resonant modes of the second resonator 22 (as indicated by the arrow 51) defines an angle greater than 0 degrees but smaller than 90 degrees with respect to that of the fundamental resonant mode of the first resonator 21 (as indicated by the arrow 50). On the other hand, the oscillation direction of one of the fundamental resonant modes of the third resonator 23 is parallel to that of its associated fundamental resonant mode of the second resonator 22 (as indicated by the arrow 51) and defines an angle greater than 0 degrees but smaller than 90 degrees with respect to that of the fundamental resonant mode of the fourth resonator 24.
In this preferred embodiment, the structure of the resonators 21 through 24 is defined by providing a conductor pattern, made of a metal film (with a thickness of 0.1 μm to 10 μm, for example), on the principal surface of the substrate 1 as shown in
The substrate 1 is made of a dielectric material such as a ceramic and has dimensions of 15 mm×4 mm×1.5 mm, for example. In a preferred embodiment, the metal film is deposited on the principal surface of the substrate 1 by some thin film deposition technique such as vacuum evaporation. The shape and location of each conductor pattern may be arbitrarily defined by performing an etching process using a mask or a liftoff process.
The elliptical conductor patterns, functioning as the respective resonators 21, 22, 23 and 24, are arranged in series to each other with gaps 61, 62 and 63 provided between them, thereby forming a planar microwave transmission line.
To the first resonator 21, located at one end of the serial arrangement of the resonators 21, 22, 23 and 24, an input coupling terminal 31 is connected at an input coupling point 41. On the other hand, to the fourth resonator 24, located at the other end of the serial arrangement of the resonators 21, 22, 23 and 24, an output coupling terminal 32 is connected at an output coupling point 42. In this preferred embodiment, an RF signal (with a frequency of 15 GHz to 20 GHz, for example) is input through the input coupling terminal 31 and a filtered RF signal component is output through the output coupling terminal 32.
As shown in
The degree of coupling between the input coupling terminal 31 and the resonator 21 and between the output coupling terminal 32 and the resonator 24 is the highest when the angle a is equal to zero degrees. However, when this angle a is equal to ninety degrees, the degree of coupling becomes zero. Accordingly, by adjusting the angle a within the range of zero degrees to less than ninety degrees (i.e., 0°≦a<90°), a desired degree of coupling is achieved. In this manner, the degree of coupling can be controlled in a wide range by adjusting the angle a and therefore the circuit can be designed with much more freedom.
The RF signal that has been input through the input coupling terminal 31 to the first resonator 21 produces a resonance state in the first resonator 21. If the angle a is equal to zero degrees, this resonance state is defined by the dipole mode that oscillates (or is polarized) in the major-axis direction of the ellipse. On the other hand, if the angle a satisfies 0°<a<90°, that resonance state is defined by the superposition of the independent modes. More specifically, the resonance state can be represented by the superposition of the dipole mode polarized in the major-axis direction and the dipole mode polarized in the minor-axis direction. In the example illustrated in
In the layout illustrated in
As shown in
In the first resonator 21, the dipole mode polarized in the major-axis direction has a resonant frequency that depends on the major-axis diameter d1. In the same way, the dipole mode polarized in the minor-axis direction has a resonant frequency that depends on the minor-axis diameter d2. In this preferred embodiment, a filter that passes an RF signal, of which the center frequency is defined by the diameter d1, is realized. For that purpose, the other resonators 22, 23 and 24 are designed so as to have their diameter in the elliptical major-axis direction match the diameter d1.
In this manner, according to this preferred embodiment, only the dipole mode in the major-axis direction is used. Accordingly, the conductor patterns of the respective resonators 21, 22, 23 and 24 are defined to be elliptical, not completely round. In the following description, “1—(minor-axis length/major-axis length)” will be referred to herein as an “ellipticity”. Thus, if the “ellipticity” is equal to zero, that shape is round. For that reason, the elliptical conductor of each resonator of this preferred embodiment has an ellipticity greater than zero. According to the present invention, the ellipticity needs to be at least 0.01%, more preferably 1% or more. Optionally, the ellipticity may be set to even 10% or more.
In this manner, the ellipticity is set greater than zero such that the resonant frequency of the dipole mode in the minor-axis direction falls out of the frequency range to be used by the circuit (i.e., the “transmission band” in this preferred embodiment). That is to say, d1 is defined so as to produce resonance at a desired frequency as to the dipole mode in the major-axis direction, while d2 is defined so as to produce resonance at a frequency, which does not affect the operation of the circuit, as to the dipole mode in the minor-axis direction. Accordingly, the “ellipticity” is defined appropriately depending on how much the difference between the frequency of the dipole mode in the major-axis direction, or the resonant frequency (i.e., the center frequency of the transmission band), and the frequency of the attenuation pole to be described later should be.
As for coupling between the resonators, the degree of coupling between the dipole modes of two adjacent resonators can be adjusted by defining an appropriate gap for the gap portion 61, 62 or 63 and the degree of coupling with the dipole mode in the major-axis direction of the resonators 21 and 24 at both ends can be adjusted by the angle a. Consequently, by appropriately setting the angle a, major-axis diameter d1 and the gaps in the gap portions 61, 62 and 63, the RF circuit component with this structure operates as a four-stage resonator coupling filter.
In this preferred embodiment, the four resonators 21, 22, 23 and 24 are arranged in line in the direction L as described above. However, the major-axis direction of the second and third resonators 22 and 23 is defined so as to tilt by b degrees with respect to that of the first and fourth resonators 21 and 24, i.e., the direction L. In such an arrangement, the dipole mode in the major-axis direction of the first resonator 21 can also be coupled with the dipole mode 52 in the minor-axis direction of the second resonator 22 by adjusting the tilt angle b. In the same way, the dipole mode in the minor-axis direction of the third resonator 23 can also be barely coupled with the dipole modes in the major-axis directions of the other resonators 21, 22 and 24.
This angle b is formed by the polarization direction (i.e., oscillation direction) of the fundamental resonant mode of the RF component to be transmitted through two resonators to couple. This angle b is defined greater than 0 degrees and equal to or smaller than 45 degrees.
As a result of this mode coupling by the resonators, a signal having a frequency component corresponding to the resonant frequency of the dipole mode in the minor-axis direction is absorbed into the dipole mode in the minor-axis direction and an attenuation pole can be created at the frequency corresponding to the resonant frequency of the dipole mode in the minor-axis direction.
Hereinafter, a specific configuration according to this preferred embodiment will be described more fully.
In this preferred embodiment, a thin plate (with a thickness of 0.5 mm), made of a glass ceramic material (with a relative dielectric constant of 5.6 and an fQ value of 33,000) including an Al2O3—MgO—Gd2O3—SiO2 based ceramic filler and SiO2—Al2O3—B2O3—MgO-ZnO based glass, may be used as the substrate 1.
The elliptical patterns of the resonators were designed such that the resonators had a center frequency of GHz. More specifically, the major-axis diameter was set to around 3 mm, the minor-axis diameter was defined to be an appropriate ratio of 0.5 to 0.9 with respect to the major-axis diameter, and the input and output lines 3 had a line width of 0.8 mm. The conductor was made of a silver thin film with a thickness of 10 μm. The number and arrangement of the resonators were determined just as shown in
As can be seen from
As also shown in
To change the filter characteristic steeply by forming these attenuation poles, the ellipse major-axis direction of the second resonator 22 and/or the third resonator 23 needs to be rotated from that of the first resonator 22 and/or the fourth resonator 24. This is because by rotating the ellipse major axis in this manner, a resonant mode oscillating in the ellipse minor-axis direction is brought about.
According to this preferred embodiment, even though the same number of resonator stages are used, a steeper filter characteristic is realized due to the presence of those attenuation poles.
In the prior art, to form such attenuation poles, a skipped coupling arrangement was usually adopted to couple the resonators together. However, if such a skipped coupling arrangement were realized with the resonators for use in the present invention, then the dipole modes in the major-axis direction of the first and fourth resonators 21 and 24 should be barely coupled together directly. But such coupling is very hard to realize. What is worse, the frequencies to form the attenuation poles become very inaccurate. In contrast, according to the present invention, the attenuation poles can be formed with a simple structure. In addition, the frequencies to form the attenuation poles are determined by the minor-axis diameter d4 of the second and third resonators 22 and 23. Accordingly, the attenuation pole frequencies can be defined with high accuracy.
An RF circuit component such as that shown in
In the preferred embodiment described above, the first and fourth resonators 21 and 24 are formed in an elliptical shape so as to satisfy the inequality d1>d2. Alternatively, the resonators 21 and 24 may also be shaped so as to satisfy d1<d2 to the contrary. In that case, d1 is preferably defined so as to make the dipole mode in the minor-axis direction of the ellipse produce resonance at a desired frequency and d2 is preferably defined so as to make the dipole mode in the major-axis direction produce resonance at a sufficiently different frequency. Also, the respective axial lengths may be matched together such that the dipole mode in the minor-axis direction of a resonator couples with the dipole mode in the major-axis direction of its adjacent resonator.
Furthermore, since the dipole modes in the minor-axis direction are used in the second and third resonators 22 and 23, the attenuation pole is created in a frequency range that exceeds the pass band (in the vicinity of the resonant frequency) as shown in
FIGS. 5(a) through 5(c) are plan views showing alternative arrangements of resonators according to this preferred embodiment. In the example illustrated in
In the example illustrated in
In the example illustrated in
As described above, according to this preferred embodiment, the arrangements of respective resonators are combined, thereby achieving the target filter characteristic in various layouts and increasing the freedom of design significantly.
In the RF circuit component shown in
The conductor pattern of each resonator preferably has a smooth profile but may have an angular profile as well.
Hereinafter, an RF circuit component according to a second preferred embodiment of the present invention will be described with reference to
The substrate 1 and resonators 21, 22, 23 and 24 of this preferred embodiment have the same structures as the counterparts of the first preferred embodiment described above. However, unlike the first preferred embodiment, the RF circuit component of this preferred embodiment further includes a metallic housing 8 that surrounds the substrate 1.
A portion of the metallic housing 8 of this preferred embodiment, which is located over the upper surface of the substrate 1 (i.e., the side that the resonator 2 faces), includes a metallic screw 9 that extends through the metallic housing 8.
The electromagnetic field, created by the two dipole modes resonating in the resonator 21, 22, 23 or 24, partially leaks out of the resonator 21, 22, 23 or 24 upward. In this preferred embodiment, the resonant frequencies of the dipole modes are finely adjusted by utilizing such a leaking magnetic field. More specifically, the screw 9 is provided in the region where the leaking magnetic field is present and the top of that screw 9 is controlled, thereby finely adjusting the resonant frequencies of the dipole modes.
By adopting such a configuration, the circuit patterning precision can be relaxed and the yield at the manufacturing stage can be increased effectively.
Also, by surrounding the entire substrate 1 with the metallic housing 8, the electromagnetic waves, radiated from the resonators 21, 22, 23 and 24, can be cut off. As a result, the loss of the circuit can be reduced and the interference with another circuit can be avoided as well.
In this preferred embodiment, the metallic screw 9 is used. However, the screw 9 does not have to be a metallic screw. Alternatively, a screw made of a dielectric material or a metallic or dielectric bar may be provided over the resonators. Even so, the resonant frequencies can also be adjusted as effectively as the preferred embodiment just described. Optionally, the screws 9 may be provided in the respective gaps 61 and 62 between the two resonators such that the degree of coupling between the resonators can be adjusted.
It is even more effective to use a superconductor as the material of the conductor patterns functioning as the resonators of the present invention. Generally speaking, if a superconductor is used as the conductor material of a resonator, then the conductor loss becomes very small and the Q value of the resonator can be increased by leaps and bounds. However, in using a superconductor, when the maximum current density in the conductor exceeds a critical current density value, which is defined with respect to the radio frequency current of the superconductor material, the superconducting property is destroyed and the conductor cannot function as a resonator anymore. Nevertheless, the resonator of the present invention can have a decreased maximum current density as described above. Accordingly, by making the conductor of a superconductor, an RF signal with higher power can be processed as compared with a resonator with a conventional structure. As a result, a resonator that exhibits a high Q value even in response to an RF signal with high power is realized. Thus, significant effects are achieved.
In the preferred embodiments described above, the substrate is made of a glass ceramic material (with a relative dielectric constant of 5.6 and an fQ value of 33,000) including an Al2O3—MgO—Gd2O3—SiO2 based ceramic filler and SiO2—Al2O3—B2O3—MgO—ZnO based glass. However, the substrate materials that can be used effectively in the present invention are never limited to such materials. Alternatively, any other general dielectric material such as single crystalline dielectric materials and resin materials may be used, too. Nevertheless, to realize the low-loss and steep filter characteristic, a material with small dielectric loss needs to be used. Also, to decrease the size, a material with a high relative dielectric constant is preferably used.
The glass ceramic material including an Al2O3—MgO—Gd2O3—SiO2 based ceramic filler and SiO2—Al2O3—B2O3—MgO—ZnO based glass for use in the preferred embodiments described above is a material that has a relatively low dielectric constant and very small dielectric loss. Thus, this material can be used particularly effectively in milliwave or sub-milliwave band applications in which not so much small size as low loss is demanded strongly.
As can be seen easily, a material with a relative dielectric constant of less than 10 is particularly effective in a radio frequency range of 10 GHz or more. Conversely, in a frequency range of less than 10 GHz in which the downsizing demand is more dominating, a material with a relative dielectric constant of 10 or more such as Ba(Mg, Ta)O3-based ceramic material is preferred. Furthermore, the conductor material does not have to be silver or a superconductor as in the preferred embodiment described above. Alternatively, gold, copper, aluminum or any other suitable metal may be used almost as effectively although the resultant loss is different to a certain extent.
According to the present invention, an RF circuit component, which exhibits a steep filter characteristic by forming an attenuation pole with high precision, can be provided easily using planar resonators.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2003-091150 | Mar 2003 | JP | national |
| Filing Document | Filing Date | Country | Kind | 371c Date |
|---|---|---|---|---|
| PCT/JP04/02586 | 3/2/2004 | WO | 12/21/2004 |
