Patent Grant
7292121
Efficient combining of RF signals of the same or similar frequencies with varying relative phases and amplitudes cannot be achieved with such devices as Wilkinson or hybrid combiners, as they rely upon specific relative power levels and phases of the input signals to operate effectively and provide isolation between their inputs.
In the case of standard quadrature hybrid combiners (such as branch line couplers), the phase of the input signals must differ by 90 degrees with equal amplitudes for optimum combining efficiency. These criteria are required to ensure optimal voltage cancellation at specific nodes within the combining network, thus providing isolation and efficient operation.
An ultra high efficiency RF power amplifier could be realized by employing different biasing, device sizing and RF drive levels to the individual high power gain stages. However, the aforementioned combining schemes cannot provide efficient combining and isolation between the stages of such a power amplifier. Therefore, efficiency would be degraded, and portions of the RF energy from a single stage will be delivered to the other stages, effectively causing load impedance shifts, complicating the amplifier's behavior.
In the case of Wilkinson combiners, specific relative amplitudes and phases (usually zero) are required to minimize loss within the isolation resistor. Using such prior combining schemes, failure of a high power gain stage will result in a dramatic drop in efficiency. In the case where two stages are combined by a quadrature hybrid or Wilkinson combiner, at least half of the power of the stage remaining in operation will be lost, due to loss within the combining network.
Two or more reflective isolator elements are joined at their outputs to produce an RF combining structure. An RF circulator element is used within each reflective isolator element to provide a different phase delay according to the direction of propagation of the RF wave. The output of the reflective isolator elements exhibits a high impedance to signals propagating backwards into it, thus preventing propagation of signals from one input to another.
In the following description, reference is made to the accompanying drawings that form a part hereof, and in which is shown by way of illustration specific embodiments which may be practiced. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention, and it is to be understood that other embodiments may be utilized and that structural, logical and electrical changes may be made without departing from the scope of the present invention. The following description is, therefore, not to be taken in a limited sense, and the scope of the present invention is defined by the appended claims.
The parallel transmission line 215 may be thought of as arm one of the reflective isolator element, with arm two having two paths that contain the RF circulator element 210. The RF circulator element 210 is coupled to input node 225 by a line 227 having an impedance of ZR and length of ⅛λ The lines may be transmission lines or waveguides, or other types of lines that are capable of carrying RF signals. Line 227 is coupled to port 1 at 235 of circulator 240. The circulator 240 has three ports in one embodiment, and restricts RF signals to travel in one direction from port 1 at 235 to port 2 at 245, and from port 2 to port 3 at 250, and from port 3 to port 1. An output portion 255 of the RF circulator element 210 has an impedance of ZR and length of ⅛λ, and is then coupled to a line 260, having an impedance of ZR and length of ¼λ As can be seen, the phase shift in both arms is the same in the forward going direction. The rectangles in the figures are merely symbols used to represent length and impedance of the lines.
An RF wave entering the reflective isolator element from the output node 230 will split equally between the RF circulator element 210 and the parallel transmission line 215. However, the signals will be out of phase at node 225, and hence no RF current will be delivered to the RF input. Therefore, the impedance of the reflective isolator element 200, looking into its output port will be infinite.
This zeroing of backward traveling waves is accomplished via the path back through the RF circulator element 210 reaching port 2 at 245, and being directed toward port 3 at 250. Port 3 is coupled to an RF short 270 via a path 275 having an impedance of ZR and a length of ¼λ. The backward traveling wave is reflected at the RF short circuit 270 and travels back to the RF circulator element to port 1 at 235, and from there to input node 225. The total shift in phase of this reverse path is λ, while that of the reverse path in the parallel arm is ½λ. In one embodiment, the path length difference is an odd multiple of ½λ to obtain cancellation.
In one embodiment, the bandwidth of the circulator may be chosen to match the application—they are inherently narrow band (frequency) devices. Hence, the combining network will be frequency specific. The insertion phase of the circulator should be considered when designing the combiner (it may dictate the distance from port 3 to the RF short or open circuit, and the phase delays of the other transmission line elements). The insertion loss of the circulator may impact combining efficiency. Optimum combining efficiency may occur when the input signals are of the same frequency.
The Abstract is provided to comply with 37 C.F.R. §1.72(b) to allow the reader to quickly ascertain the nature and gist of the technical disclosure. The Abstract is submitted with the understanding that it will not be used to interpret or limit the scope or meaning of the claims.
| Number | Name | Date | Kind |
|---|---|---|---|
| 6741144 | Maeda et al. | May 2004 | B2 |
