Patent Grant
7917106
1. Field of the Invention
The present invention relates to a circuit for controlling RF PAs (Radio Frequency Power Amplifiers), and more specifically, to an RF PA controller circuit that controls the supply voltage of a PA using a closed amplitude control loop with an amplitude correction signal.
2. Description of the Related Art
RF (Radio Frequency) transmitters and RF power amplifiers are widely used in portable electronic devices such as cellular phones, laptop computers, and other electronic devices. RF transmitters and RF power amplifiers are used in these devices to amplify and transmit the RF signals remotely. RF PAs are one of the most significant sources of power consumption in these electronic devices, and their efficiency has a significant impact on the battery life on these portable electronic devices. For example, cellular telephone makers make great efforts to increase the efficiency of the RF PA systems, because the efficiency of the RF PAs is one of the most critical factors determining the battery life of the cellular telephone and its talk time.
The RF power amplifier 104 in general includes an output transistor (not shown) for its last amplification stage. When an RF modulated signal 106 is amplified by the RF PA 104, the output transistor tends to distort the RF modulated signal 106, resulting in a wider spectral occupancy at the output signal 110 than at the input signal 106. Since the RF spectrum is shared amongst users of the cellular telephone, a wide spectral occupancy is undesirable. Therefore, cellular telephone standards typically regulate the amount of acceptable distortion, thereby requiring that the output transistor fulfill high linearity requirements. In this regard, when the RF input signal 106 is amplitude-modulated, the output transistor of the PA 104 needs to be biased in such a way that it remains linear at the peak power transmitted. This typically results in power being wasted during the off-peak of the amplitude of the RF input signal 106, as the biasing remains fixed for the acceptable distortion at the peak power level.
Certain RF modulation techniques have evolved to require even more spectral efficiency, and thereby forcing the RF PA 104 to sacrifice more efficiency. For instance, while the efficiency at peak power of an output transistor of the PA 104 can be above 60%, when a modulation format such as WCDMA is used, with certain types of coding, the efficiency of the RF PA 104 falls to below 30%. This change in performance is due to the fact that the RF transistor(s) in the RF PA 104 is maintained at an almost fixed bias during the off-peak of the amplitude of the RF input signal 106.
Certain conventional techniques exist to provide efficiency gains in the RF PA 104. One conventional technique is EER (Envelope Elimination and Restoration). The EER technique applies the amplitude signal (not shown in
The conventional EER technique can function better only if a variable power supply with a very large variation range is used to adjust the supply voltage based on the amplitude signal of the RF input signal 106, while not reducing the efficiency of the RF transmitter by power consumed by the power supply itself. However, the variable power supply, which is typically comprised of a linear regulator (not shown in
Quite often, the conventional methods of controlling a PA fail to address the amplitude-to-phase re-modulation (AM-to-PM) which occurs in a non-frequency linear device such as a PA. Thus, the conventional methods are not suitable for the common types of PAs for use in common mobile telephony or mobile data systems because the required spectral occupancy performance is compromised by the AM to PM distortion.
Finally, PAs are typically used in conjunction with band pass filters that have a high electric coefficient of quality. These filters are typically of the SAW (surface acoustic wave) type. Due to their high coefficient of quality, the filters exhibit a relatively high group delay. The group delay makes it very difficult for a correction loop to work around the arrangement of the SAW filter and the PA while still meeting the high bandwidth requirements needed for the correction of the AM-to-PM.
Thus, there is a need for an RF PA system that is efficient over a wide variety of modulation techniques and results in a significant net decrease in power consumption by the RF PA system. There is also a need for a PA controller that can correct the AM to PM effects, while not relying on a PA specially designed for low AM to PM at the expense of efficiency. In addition, there is a need for a PA controller that can exclude the use of SAW filters from the path of the correction loop in the PA circuitry.
One embodiment of the present invention disclosed is a power amplifier controller circuit for controlling a power amplifier based upon an amplitude correction signal or amplitude error signal. The power amplifier receives and amplifies an input signal to the power amplifier and generates an output signal, and the power amplifier controller circuit controls the power amplifier so that it operates in an efficient manner.
The PA controller circuit comprises an amplitude control loop and a phase control loop. The amplitude control loop determines the amplitude correction signal (also referred to herein as the amplitude error signal), which is indicative of the amplitude difference between the amplitude of the input signal and the attenuated amplitude of the output signal, and adjusts the supply voltage to the power amplifier based upon the amplitude correction signal. The phase control loop determines a phase error signal, which indicates a phase difference between phases of the input signal and the output signal, and adjusts the phase of the input signal based upon the phase error signal to match the phase of the output signal. Thus, the phase control loop corrects for unwanted phase modulation introduced by the AM to PM non-ideality of the power amplifier and thus reduces phase distortion generated by the power amplifier.
In a first embodiment of the present invention, the amplitude control loop comprises an amplitude comparator comparing the amplitude of the input signal with an attenuated amplitude of the output signal to generate an amplitude correction signal, and a power supply coupled to receive the amplitude correction signal and generating the adjusted supply voltage provided to the power amplifier based upon the amplitude correction signal. The power supply can be a switched mode power supply. By using the amplitude correction signal to control the supply voltage to the power amplifier, a high-efficiency yet low-bandwidth power supply such as the switched mode power supply may be used to provide the adjusted supply voltage to the power amplifier.
In a second embodiment of the present invention, the amplitude correction signal is split into two or more signals with different frequency ranges and provided respectively to different types of power supplies with different levels of efficiency to generate the adjusted supply voltage provided to the power amplifier. For example, in the second embodiment, the power supplies include a first power supply with a first efficiency and a second power supply with a second efficiency higher than the first efficiency. The first power supply receives a first portion of the amplitude correction signal in a first frequency range and generates a first adjusted supply output based upon the first portion of the amplitude correction signal, and the second power supply receives a second portion of the amplitude correction signal in a second frequency range lower than the first frequency range and generates a second adjusted supply output based upon the second portion of the amplitude correction signal. The first and second adjusted supply outputs are combined to form the adjusted supply voltage provided to the power amplifier. The first power supply can be a linear regulator, and the second power supply can be a switched mode power supply. By dividing the amplitude correction signal into two or more signals with different frequency ranges, the second embodiment of the present invention has the additional advantage that the switched mode power supply may be implemented with even narrower bandwidth as compared to the first embodiment without significantly sacrificing efficiency. A narrower bandwidth power supply or a variable power supply with a smaller range of voltage variation is easier to implement.
In a third embodiment of the present invention, the amplitude control loop further comprises a gain control module receiving the amplitude correction signal to generate a gain control signal, and a variable gain amplifier adjusting the amplitude of the input signal according to the gain control signal. The third embodiment has the advantage that it is possible to operate the power amplifier at any given depth beyond its compression point, resulting in an extra degree of freedom in designing the PA circuit. This is useful in optimizing the efficiency gain versus spectral occupancy performance. By adding the variable gain amplifier, the amplitude of variation of the Vcc or bias voltage to the PA is further reduced, resulting in further significant efficiency gains.
In a fourth embodiment of the present invention, the phase control loop typically comprises one or more variable phase delays that introduce a relative phase delay in the phase control loop to allow the phase differences between the input and output signals of PA circuit to be within a range compatible with a phase comparator that generates the phase error signal. The phase control loop may also additionally comprise a low frequency blocking module such as a capacitor that removes the larger extent, lower frequency components of the phase error signal, so that the phase error signal is compatible with the phase shifter that adjusts the phase of the input signal based upon the phase error signal and is typically suitable for correcting smaller extent phase changes occurring at higher frequencies.
The features and advantages described in the specification are not all inclusive and, in particular, many additional features and advantages will be apparent to one of ordinary skill in the art in view of the drawings, specification, and claims. Moreover, it should be noted that the language used in the specification has been principally selected for readability and instructional purposes, and may not have been selected to delineate or circumscribe the inventive subject matter.
The teachings of the present invention can be readily understood by considering the following detailed description in conjunction with the accompanying drawings.
The Figures (FIG.) and the following description relate to preferred embodiments of the present invention by way of illustration only. It should be noted that from the following discussion, alternative embodiments of the structures and methods disclosed herein will be readily recognized as viable alternatives that may be employed without departing from the principles of the claimed invention.
Reference will now be made to several embodiments of the present invention(s), examples of which are illustrated in the accompanying figures. Wherever practicable similar or like reference numbers may be used in the figures and may indicate similar or like functionality. The figures depict embodiments of the present invention for purposes of illustration only. One skilled in the art will readily recognize from the following description that alternative embodiments of the structures and methods illustrated herein may be employed without departing from the principles of the invention described herein.
The PA controller circuit 202 may also adjust the phase and amplitude of the signal 204 to allow for power control and PA ramping, in accordance with information received through the configuration signals 209. Since the PA controller circuit 202 is aware of the voltage at the output and the current in the power amplifier 104, it can also adjust for load variations at an antenna (not shown herein) that may be used with the PA. If a directional coupler (not shown) is used to feed the attenuated amplitude of the signal 204, the PA controller 202 can adjust the forward power while controlling the PA operation point as it is aware of the voltage and current at node 208.
The phase control loop includes two limiters 312, 314, a phase comparator 316, a loop filter (PLF (Phase Loop Filter)) 318, and a phase shifter 320. To achieve stability over all conditions, the phase comparator 316 is of an adequate type with a capture range greater than 2*PI. To achieve this, a combination of adjustable delay elements and frequency dividers may be used. Also a phase sub-ranging system can be used since the dynamic phase variations that the phase correction loop processes are limited in amplitude. A sub-ranging phase control block (not shown) could be one of the constituents of the phase comparator 316 used with this system. Advantages of using sub-ranging in the phase comparator 316 are stability and good noise.
The amplitude control loop includes an adjusted variable attenuator (RFFA (RF Feedback Attenuator)) 306, two matched amplitude detectors 302, 304, a comparator 308, and a switched mode power supply (SMPS) 310. Note that the limiter 312 and the detector 302, and the limiter 314 and the detector 304, can be combined into a single limiter/power detector blocks without altering the functionality of the system.
Referring to
The function of the phase control loop is to counteract the AM (Amplitude Modulation) to PM (Phase Modulation) characteristics of the PA 104, which is part of the normal distortion characteristics of transistor-based amplifiers, allowing for the phase of the RF signal to be held constant at the output 110 of the PA 104 compared with the input 204 of the phase shifter 320 and thus reducing phase distortion generated by the PA 104. This phase control loop contributes to linearizing the PA 104 as the AM to PM phase shift of the PA 104 tends to become higher at higher power levels. By limiting the effects of AM to PM of the PA 104, the phase control loop allows the PA 104 to function at higher power levels with less distortion for the output signal 110, thus allowing the use of the PA 104 in more favorable efficiency conditions. In addition, the phase control loop also helps in correcting any additional AM to PM characteristics that the amplitude control loop (described below) may cause. While
Note that the phase control loop is of the error correction only type. In other words, the phase control loop does not modify the phase of the input signal 204 to the PA 104 unless the PA 104 or the amplitude control loop introduces a phase error. Since the noise contributions of the feedback loops affect the overall signal quality of the RF transmitter, an error correction only loop such as the phase control loop shown in
The amplitude control loop is also of the error correction only type, and thus is referred to herein as the amplitude correction loop. Thus, amplitude control loop and amplitude correction loop are used synonymously herein. Referring to
For a given output power, adjusting the supply voltage 208 of the PA 104 has the effect of varying its gain, as well as changing its efficiency. For a given output power, lowering the supply voltage 208 to the PA 104 provides better efficiency for the PA 104. The adjusted supply voltage 208 of the PA 104 is adjusted to ensure that the PA 104 stays in its most efficient amplification zone. Because adjusting the supply voltage 208 of the PA 104 does make a change to the gain of the PA 104, the output amplitude of the PA 104 changes with the supply voltage 208 from the SMPS 310, and the amplitude control loop can be closed. The principles of such operation can be explained as follows.
When the input to the PA 104 increases, the output of the PA 104 also increases. As the PA 104 stays in its linear region of operation, which corresponds to small input signals, its output will increase linearly with its input. Thus, both inputs to the comparator 308 will rise by the same amount, resulting in no error correction and no change to the supply voltage 208. This is the case when the output power is relatively small and well below the saturation point. As the input power continues to rise at the input of PA 104, there will be a point beyond which the output of the PA 104 will no longer be directly proportional with the input to the PA 104. The amplitude control loop will detect this error between the output and input of the PA 104, and raise the supply voltage to the PA 104 such that the initially-desired output power is delivered, resulting in linear operation of the system, even with a non-linear PA 104.
In a practical application, the PA 104 will be fully or partially saturated from its Vcc, for example, the highest 10 dB of its output power range, and as the RF modulation of the RF signal 204 forces the amplitude to vary, the amplitude control loop will only be actively controlling the supply voltage 208 to the PA 104 when the highest powers are required. For lower input power, the amplitude control loop will leave the supply voltage 208 at a fixed level because it detects no gain error, resulting in a fixed gain for the PA 104. The depth beyond compression can be adjusted by setting the level of the input signal 204 and the level of the attenuator 306, as well as the default supply voltage Vcc (not shown in
Varying the supply voltage to the PA 104 also results in a phase change. Thus, the phase control loop described above operates in conjunction with the amplitude control loop to maintain the accuracy of RF modulation at the output signal of the PA 104. Note that the phase control loop is also an error correction loop only, and therefore minimally contributes to noise.
Furthermore, the amplitude correction loop has the advantage that an SMPS 310, which does not consume any significant power by itself and thus actually increases the efficiency of the overall RF power amplifier system, can be used to generate the adjusted supply voltage 208 to the PA 104. This is possible because the adjusted supply voltage 208 to the PA 104 is generated by the SMPS 310 based upon the amplitude correction signal 309 which by nature has a much narrower range of variation or fluctuation rather than the actual amplitude of the RF input signal 204 which by nature has a much wider range of variation or fluctuation. An SMPS 310 is easier to implement to follow the amplitude correction signal 309 with a narrow range of variation, but would be more difficult to implement if it had to follow the unmodified amplitude of the RF input signal 204. This is related to the fact that the amplitude signal itself has its fastest variations when the amplitude itself is low. The amplitude correction loop does not need to make any changes to its output when the PA is operating in linear mode. For example, the amplitude correction signal 309 may be only active for the highest 10 dB of the actual output power variation. In contrast, the amplitude signal itself may vary by 40 dB, and varies much faster between −10 dBc to −40 dBc than it does between 0 dBc to −10 dBc. Thus the bandwidth requirements on the SMPS 310, which are coupled with the rate of change of the voltage, are reduced when an amplitude correction signal 309 rather than the amplitude signal itself is used to control the supply of the PA 104. The SMPS 310 does not consume any significant power by itself, and thus does not significantly contribute to usage of the battery power, and actually increases the efficiency of the RF power amplifier system. In contrast, a conventional polar modulation technique typically utilizes the amplitude signal itself to adjust the supply voltage to the PA 104, which prevents the use of an SMPS 310 for wideband RF signals because of the higher bandwidth requirements. Therefore, conventional RF power amplifier control systems typically use linear regulators (rather than an SMPS) to adjust the supply voltage to the PA 104. Such a linear regulator by itself consumes power resulting from its current multiplied by the voltage drop across the linear regulator. When there is a large drop in the amplitude signal, this can result in significant power being lost and results in none or little reduction in the overall battery power being consumed by the RF transmitter. This is because any efficiency gained in the RF PA is mostly lost in the linear regulator itself.
The amplitude correction signal 309 is split into the high frequency amplitude correction signal 401 and the low frequency amplitude correction signal 403 using the high pass filter 410 and the low pass filter 411, respectively. The high frequency amplitude correction signal 401 comprised of components of the amplitude correction signal 309 higher than a predetermined frequency and the low frequency amplitude correction signal 403 is comprised of components of the amplitude correction signal 309 lower than the predetermined frequency. The predetermined frequency used to split the amplitude correction signal 309 can be set at any frequency, but is preferably set at an optimum point where the efficiency of the overall RF transmitter system becomes sufficiently improved. For example, the predetermined frequency can be as low as 1/20th of the spectrally occupied bandwidth for the RF signal. In other embodiments, the predetermined frequency may not be fixed but may be adjusted dynamically to achieve optimum performance of the RF transmitter system.
Power consumed by the linear regulator 401 from a power source such as a battery (not shown) for a given control voltage 208 on the PA 104 can be approximated as follows:
Pbat≈Ipa×Vpa+Effl×(Vcc−Vpa)×Ipa
≈Effl×Vcc×Ipa
with Effl=1.05, which is sufficiently close to 1 to allow for this approximation, where Pbat is the power from the battery, Ipa is the input current to the PA 104, Vpa is the input supply voltage to the PA 104, and Vcc is the supply voltage of the battery. In addition, power consumed by the SMPS 404 from a power source such as a battery (not shown) for a given control voltage 208 on the PA 104 can be approximated as follows:
Pbat=Effs*Ipa*Vpa
with Effs=1.1,
and the efficiency of the switch (not shown) in the SMPS generally exceeding 90%.
If the average input voltage Vpa to the PA 104 is significantly lower than supply voltage Vcc of the battery, the SMPS 404 achieves much lower power consumption. While the linear regulator 402 is generally less efficient than the SMPS 404, the linear regulator 402 processing the high frequency part 401 of the amplitude correction signal 309 does not make the overall RF PA system inefficient in any significant way, because most of the energy of the amplitude correction signal 309 is contained in the low frequency part 403 rather than the high frequency part 401. This is explained below with reference to
Using both a high efficiency path comprised of the SMPS 404 carrying the low frequency portion 403 of the amplitude correction signal 309 and a low efficiency path comprised of the linear regulator 402 carrying the high frequency portion 401 of the amplitude correction signal 309 has the advantage that it is possible to use an SMPS 404 with a limited frequency response. In other words, the SMPS 404 need not accommodate for very high frequencies but just accommodates for a limited range of lower frequencies of the amplitude correction signal 309, making the SMPS 404 much easier and more cost-effective to implement. Combining the SMPS 404 with the linear regulator 402 enables high bandwidths of operation accommodating for full frequency ranges of the amplitude correction signal 309 without sacrificing the overall efficiency of the RF PA system in any significant way, since most of the energy of the amplitude correction signal 309 that is contained in the low frequency part 403 of the amplitude correction signal 309 is processed by the more efficient SMPS 404 rather than the less efficient linear regulator 402.
For example, Table 1 below illustrates the percentage of energy contained in the various frequency ranges in a hypothetical simple 4QAM (Quadrature Amplitude Modulation) signal used in WCDMA cellular telephones and the overall efficiency that can be expected to be achieved by the RF transmitter according to the embodiment of
Despite the extremely narrow bandwidth (100 KHz) of the SMPS 404 shown in the example of Table 1, 71% efficiency in the RF power amplifier supply system according to the embodiment of
More specifically, the gain control block 506 receives the amplitude correction signal 309 and adjusts the gain of the variable gain amplifier 502 based upon the amplitude correction signal 309, as well as passing the low frequency and high frequency parts 403, 401 of the amplitude correction signal 309 to the SMPS 404 and the linear regulator 402, respectively, to generate the adjusted supply voltage 208 as explained above with reference to
With the addition of the variable gain amplifier 502 and the gain control block 506, it is possible to use the PA 104 at any given depth beyond its compression point. The term “depth beyond compression” is used herein to refer to the difference between the averaged input compression level of the PA 104 and the actual averaged input power at the PA 104. For instance, when the peak output power is required, the input to the PA 104 can be overdriven by 10 dB beyond the 1 dB compression point of the PA 104. It is also possible to adjust the supply voltage of the PA 104 at the instant when the peak power is required, such that the 1 dB compression point is set higher and it is only necessary to overdrive the PA 104 input by 3 dB to obtain the same output peak power. A dynamic adjustment of both the input level and the supply voltage allows this loop system to reduce significantly further the amplitude of the control voltage 208.
In the embodiment of
In addition, the third embodiment of
The phase control loop illustrated in the embodiments of
The phase control loop shown in
Using the low frequency blocking module 1006 improves the performance of the phase control loop of the RF PA system of the fourth embodiment of
The low frequency blocking module 1006 has another benefit under some conditions. Some commonly used code-division multiple access cellular radio standards do not allow more than 30 degrees of phase discontinuity in the modulation when the level of the output power is going up and down according to the inner loop power control (base station). Note that the phase control loop may be configured to turn on and off as the PA output power is turned on or off (below a certain level), which could cause a phase glitch. However, the low frequency blocking module 1006 would prevent such phase glitch from occurring even when the phase control loop is turned on and off.
In one embodiment, the low frequency blocking module 1006 may be implemented using a capacitor. The value of the capacitance of the capacitor may be set to determine the frequency Fc (
Note that the phase comparator (also referred to herein as phase detector) 316 generally has a relatively wide operating range, for example, +/−90 degrees about a center point, which in this example may be 90 degrees. For most effective operation, it is desired that the RF PA system be configured so that at a desired transmission frequency of the RF signal under normal operating conditions the phase difference between the two RF inputs 324 and 325 to the phase comparator 316 is near the center point of the phase comparator 316 (in this example 90 degrees), which would result in a phase error signal 317 of approximately zero. The benefit to centering the operating point of the phase comparator 316 in this way is that the AM to PM distortion that occur during transmission leads to relative phase variations to the phase comparator input signals 1018 (324), 1022 (325) that remain within the operating range of the phase comparator 316. It is typically not possible to achieve a phase difference equal to exactly the center point of the phase comparator 316 (in this example 90 degrees) between the RF inputs 324 and 325 to the phase comparator 316, because this would require a phase difference close to 90 degrees between the RF input signal 204 and the RF output signal 110, which may not naturally occur. For RF PA systems designed for use only at a single frequency or a narrow band of frequencies, this can be handled by ensuring that the appropriate relative phase delays are present in the paths leading from the RF PA input 204 and the RF PA output 110 to the respective two inputs 324, 325 of the phase comparator 316. In the fourth embodiment of
Methods of setting the amount of relative phase delay to be introduced by the variable phase delays 1002, 1004 are explained below with reference to
Referring to
If the phase control loop is not locked in step 1208, then the process of adjusting the parameter VarDel, and thus the “delay” for the one or more variable phase delays 1002, 1004, can begin. First, the polarity of the phase comparator output 317 is checked 1211. If the polarity of the phase comparator output 317 indicates excessive delay (positive), VarDel is decremented 1212 by predetermined amount “Step_2” (e.g., 45 degrees). Similarly, if the polarity of the phase comparator output 317 indicates insufficient delay (negative), VarDel is incremented 1213 by the predetermined amount “Step_2” (e.g., 45 degrees). In this case, incrementing VarDel means increasing the value of “delay,” and decrementing VarDel means decreasing the value of “delay.” Then, another check is made 1214 to determine whether the phase control loop is apparently locked. Steps 1212 and 1213 are performed using the relatively large increment value Step_2 to determine generally in what range the appropriate phase delay for the phase control loop is. Thus, steps 1212 or 1213, and step 1214 are relatively coarse searching steps in search for the appropriate VarDel value. If the phase control loop is apparently locked in step 1214, the current value of VarDel is saved in step 1222 as the phase delay to use for the current output power level and center frequency. If the phase control loop is not apparently locked in step 1214, it is determined in step 1216 whether the phase comparator 316 flipped polarity, i.e., whether the phase error signal 317 has an opposite polarity relative to its value prior to the decrease or increase of Step_2 in step 1212 or 1213, respectively. If the polarity of the phase error signal 317 is determined in step 1216 to have flipped, this means that decrementing or incrementing Step_2 in step 1212 or step 1213 caused the phase control loop to overshoot the appropriate phase delay value that would have resulted in a locked condition. Thus, the appropriate VarDel value can be obtained by adjusting the VarDel value by a small amount smaller than Step_2 in a direction opposite to the direction of adjustment of Step_2. To accomplish this, VarDel is either incremented (if it was previously decremented in step 1212) or decremented (if it was previously incremented in step 1213) in steps of a predetermined value “Step_3” (e.g., 6 degrees) smaller than Step_2 until it is determined 1220 that the phase control loop is locked, at which point the corresponding value of VarDel is saved in step 1222 as the phase delay to use for the current output power level and center frequency. The value of “Step_3” is set to be small enough so that the phase control loop can achieve lock without overshoot as VarDel is stepped. As before, incrementing VarDel means increasing the value of “delay,” and decrementing VarDel means decreasing the value of “delay.” Note that steps 1218 and 1220 are relatively fine searching steps in search for the appropriate VarDel value. If the polarity of the phase error signal 317 did not flip in step 1216, the process goes back to step 1211 and step 1212 or step 1213 to adjust VarDel by Step_2 (e.g., 45 degrees) again, and the subsequent steps 1214, 1216, 1218, 1220, 1222 are repeated as necessary. In step 1222, after obtaining the appropriate value of phase delay to be introduced to the phase control loop of the RF PA system for the center frequency and the output power level, the variable phase delay(s) 1002, 1004 are set accordingly to introduce such relative phase delay in the phase control loop. For cases where the phase of the RF PA does not change substantially when operating at different output power levels, the calibration procedure of steps 1205 through 1222 can be performed at a single output power level. However, if the phase of the RF PA does change substantially when operating at difference output power levels, and thus there are more output power levels to set phase delays for (step 1223), the process returns to step 1205 to repeat steps 1205 through 1222 for a different output power level, and the appropriate settings for the variable phase delays 1002, 1004 are stored 1222 separately for each of the output power levels, at the currently set center frequency. Such settings are recalled in accordance with the output power level being used with the RF PA system. Similarly, for cases where the operating frequency range is narrow, for example, 1920-1980 MHz, the calibration procedure of steps 1204 through 1222 can be performed at a single center frequency and the process may end 1226. However, if the RF PA system must operate across a wide range of frequencies and thus there are more frequencies to set phase delays for (step 1224), the process returns to step 1204 to repeat steps 1204 through 1222 for a different center frequency.
In determining the appropriate settings for the one or more variable phase delay elements 1002, 1004, care must be taken to ensure that the phase control loop is not in an inverted condition, which is often referred to as a “false lock” condition, in which the phase comparator 316 would cause the phase shifter 320 to adjust in precisely the opposite of the appropriate direction. An inverted condition can arise because the phase comparator 316 generally operates with a limited range of phase difference at its inputs 324, 325. In this example, this range may be limited to approximately +/−90 degrees about a center point, which may be at 90 degrees. If the phase difference at 324, 325 is −90 degrees, which in this example is 180 degrees offset from the center point of 90 degrees, the phase comparator 316 can generate a zero signal at its output 317 indicating that the phase is locked when in fact the phase comparator 316 is in an inverted condition. Steps 1244, 1246, 1248, 1250 deal with ensuring that the phase control loop is not in an inverted condition. To accomplish this, VarDel is changed 1244 by a predetermined amount, Offset_Check. Offset_Check may be typically a moderate amount (e.g., 20 degrees), and VarDel may be changed in either direction (either incremented or decremented). Then, the behavior of the phase comparator 316 is observed to test for an expected polarity for a given polarity of Offset_Check. If the phase control loop was in an inverted condition with VarDel_Candidate applied, the polarity of the phase error signal 317 output from the phase comparator 316 would be at a polarity opposite to the expected polarity if the phase control loop was in a normal, non-inverted condition. Thus, step 1246 tests whether the value of VarDel_Candidate resulted in the phase control loop's proper locked condition, or an inverted condition. It is important that the magnitude of Offset_Check is large enough to ensure a reliable measurement of the polarity—e.g., substantially larger than noise levels in the circuit. If it is determined that the phase control loop is operating in an inverted condition in step 1246, VarDel is adjusted 1248 by another predetermined amount “Step_5” (e.g., 180 degrees) and the process goes back to step 1238 with this new VarDel value. The value of Step_5 may be set to be approximately 180 degrees since in this example an inverted condition occurs when the phase delay in the phase control loop is offset from a non-inverted condition by approximately 180 degrees. If it is determined that the phase control loop is operating in a proper non-inverted locked condition in step 1246, the value of VarDel_Candidate is stored 1250 as the final relative phase delay to be introduced to the phase control loop of the RF PA system for that output power level and center frequency, and the variable phase delay(s) 1002, 1004 are set accordingly to introduce such relative phase delay in the phase control loop. For cases where the phase of the RF PA does not change substantially when operating at different output power levels, the calibration procedure of steps 1235 through 1250 can be performed at a single output power level. However, if the phase of the RF PA does change substantially when operating at difference output power levels, and thus there are more output power levels to set phase delays for (step 1251), the process returns to step 1235 to repeat steps 1235 through 1250 for a different output power level, and the appropriate settings for the variable phase delays 1002, 1004 are stored 1250 separately for each of the output power levels, at the currently set center frequency. Such settings are recalled in accordance with the output power level being used with the RF PA system. Similarly, for cases where the operating frequency range is narrow, for example, 1920-1980 MHz, the calibration procedure of steps 1234 through 1250 can be performed at a single center frequency and the process may end 1254. If the RF PA system must operate across a wide range of frequencies and thus there are more frequencies to set phase delays for (step 1252), the process returns to step 1234 to repeat steps 1234 through 1250 for a different center frequency.
Upon reading this disclosure, those of skill in the art will appreciate still additional alternative structural and functional designs for the RF power amplifier controller through the disclosed principles of the present invention. For example, although the embodiment in
Thus, while particular embodiments and applications of the present invention have been illustrated and described, it is to be understood that the invention is not limited to the precise construction and components disclosed herein and that various modifications, changes and variations which will be apparent to those skilled in the art may be made in the arrangement, operation and details of the method and apparatus of the present invention disclosed herein without departing from the spirit and scope of the invention as defined in the appended claims.
This application claims priority under 35 U.S.C. §119(e) from co-pending U.S. Provisional Patent Application No. 60/764,947, entitled “RF Power Amplifier with Efficiency Improvement for High Peak to Average Modulation Types,” filed on Feb. 3, 2006; and this application is a continuation-in-part application of, and claims the benefit under 35 U.S.C. §120 from, co-pending U.S. patent application Ser. No. 11/429,119, entitled “Power Amplifier Controller Circuit,” filed on May 4, 2006, the subject matter of both of which are incorporated by reference herein in their entirety.
| Number | Name | Date | Kind |
|---|---|---|---|
| 3900823 | Sokal et al. | Aug 1975 | A |
| 4262264 | Vandegraaf | Apr 1981 | A |
| 4420723 | de Jager | Dec 1983 | A |
| 4591800 | Opas | May 1986 | A |
| 4754260 | Nelson et al. | Jun 1988 | A |
| 5023937 | Opas | Jun 1991 | A |
| 5087829 | Ishibashi et al. | Feb 1992 | A |
| 5128629 | Trinh | Jul 1992 | A |
| 5142240 | Isota et al. | Aug 1992 | A |
| 5305468 | Bruckert et al. | Apr 1994 | A |
| 5410276 | Hwang et al. | Apr 1995 | A |
| 5590408 | Weiland et al. | Dec 1996 | A |
| 5606285 | Wang et al. | Feb 1997 | A |
| 5675288 | Peyrotte et al. | Oct 1997 | A |
| 5732334 | Miyake | Mar 1998 | A |
| 5815531 | Dent | Sep 1998 | A |
| 5822442 | Agnew et al. | Oct 1998 | A |
| 5880633 | Leizerovich et al. | Mar 1999 | A |
| 5936464 | Grondahl | Aug 1999 | A |
| 5973556 | Su | Oct 1999 | A |
| 6002300 | Herbster et al. | Dec 1999 | A |
| 6031421 | McEwan | Feb 2000 | A |
| 6043707 | Budnik | Mar 2000 | A |
| 6133792 | Hansson | Oct 2000 | A |
| 6141541 | Midya et al. | Oct 2000 | A |
| 6166596 | Higashiyama et al. | Dec 2000 | A |
| 6166598 | Schlueter | Dec 2000 | A |
| 6175273 | Sigmon et al. | Jan 2001 | B1 |
| 6198347 | Sander et al. | Mar 2001 | B1 |
| 6208199 | Andersson | Mar 2001 | B1 |
| 6275685 | Wessel et al. | Aug 2001 | B1 |
| RE37407 | Eisenberg et al. | Oct 2001 | E |
| 6300826 | Mathe et al. | Oct 2001 | B1 |
| 6353359 | Leizerovich | Mar 2002 | B1 |
| 6377784 | McCune | Apr 2002 | B2 |
| 6404823 | Grange et al. | Jun 2002 | B1 |
| 6437641 | Bar-David | Aug 2002 | B1 |
| 6438360 | Alberth, Jr. et al. | Aug 2002 | B1 |
| 6445249 | Khan et al. | Sep 2002 | B1 |
| 6449465 | Gailus et al. | Sep 2002 | B1 |
| 6472934 | Pehlke | Oct 2002 | B1 |
| 6528975 | Sander | Mar 2003 | B2 |
| 6539072 | Donnelly et al. | Mar 2003 | B1 |
| 6583664 | Mathe et al. | Jun 2003 | B2 |
| 6633199 | Nielsen et al. | Oct 2003 | B2 |
| 6646501 | Wessel | Nov 2003 | B1 |
| 6661210 | Kimball et al. | Dec 2003 | B2 |
| 6734724 | Schell et al. | May 2004 | B1 |
| 6741127 | Sasho et al. | May 2004 | B2 |
| 6781452 | Cioffi et al. | Aug 2004 | B2 |
| 6825726 | French et al. | Nov 2004 | B2 |
| 6917244 | Rosnell et al. | Jul 2005 | B2 |
| 6924695 | Cioffi et al. | Aug 2005 | B2 |
| 6924700 | Taura et al. | Aug 2005 | B2 |
| 6924711 | Liu | Aug 2005 | B2 |
| 6928272 | Doi | Aug 2005 | B2 |
| 6968163 | Kuechler et al. | Nov 2005 | B2 |
| 7058373 | Grigore | Jun 2006 | B2 |
| 7068743 | Suzuki | Jun 2006 | B1 |
| 7109897 | Levesque | Sep 2006 | B1 |
| 7197286 | Ode et al. | Mar 2007 | B2 |
| 7250815 | Taylor et al. | Jul 2007 | B2 |
| 7359685 | Jafari et al. | Apr 2008 | B2 |
| 7440731 | Staudinger et al. | Oct 2008 | B2 |
| 7761065 | Drogi et al. | Jul 2010 | B2 |
| 20020053897 | Kajiwara et al. | May 2002 | A1 |
| 20020137481 | Chen et al. | Sep 2002 | A1 |
| 20020168949 | Johannisson et al. | Nov 2002 | A1 |
| 20020175764 | Matsuura et al. | Nov 2002 | A1 |
| 20030017840 | Katagishi et al. | Jan 2003 | A1 |
| 20030155978 | Pehlke | Aug 2003 | A1 |
| 20040071225 | Suzuki et al. | Apr 2004 | A1 |
| 20040162039 | Marque-Pucheu | Aug 2004 | A1 |
| 20040189378 | Suzuki et al. | Sep 2004 | A1 |
| 20040198257 | Takano et al. | Oct 2004 | A1 |
| 20040263254 | Tahara et al. | Dec 2004 | A1 |
| 20050046474 | Matsumoto et al. | Mar 2005 | A1 |
| 20050059362 | Kalajo et al. | Mar 2005 | A1 |
| 20050064830 | Grigore | Mar 2005 | A1 |
| 20050122163 | Chu | Jun 2005 | A1 |
| 20050156662 | Raghupathy et al. | Jul 2005 | A1 |
| 20050242880 | Domokos et al. | Nov 2005 | A1 |
| 20060001483 | Cioffi et al. | Jan 2006 | A1 |
| 20060040625 | Saito et al. | Feb 2006 | A1 |
| 20060232332 | Braithwaite | Oct 2006 | A1 |
| 20070096806 | Sorrells et al. | May 2007 | A1 |
| 20070184791 | Vinayak et al. | Aug 2007 | A1 |
| 20070184792 | Drogi et al. | Aug 2007 | A1 |
| 20070184793 | Drogi et al. | Aug 2007 | A1 |
| 20070184795 | Drogi et al. | Aug 2007 | A1 |
| 20070184796 | Drogi et al. | Aug 2007 | A1 |
| 20070218848 | Drogi et al. | Sep 2007 | A1 |
| 20070247253 | Carey et al. | Oct 2007 | A1 |
| Number | Date | Country |
|---|---|---|
| 0473299 | Mar 1992 | EP |
| 1225690 | Jul 2002 | EP |
| 0812064 | Sep 2003 | EP |
| 04-192907 | Jul 1992 | JP |
| 06-164249 | Jun 1994 | JP |
| 8-204774 | Aug 1996 | JP |
| 2000-507751 | Jun 2000 | JP |
| 3207153 | Jul 2001 | JP |
| 2001-519612 | Oct 2001 | JP |
| 2005-117315 | Apr 2005 | JP |
| WO 9534128 | Dec 1995 | WO |
| WO 9728598 | Aug 1997 | WO |
| WO 9918663 | Apr 1999 | WO |
| WO 9959243 | Nov 1999 | WO |
| WO 0016473 | Mar 2000 | WO |
| WO 0165685 | Sep 2001 | WO |
| WO 2005036739 | Apr 2005 | WO |
| WO 2005041438 | May 2005 | WO |
| WO 2005101678 | Oct 2005 | WO |
| Number | Date | Country | |
|---|---|---|---|
| 20070184794 A1 | Aug 2007 | US |
| Number | Date | Country | |
|---|---|---|---|
| 60764947 | Feb 2006 | US |
| Number | Date | Country | |
|---|---|---|---|
| Parent | 11429119 | May 2006 | US |
| Child | 11669648 | US |
