Patent Grant
8913974
1. Field of the Invention
The present invention relates generally to modulation and on-frequency power amplification. More particularly, the invention relates to methods and systems for vector combining power amplification.
2. Background Art
In power amplifiers, a complex tradeoff typically exists between linearity and power efficiency.
Linearity is determined by a power amplifier's operating range on a characteristic curve that relates its input to output variables—the more linear the operating range the more linear the power amplifier is said to be. Linearity is a desired characteristic of a power amplifier. In one aspect, for example, it is desired that a power amplifier uniformly amplifies signals of varying amplitude, and/or phase and/or frequency. Accordingly, linearity is an important determiner of the output signal quality of a power amplifier.
Power efficiency can be calculated using the relationship of the total power delivered to a load divided by the total power supplied to the amplifier. For an ideal amplifier, power efficiency is 100%. Typically, power amplifiers are divided into classes which determine the amplifier's maximum theoretical power efficiency. Power efficiency is clearly a desired characteristic of a power amplifier—particularly, in wireless communication systems where power consumption is significantly dominated by the power amplifier.
Unfortunately, the traditional tradeoff between linearity and efficiency in power amplifiers is such that the more linear a power amplifier is the less power efficient it is. For example, the most linear amplifier is biased for class A operation, which is the least efficient class of amplifiers. On the other hand, higher class amplifiers such as class B,C,D,E, etc, are more power efficient, but are considerably non-linear which can result in spectrally distorted output signals.
The tradeoff described above is further accentuated by typical wireless communication signals. Wireless communication signals, such as OFDM, CDMA, and W-CDMA for example, are generally characterized by their peak-to-average power ratios. The larger the signal's peak to average ratio the more non-linear distortion will be produced when non-linear amplifiers are employed.
Outphasing amplification techniques have been proposed for RF amplifier designs. In several aspects, however, existing outphasing techniques are deficient in satisfying complex signal amplification requirements, particularly as defined by wireless communication standards, for example.
In one aspect, existing outphasing techniques employ an isolating and/or a combining element when combining constant envelope constituents of a desired output signal. For example, it is commonly the case that a power combiner is used to combine the constituent signals. This combining approach, however, typically results in a degradation of output signal power due to insertion loss and limited bandwidth, and, correspondingly, a decrease in power efficiency.
In another aspect, the typically large size of combining elements precludes having them in monolithic amplifier designs.
What is needed therefore are power amplification methods and systems that solve the deficiencies of existing power amplifying techniques while maximizing power efficiency and minimizing non-linear distortion. Further, power amplification methods and systems that can be implemented without the limitations of traditional power combining circuitry and techniques are needed.
Embodiments for vector combining power amplification are disclosed herein.
In one embodiment, a plurality of substantially constant envelope signals are individually amplified, then combined to form a desired time-varying complex envelope signal. Phase and/or frequency characteristics of one or more of the signals are controlled to provide the desired phase, frequency, and/or amplitude characteristics of the desired time-varying complex envelope signal.
In another embodiment, a time-varying complex envelope signal is decomposed into a plurality of substantially constant envelope constituent signals. The constituent signals are amplified, and then re-combined to construct an amplified version of the original time-varying envelope signal.
Embodiments of the invention can be practiced with modulated carrier signals and with baseband information and clock signals. Embodiments of the invention also achieve frequency up-conversion. Accordingly, embodiments of the invention represent integrated solutions for frequency up-conversion, amplification, and modulation.
Embodiments of the invention can be implemented with analog and/or digital controls. The invention can be implemented with analog components or with a combination of analog components and digital components. In the latter embodiment, digital signal processing can be implemented in an existing baseband processor for added cost savings.
Additional features and advantages of the invention will be set forth in the description that follows. Yet further features and advantages will be apparent to a person skilled in the art based on the description set forth herein or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure and methods particularly pointed out in the written description and claims hereof as well as the appended drawings.
It is to be understood that both the foregoing summary and the following detailed description are exemplary and explanatory and are intended to provide further explanation of embodiments of the invention as claimed.
Embodiments of the present invention will be described with reference to the accompanying drawings, wherein generally like reference numbers indicate identical or functionally similar elements. Also generally, the leftmost digit(s) of the reference numbers identify the drawings in which the associated elements are first introduced.
The present invention will he described with reference to the accompanying drawings. The drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number.
Table of Contents
Methods, apparatuses and systems for vector combining power amplification are disclosed herein.
Vector combining power amplification is an approach for optimizing linearity and power efficiency simultaneously. Generally speaking, and referring to flowchart 502 in
Accordingly, vector combining power amplification allows for non-linear power amplifiers to be used to efficiently amplify complex signals whilst maintaining minimal non-linear distortion levels.
For purposes of convenience, and not limitation, methods and systems of the present invention are sometimes referred to herein as vector power amplification (VPA) methods and systems.
A high-level description of VPA methods and systems according to embodiments of the present invention is now provided. For the purpose of clarity, certain terms are first defined below. The definitions described in this section are provided for convenience purposes only, and are not limiting. The meaning of these terms will be apparent to persons skilled in the art(s) based on the entirety of the teachings provided herein. These terms may be discussed throughout the specification with additional detail.
The term signal envelope, when used herein, refers to an amplitude boundary within which a signal is contained as it fluctuates in the time domain. Quadrature-modulated signals can be described by r(t)=i(t)·cos(ωc·t)+q(t)·sin(ωc·t) where i(t) and q(t) represent in-phase and quadrature signals with the signal envelope e(t), being equal to e(t)=√{square root over (i(t)2+q(t)2)}{square root over (i(t)2+q(t)2)} and the phase angle associated with r(t) is related to arctan (q(t)/i(t).
The term constant envelope signal, when used herein, refers to in-phase and quadrature signals where e(t)=√{square root over (i(t)2+q(t)2)}{square root over (i(t)2+q(t)2)}, with e(t) having a relatively or substantially constant value.
The term time-varying envelope signal, when used herein, refers to a signal having a time-varying signal envelope. A time-varying envelope signal can be described in terms of in-phase and quadrature signals as e(t)=√{square root over (i(t)2+q(t)2)}{square root over (i(t)2+q(t)2)}, with e(t) having a time-varying value.
The term phase shifting, when used herein, refers to delaying or advancing the phase component of a time-varying or constant envelope signal relative to a reference phase.
1.1) Example Generation of Complex Envelope Time-Varying Input Signals
Time-varying complex signals may also be generated as illustrated in
1.2) Example Generation of Time-Varying Complex Envelope Signals from Constant Envelope Signals
The description in this section generally relates to the operation of step 508 in
In example 1 of
In example 2 of
In example 3 of
In summary, the examples of
It is noted that signals in the examples of
1.3) Vector Power Amplification Overview
A high-level overview of vector power amplification is now provided.
In the example of
in the example of
where output signal 178 is a power amplified version of input signal 172.
Linear (or substantially linear) power amplification of time-varying complex signals, as illustrated in
Referring to
Still referring to
In the example of
2. General Mathematical Overview
2.1) Phasor Signal Representation
Still referring to
r(t)=I(t)·cos(ωt)+Q(t)·sin(ωt)=R(t)·cos(φ(t))·cos(ωt)+R(t)·sin(φ(t))·sin(ωt) (1)
Note that in the example of
2.2) Time-Varying Complex Envelope Signals
It is further noted, from
In the example of
From
2.3) Constant Envelope Decomposition of Time-Varying Envelope Signals
Any phasor of time-varying magnitude and phase can be obtained by the sum of two or more constant magnitude phasors having appropriately specified phase shifts relative to a reference phasor.
For the purpose of illustration, three views are provided in
The example of
3. Vector Power Amplification Methods and Systems
Vector power amplification methods and systems according to embodiments of the present invention rely on the ability to decompose any time-varying envelope signal into two or more substantially constant envelope constituent signals or to receive or generate such constituent signals, amplify the constituent signals, and then sum the amplified signals to generate an amplified version of the time-varying complex envelope signal.
In, sections 3.1-3.3, vector power amplification (VPA) embodiments of the present invention are provided, including 4-branch and 2-branch embodiments. In the description, each VPA embodiment is first presented conceptually using a mathematical derivation of underlying concepts of the embodiment. An embodiment of a method of operation of the VPA embodiment is then presented, followed by various system level embodiments of the VPA embodiment.
Section 3.4 presents various embodiments of control modules according to embodiments of the present invention. Control modules according to embodiments of the present invention may be used to enable certain VPA embodiments of the present invention. In some embodiments, the control modules are intermediary between an input stage of the VPA embodiment and a subsequent vector modulation stage of the VPA embodiment.
Section 3.5 describes VPA output stage embodiments according to embodiments of the present invention. Output stage embodiments are directed to generating the output signal of a VPA embodiment.
Section 3.6 is directed to harmonic control according to embodiments of the present invention. Harmonic control may be implemented in certain embodiments of the present invention to manipulate the real and imaginary power in the harmonics of the VPA embodiment, thus increasing the power present in the fundamental frequency at the output.
Section 3.7 is directed to power control according to embodiments of the present invention. Power control may be implemented in certain embodiments of the present invention in order to satisfy power level requirements of applications where VPA embodiments of the present invention may be employed.
3.1) Cartesian 4-Branch Vector Power Amplifier
According to one embodiment of the invention, herein called the Cartesian 4-Branch VPA embodiment for ease of illustration and not limitation, a time-varying complex envelope signal is decomposed into 4 substantially constant envelope constituent signals. The constituent signals are equally or substantially equally amplified individually, and then summed to construct an amplified version of the original time-varying complex envelope signal.
It is noted that 4 branches are employed in this embodiment for purposes of illustration, and not limitation. The scope of the invention covers use of other numbers of branches, and implementation of such variations will be apparent to persons skilled in the art based on the teachings contained herein.
In one embodiment, a time-varying complex envelope signal is first decomposed into its in-phase and quadrature vector components. In phasor representation, the in-phase and quadrature vector components correspond to the signal's real part and imaginary part phasors, respectively.
As described above, magnitudes of the in-phase and quadrature vector components of a signal vary proportionally to the signal's magnitude, and are thus not constant envelope when the signal is a time-varying envelope signal. Accordingly, the 4-Branch VPA embodiment further decomposes each of the in-phase and quadrature vector components of the signal into four substantially constant envelope components, two for the in-phase and two for the quadrature signal components. This concept is illustrated in
In the example of
Still referring to
The phase shifts of phasors {right arrow over (IU
As an example, it can be further verified that, for the case illustrated in
and
in
wherein I1 and I2 represent the normalized magnitudes of phasors {right arrow over (I1)} and {right arrow over (I2)}, respectively, and wherein the domains of I1 and I2 are restricted appropriately according to the domain over which equation (2) and (3) are valid. It is noted that equations (2) and (3) are one representation for relating the relative phase shifts to the normalized magnitudes. Other, solutions, equivalent representations, and/or simplified representations of equations (2) and (3) may also be employed. Look up tables relating relative phase shifts to normalized magnitudes may also be used.
The concept describe above can be similarly applied to the imaginary phasor or the quadrature component part of a signal r(t) as illustrated in
It follows from the above discussion that, in phasor representation, any phasor {right arrow over (R)} of variable magnitude and phase can be constructed by the sum of four substantially constant magnitude phasor components:
{right arrow over (R)}={right arrow over (IU)}+{right arrow over (IL)}+{right arrow over (QU)}+{right arrow over (QL)};
{right arrow over (IU)}+{right arrow over (IL)}={right arrow over (I)};
{right arrow over (QU)}+{right arrow over (QL)}={right arrow over (Q)};
IU=IL=constant;
QU=QL=constant; (4)
where IU, IL, QU, and QL represent the magnitudes of phasors {right arrow over (IU)}, {right arrow over (IL)}, {right arrow over (QU)}, and {right arrow over (QL)}, respectively.
Correspondingly, in the time domain, a time-varying complex envelope sinusoidal signal r(t)=R(t)cos(ωt+φ) is constructed by the sum of four constant envelope signals as follows:
where sgn({right arrow over (I)})=±1 depending on whether {right arrow over (I)} is in-phase or 180° degrees out-of-phase with the positive real axis. Similarly, sgn({right arrow over (Q)})=±1 depending on whether {right arrow over (Q)} is in-phase or 180° degrees out-of-phase with the imaginary axis.
corresponds to the phase shift of {right arrow over (IU)} and {right arrow over (IL)} relative to the real axis. Similarly,
corresponds to the phase shift of {right arrow over (QU)} and {right arrow over (QL)} relative to the imaginary axis.
and
can be calculated using the equations given in (2) and (3).
Equations (5) can be farther simplified as:
r(t)=IU(t)+IL(t)+QU(t)+QL(t);
IU(t)=sgn({right arrow over (I)})×IUX×cos(ωt)+IUY×sin(ωt);
IL(t)=sgn({right arrow over (I)})×IUX×cos(ωt)−IUY×sin(ωt);
QU(t)=−QUX×cos(ωt)+sgn({right arrow over (Q)})×QUY×sin(ωt);
QL(t)=QUY×cos(ωt)−sgn({right arrow over (Q)})×QUY×sin(ωt). (6)
where
It can be understood by a person skilled in the art that, whereas the time domain representations in equations (5) and (6) have been provided for the case of a sinusoidal waveform, equivalent representations can be developed for non-sinusoidal waveforms using appropriate basis functions. Further, as understood by a person skilled in the art based on the teachings herein, the above-describe two-dimensional decomposition into substantially constant envelope signals can be extended appropriately into a multi-dimensional decomposition.
In the example of
Referring to
Still referring to
The actual implementation of each of vector modulators 520, 530, 540, and 550 may vary. It will be understood by a person skilled in the art, for example, that various techniques exist for generating the constant envelope constituents according to the equations in (6).
In the example embodiment of
In each vector modulator 520, 530, 540, 550, the in-phase and quadrature components are multiplied with amplitude information. In
To generate the IU(t) constant envelope constituent signals 525 and 527 are summed using phase splitter 528 or alternate summing techniques. The resulting signal 529 corresponds to the IU(t) component of signal r(t).
In similar fashion as described above, vector modulators 530, 540, and 55U, respectively, generate the IL(t), QU(t), and QL(t) components of signal r(t). IL(t), QU(t), and QL(t), respectively, correspond to signals 539, 549, and 559 in
Further, as described above, signals 529, 539, 549, and 559 are characterized by having substantially equal and constant magnitude envelopes. Accordingly, when signals 529, 539, 549, and 559 input into corresponding power amplifiers (PA) 562, 564, 566, and 568, corresponding amplified signals 563, 565, 567, and 569 are substantially constant envelope signals.
Power amplifiers 562, 564, 566, and 568 amplify each of the signals 529, 539, 549, 559, respectively. In an embodiment, substantially equal power amplification is applied to each, of the signals 529, 539, 549, and 559. In an embodiment, the power amplification level of PAs 562, 564, 566, and 568 is set according to the desired power level of output signal r(t).
Still referring to
Signals 573 and 575 are summed using summer 576, as shown in
It must be noted that, in the example of
Operation of the Cartesian 4-Branch VPA embodiment shall now be further described with reference to the process flowchart of
Step 620 includes receiving a clock signal set according to a desired output signal frequency of the desired output signal. In the example of
Step 630 includes processing the I component to generate first and second signals having the output signal frequency. The first and second signals have substantially constant and equal magnitude envelopes and a sum equal to the I component. The first and second signals correspond to the IU(t) and IL(t) constant envelope constituents described above. In the example of
Step 640 includes processing the Q component to generate third and fourth signals having the output signal frequency. The third and fourth signals have substantially constant and equal magnitude envelopes and a sum equal to the Q component. The third and fourth signals correspond to the QU(t) and QL(t) constant envelope constituents described above. In the example of
Step 650 includes individually amplifying each of the first, second, third, and fourth signals, and summing the amplified signals to generate the desired output signal. In an embodiment, the amplification of the first, second, third, and fourth signals is substantially equal and according to a desired power level of the desired output signal. In the example of
Vector power amplifier 700 includes an in-phase (I) branch 703 and a quadrature (Q) branch 705. Each of the I and Q branches further comprises a first branch and a second branch.
In-phase (I) information signal 702 is received by an I Data Transfer Function module 710. In an embodiment, I information signal 702 includes a digital baseband signal. In an embodiment, I Data Transfer Function module 710 samples I information signal 702 according to a sample clock 706. In another embodiment, I information signal 702 includes an analog baseband signal, which is converted to digital using an analog-to-digital converter (ADC) (not shown in
I Data Transfer Function module 710 processes I information signal 702, and determines in-phase and quadrature amplitude information of at least two constant envelope constituent signals of I information signal 702. As described above with reference to
I Data Transfer Function module 710 outputs information signals 722 and 724 used to control the in-phase and quadrature amplitude components of vector modulators 760 and 762. In an embodiment, signals 722 and 724 are digital signals. Accordingly, each of signals 722 and 724 is fed into a corresponding digital-to-analog converter (DAC) 730 and 732, respectively. The resolution and sample rate of DACs 730 and 732 is selected to achieve the desired component of the output signal 782. DACs 730 and 732 are controlled by DAC clock signals 723 and 725, respectively. DAC clock signals 723 and 725 may be derived from a same clock signal or may be independent.
In another embodiment, signals 722 and 724 are analog signals, and DACs 730 and 732 are not required.
In the exemplary embodiment of
Vector modulators 760 and 762 multiply signals 740-742 with appropriately phased clock signals to generate constant envelope constituents of I information signal 702. The clock signals are derived from a channel clock signal 708 having a rate according to a desired output signal frequency. A plurality of phase splitters, such as 750 and 752, for example, and phasors associated with the vector modulator multipliers may be used to generate the appropriately phased clock signals.
In the embodiment of
In parallel and in similar fashion, the Q branch of vector power amplifier 700 generates at least two constant envelope constituent signals of quadrature (Q) information, signal 704.
In the embodiment of
As described above with respect to
Circuitry 714 and 716 (herein referred to as “autobias circuitry” for ease of reference, and not limitation) and in this embodiment, control the bias of PAs 770, 772, 774, and 776 according to I and Q information signals 702 and 704. In the embodiment of
In an embodiment, PAs 770, 772, 774, and 776 apply substantially equal power amplification to respective substantially constant envelope signals 761, 763, 765, and 767. In other embodiments, PA drivers are additionally employed to provide additional power amplification. In the embodiment of
The outputs of PAs 770, 772, 774, and 776 are coupled together to generate output signal 782 of vector power amplifier 700. In an embodiment, the outputs of PAs 770, 772, 774, and 776 are directly coupled together using a wire. Direct coupling in this manner means that there is minimal or no resistive, inductive, or capacitive isolation between the outputs of PAs 770, 772, 774, and 776. In other words, outputs of PAs 770, 772, 774, and 776, are coupled together without intervening components. Alternatively, in an embodiment, the outputs of PAs 770, 772, 774, and 776 are coupled together indirectly through inductances and/or capacitances that result in low or minimal impedance connections, and/or connections that result in minimal isolation and minimal power loss. Alternatively, outputs of PAs 770, 772, 774, and 776 are coupled using well known combining techniques, such as Wilkinson, hybrid, transformers, or known active combiners. In an embodiment, the PAs 770, 772, 774, and 776 provide integrated amplification and power combining in a single operation. In an embodiment, one or more of the power amplifiers and/or drivers described herein are implemented using multiple input, single output power amplification techniques, examples of which are shown in
Output signal 782 includes the I and Q characteristics of I and Q information signals 702 and 704. Further, output signal 782 is of the same frequency as that of its constituents, and thus is of the desired up-converted output frequency. In embodiments of vector power amplifier 700, a pull-up impedance 780 is coupled between the output of vector amplifier 700 and a power supply. Output stage embodiments according to power amplification methods and systems of the present invention will be further described below in section 3.5.
In other embodiments of vector power amplifier 700, process detectors are employed to compensate for any process variations in circuitry of the amplifier. In the embodiment of
The embodiment illustrates a multiple-input single-output (MISO) implementation of the amplifier of
In the embodiment of
DAC 830 receives in-phase and quadrature information signals 810 and 820 from I Data Transfer Function module 710 and Q Data Transfer Function module 712, respectively, as described above. In an embodiment, a input selector 822 selects the order of signals 810 and 820 being input into DAC 830.
DAC 830 may output a single analog signal at a time. In an embodiment, a sample and hold architecture may be used to ensure proper signal timing to the four branches of the amplifier, as shown in
DAC 830 sequentially outputs analog signals 832, 834, 836, 838 to a first set of sample-and-hold circuits 842, 844, 846, and 848. In an embodiment, DAC 830 is clocked at a sufficient rate to emulate the operation of DACs 730, 732, 734, and 736 of the embodiment of
DAC 830's DAC clock signal 826, output selector signal 824, input selector 822, and sample-and-hold clocks 840A-D, and 850 are controlled by a control module that can be independent or integrated into transfer function modules 710 and/or 712.
In an embodiment, sample-and-hold circuits (S/H) 842, 844, 846, and 848 sample and hold the received analog values from DAC 830 according to a clock signals 840A-D. Sample-and-hold circuits 852, 854, 856, and 858 sample and hold the analog values from sample and hold circuits 842, 844, 846, and 848 respectively. In turn, sample-and-hold circuits 852, 854, 856, and 858 hold the received analog values, and simultaneously release the values to vector modulators 760, 762, 764, and 766 according to a common clock signal 850. In another embodiment, sample-and-hold circuits 852, 854, 856, and 858 release the values to optional interpolation filters 731, 733, 735, and 737 which are also anti-aliasing filters. In an embodiment, a common clock signal 850 is used in order to ensure that the outputs of S/H 852, 854, 856, and 858 are time-aligned.
Other aspects of vector power amplifier 800A substantially correspond to those described above with respect to vector power amplifier 700.
Embodiment 800B illustrates another single DAC implementation of the vector power amplifier. However, in contrast to the embodiment of
Other aspects of vector power amplifier 800B substantially correspond to those described above with respect to vector power amplifiers 700 and 800A.
Other aspects of vector power amplifier 800C substantially correspond to those described above with respect to vector power amplifiers 700 and 800A.
Other aspects of vector power amplifier 800D substantially correspond to those described above with respect to vector power amplifiers 700 and 800B.
3.2) Cartesian-Polar-Cartesian-Polar 2-Branch Vector Power Amplifier
A Cartesian-Polar-Cartesian-Polar (CPCP) 2-Branch VPA embodiment shall now be described (The name of this embodiment is provided for ease of reference, and is not limiting).
According, to the Cartesian-Polar-Cartesian-Polar (CPCP) 2-Branch VPA method, a time-varying complex envelope signal is decomposed into 2 substantially constant envelope constituent signals. The constituent signals are individually amplified, and then summed to construct an amplified version of the original time-varying complex envelope signal. In addition, the phase angle of the time-varying complex envelope signal is determined and the resulting summation of the constituent signals are phase shifted by the appropriate angle.
In one embodiment of the CPCP 2-Branch VPA method, a magnitude and a phase angle of a time-varying complex envelope signal are calculated from in-phase and quadrature components of a signal. Given the magnitude information, two substantially constant envelope constituents are calculated from a normalized version of the desired, time-varying envelope signal, wherein the normalization includes implementation specific manipulation of phase and/or amplitude. The two substantially constant envelope constituents are then phase shifted by an appropriate angle related to the phase shift of the desired time-varying envelope signal. The substantially constant envelope constituents are then individually amplified substantially equally, and summed to generate an amplified version of the original desired time-varying envelope signal.
Referring to
Still referring to
The phase shifts of phasors {right arrow over (U′)} and {right arrow over (L′)} relative to {right arrow over (R′)} are set according to the desired magnitude R of {right arrow over (R′)}. In the simplest case, when upper and lower phasors {right arrow over (U′)} and {right arrow over (L′)} are selected to have equal magnitude, upper and lower phasors {right arrow over (U′)} and {right arrow over (L′)} are substantially symmetrically shifted in phase relative to {right arrow over (R′)}. This is illustrated in the example of
It can be verified that, for the case illustrated in
in
where R represents a normalized magnitude of phasor {right arrow over (R′)}.
Equation (7) can further be reduced to
where R represents a normalized magnitude of phasor {right arrow over (R′)}.
Alternatively, any substantially equivalent mathematical equations or other substantially equivalent mathematical techniques such as look up tables can be used.
It follows from the above discussion that, in phasor representation, any phasor {right arrow over (R′)} of variable magnitude and phase can be constructed by the sum of two constant magnitude phasor components:
{right arrow over (R′)}={right arrow over (U′)}+{right arrow over (L′)}
|{right arrow over (U)}|={right arrow over (L)}|=A=constant (8)
Correspondingly, in the time domain, a time-varying envelope sinusoidal signal r′(t)=R(t)×cos(ωt) is constructed by the sum of two constant envelope signals as follows:
where A is a constant and
is as shown in equation (7).
From
r′(t)=U′(t)+L′(t);
U′(t)=C cos(ωt)+α sin(ωt);
L′(t)=C cos(ωt)−β sin(ωt); (10)
where C denotes the real part component of phasors {right arrow over (U′)} and {right arrow over (L′)} and is equal to
Note that C is a common component of {right arrow over (U′)} and {right arrow over (L′)}, α and β denote the imaginary part components of phasors {right arrow over (U′)} and {right arrow over (L′)}, respectively.
Accordingly, from equations (12),
As understood by a person skilled in the art based on the teachings herein, other equivalent and/or simplified representations of the above representations of the quantities A, B, and C may also be used, including look up tables, for example.
Note that {right arrow over (Rin)} is shifted by θ degrees relative to {right arrow over (R′)} Accordingly, using equations (8), it can be deduced that:
{right arrow over (Rin)}={right arrow over (R′)}ejθ=({right arrow over (U′)}+{right arrow over (L′)})ejθ={right arrow over (U′)}ejθ+{right arrow over (L′)}ejθ (11)
Equations (11) imply that a representation of {right arrow over (Rin)} be obtained by slimming phasors {right arrow over (U′)} and {right arrow over (L′)}, described above, shifted by θ degrees. Further, an amplified output version, {right arrow over (Rout)}, of {right arrow over (Rin)} can be obtained by separately amplifying substantially equally each of the θ degrees shifted versions of phasors {right arrow over (U′)} and {right arrow over (L′)}, and summing them.
Equivalently, in the time domain, it can be shown that:
rout(t)=U(t)+L(t);
U(t)=K[C cos(ωt+θ)+α sin(ωt+θ)];
L(t)=K[C cos(ωt+θ)−β sin(ωt+θ)]. (12)
where rout (t) corresponds to the time domain signal represented by phasor {right arrow over (Rout)}, U(t) and L(t) correspond to the time domain signals represents by phasors {right arrow over (U)} and {right arrow over (L)}, and K is the power amplification factor.
A person skilled in the art will appreciate that, whereas the time domain representations in equations (9) and (10) have been provided for the case of a sinusoidal waveform, equivalent representations can be developed for non-sinusoidal waveforms using appropriate basis functions.
In the example of
Referring to
Still referring to
Still referring to
Output signals 1040 and 1042 of respective vector modulators 1060 and 1062 correspond, respectively, to the U(t) and L(t) constant envelope constituents of input signal r(t).
As described above, signals 1040 and 1042 are characterized by having substantially equal and constant magnitude envelopes. Accordingly, when signals 1040 and 1042 are input into corresponding power amplifiers (PA) 1044 and 1046, corresponding amplified signals 1048 and 1050 are substantially constant envelope signals.
Power amplifiers 1044 and 1046 apply substantially equal power amplification to signals 1040 and 1042, respectively. In an embodiment, the power amplification level of PAs 1044 and 1046 is set according to the desired power level of output signal r(t). Further, amplified signals 1048 and 1050 are in-phase relative to each other. Accordingly, when summed together, as shown in
In embodiment 1000A, constant envelope signals 1040 and 1042, output from vector modulators 1060 and 1062, are input into MISO PA 1054. MISO PA 1054 is a two-input single-output power amplifier. In an embodiment, MISO PA 1054 may include various elements, such as pre-drivers, drivers, power amplifiers, and process detectors (not shown in
Operation of the CPCP 2-Branch VPA embodiment is depicted in the process flowchart 1100 of
The process begins at step 1110, which includes receiving a baseband representation of the desired output signal. In an embodiment, this involves receiving in-phase (I) and quadrature (Q) components of the desired output signal. In another embodiment, this involves receiving magnitude and phase of the desired output signal.
Step 1120 includes receiving a clock signal set according to a desired output signal frequency of the desired output signal. In the example of
Step 1130 includes processing the clock signal to generate a normalized clock signal having a phase shift angle according to the received I and Q components. In an embodiment, the normalized clock signal is a constant envelope signal having a phase shift angle according to a ratio of the I and Q components. The phase shift angle of the normalized clock is relative to the original clock signal. In the example of
Step 1140 includes the processing of the I and Q components to generate the amplitude information required to produce first and second substantially constant envelope constituent signals.
Step 1150 includes processing the amplitude information of step 1140 and the normalized clock signal Rclk to generate the first and second constant envelope constituents of the desired output signal. In an embodiment, step 1150 involves phase shifting the first and second constant envelope constituents of the desired output signal by the phase shift angle of the normalized clock signal. In the example of
Step 1160 includes individually amplifying the first and second constant envelope constituents, and summing the amplified signals to generate the desired output signal. In an embodiment, the amplification of the first and second constant envelope constituents is substantially equal and according to a desired power level of the desired output signal. In the example of
Referring to
In an embodiment, I and Q Data Transfer Function module 1216 processes information signal 1210 to generate information signals 1220, 1222, 1224, and 1226. The operation of I and Q Data Transfer Function module 1216 is further described below in section 3.4.
Referring to
Still referring to
where θ represents the phase of the desired: output signal, represented b
phasor {right arrow over (Rout)} in
In the exemplary embodiment of
In other embodiments, information signals 1220, 1222, 1224, and 1226 are generated in analog, format and, no DACs are required.
Referring to
Still referring to
Vector modulator 1238 combines the two modulated signals to generate Rclk signal 1250. Rclk signal 1250 is a substantially constant envelope signal having the desired output frequency and a phase shift angle according to the I and Q data included in signal 1210.
Still referring to
Vector modulator 1260 combines signal 1240, multiplied with a 90° shifted version of Rclk signal 1250, and signal 1248, multiplied with a 0° shifted version of Rclk signal 1250, to generate output signal 1264. In parallel, vector modulator 1262 combines signal 1242 multiplied with a 90° shifted version of Rclk signal 1250, and signal 1248, modulated with a 0° shifted version of Rclk signal 1250, to generate output signal 1266.
Output signals 1264 and 1266 represent substantially constant envelope signals. Further, phase shifts of output signals 1264 and 1266 relative to Rclk signal 1250 are determined by the angle relationships associated with the ratios α/C and β/C, respectively. In an embodiment, α=−β and therefore output signals 1264 and 1266 are symmetrically phased relative to Rclk signal 1250. With reference to
A sum of output signals 1264 and 1266 results in a channel-clock-modulated signal having the I and Q characteristics of baseband signal r(t). To achieve a desired power level at the output of vector power amplifier 1200, however, signals 1264 and 1266 are amplified to generate an amplified output signal. In the embodiment of
In an embodiment, PAs 1270 and 1272 apply substantially equal power amplification to respective constant envelope signals 1264-1266. In an embodiment, the power amplification is set according to the desired output power level. In other embodiments of vector power amplifier 1200, PA drivers and/or pre-drivers are additionally employed to provide additional power amplification capability to the amplifier. In the embodiment of
Respective output signals 1274 and 1276 of PAs 1270 and 1272 are substantially constant envelope signals. Further, when output signals 1274 and 1276 are summed, the resulting signal has minimal non-linear distortion. In the embodiment of
Output signal 1280 represents a signal having, the I and Q characteristics of baseband signal r(t) and the desired output power level and frequency. In embodiments of vector power amplifier 1200, a pull-up impedance 1288 is coupled between the output of vector power amplifier 1200 and a power supply. In other embodiments, an impedance matching network 1290 is coupled at the output of vector power amplifier 1200. Output stage embodiments according to power amplification methods and systems of the present invention will be further described below in section 3.5.
In other embodiments of vector power amplifier 1200, process detectors are employed to compensate for any process variations in circuitry of the amplifier. In the exemplary embodiment of
Embodiment 1200A illustrates a multiple-input single-output (MISO) implementation of embodiment 1200. In embodiment 1200A, constant envelope signals 1261 and 1263, output from vector modulators 1260 and 1262, are input into MISO PA 1292. MISO PA 1292 is a two-input single-output power amplifier. In an embodiment, MISO PA 1292 includes elements 1270, 1272, 1282, 1284, and 1286 as shown in the embodiment of
Still referring to
In another implementation, shown in embodiment 1200B of
Other aspects of vector power amplifiers 1200A and 1200B substantially correspond to those described above with respect to vector power amplifier 1200.
In the exemplary embodiment of
DAC 1320 receives information signal 1310 from I and Q Data Transfer Function module 1216. Information signal 1310 includes identical information content to signals 1220, 1222, 1224 and 1226 in the embodiment of
DAC 1320 may output a single analog signal at a time. Accordingly, a sample-and-hold architecture may be used as shown in
DAC 1320 sequentially outputs analog signals 1332, 1334, 1336, 1336 to a first set of sample-and-hold circuits 1342, 1344, 1346, and 1348. In an embodiment, LAC 1230 is clocked at a sufficient rate to replace DACs 1230, 1232, 1234, and 1236 of the embodiment of
DAC 1320's DAC clock signal 1324, output selector signal 1322, and sample-and-hold clocks 1340A-D and 1350 are controlled by a control module that can be independent or integrated into transfer function module 1216.
In an embodiment, sample-and-hold circuits (S/H) 1342, 1344, 1346, and 1348 hold the received analog values and, according to a clock signal 1340A-D, release the values to a second set of sample-and-hold circuits 1352, 1354, 1356, and 1358. For example, S/H 1342 release its value to S/H 1352 according to a received clock signal 1340A. In turn, sample-and-hold circuits 1352, 1354, 1356, and 1358 hold the received analog values, and simultaneously release the values to interpolation filters 1231, 1233, 1235, and 1237 according to a common clock signal 1350. A common clock signal 1350 is used in order to ensure that the outputs of S/H 1352, 1354, 1356, and 1358 are time-aligned.
In another embodiment, a single layer of S/H circuitry that includes S/H 1342, 1344, 1346, and 1348 can be employed. Accordingly, S/H circuits 1342, 1344, 1346, and 1348 receive analog values from DAC 1320, and each releases its received value according to a clock independent of the others. For example, S/H 1342 is controlled by clock 1340A, which may not be synchronized with clock 1340B that controls S/H 1344. To ensure that outputs of S/H circuits 1342, 1344, 1346, and 1348 are time-aligned, delays between clocks 1340A-D are pre-compensated for in prior stages of the amplifier. For example, DAC 1320 outputs signal 1332, 1334, 1336, and 1338 with appropriately selected delays to S/H circuits 1342, 1344, 1346, and 1348 in order to compensate for the time differences between clocks 1340A-D.
Other aspects of vector power amplifier 1300 are substantially equivalent to those described above with respect to vector power amplifier 1200.
In the embodiment of
The embodiment of
Embodiment 1300A also illustrates optional bias control circuitry 1218 and associated bias control signal 1325, 1326, and 1327. Signals 1325, 1326, and 1327 may be used to bias different stages of MISO PA 1360 in certain embodiments.
Other aspects of vector power amplifier 1300A are equivalent to those described above with respect to vector power amplifiers 1200 and 1300.
3.3) Direct Cartesian 2-Branch Vector Power Amplifier
A Direct Cartesian 2-Branch VPA embodiment shall now be described. This name is used herein for reference purposes, and is not functionally or structurally limiting.
According to the Direct Cartesian 2-Branch VPA embodiment, a time-varying envelope signal is decomposed into two constant envelope constituent signals. The constituent signals are individually amplified equally or substantially equally, and then summed to construct an amplified version of the original time-varying envelope signal.
In one embodiment of the Direct Cartesian 2-Branch VPA embodiment, a magnitude and a phase angle of a time-varying envelope signal are calculated from in-phase and quadrature components of an input signal. Using the magnitude and phase information, in-phase and quadrature amplitude components are calculated for two constant envelope constituents of the time-varying envelope signal. The two constant envelope constituents are then generated, amplified equally or substantially equally, and summed to generate an amplified version of the original time-varying envelope signal Rin.
The concept of the Direct Cartesian 2-Branch VPA will now be described with reference to
As described and verified above with respect to
For the case illustrated in
in
where R represents the normalized magnitude of phasor {right arrow over (R′)}.
In the time domain, it was shown that a time-varying envelope signal, r′(t)=R(t) cos(ωt) for example, can be constructed by the sum of two constant envelope signals as follows:
r(t)=U′(t)+L′(t);
U′(t)=C×cos(ωt)+α×sin(ωt);
L′(t)=C×cos(ωt)−β×sin(ωt). (14)
where C denotes the in-phase amplitude component of phasors {right arrow over (U′)} and {right arrow over (L′)} and is equal or substantially equal to
(A being a constant). α and β denote the quadrature amplitude components of phasors {right arrow over (U′)} and {right arrow over (U′)}, respectively.
Note that equations (14) can be modified for non-sinusoidal signals by changing the basis function from sinusoidal to the desired function.
{right arrow over (R)}={right arrow over (R′)}×ejθ=({right arrow over (U′)}+{right arrow over (L′)})×ejθ={right arrow over (U)}+{right arrow over (L)};
{right arrow over (U)}={right arrow over (U′)}×ejθ;
{right arrow over (L)}={right arrow over (L′)}×ejθ, (15)
From equations (15), it can be further shown that:
{right arrow over (U)}={right arrow over (U′)}×ejθ=(C+jα)×ejθ;{right arrow over (U)}=(C+jα)(cos θ+j sin θ)=(C cos θ−α sin θ)+j(C sin θ+α cos θ). (16)
Similarly, it can be shown that:
{right arrow over (L)}={right arrow over (L′)}×ejθ=(C+jβ)×ejθ;{right arrow over (L)}=(C+jβ)(cos θ+j sin θ)=(C cos θ−β sin θ)+j(C sin θ+β cos θ). (17)
Equations (16) and (17) can be re-written as:
{right arrow over (U)}=(C cos θ−α sin θ)+j(C sin θ+α cos θ)=Ux+jUy;
{right arrow over (L)}=(C cos θ−β sin θ)+j(C sin θ+β cos θ)=Lx+jLy. (18)
Equivalently, in the time domain:
U(t)=Uxφ1(t)+Uyφ2(t);
L(t)=Lxφ1(t)+Lyφ2(t); (19)
where φ1(t) and φ2(t) represent an appropriately selected orthogonal basis functions.
From equations (18) and (19), it is noted that A is sufficient to calculate the values of α, β, C and sin(Θ) and cos(Θ) in order to determine the two constant envelope constituents of a time-varying envelope signal r(t). Further, α, β and C can be entirely determined from magnitude and phase information, equivalently I and Q components, of signal r(t).
In the example of
r Referring to
Still referring to
Accordingly, respective output signals 1540 and 1542 of vector modulators 1520 and 1530 correspond, respectively, to the U(t) and L(t) constant envelope constituents of signal r(t) as described above in equations (19). As described above, signals 1540 and 1542 are characterized by having equal and constant or substantially equal and constant magnitude envelopes.
Referring to
In an embodiment, power amplifiers 1550 and 1560 apply equal or substantially equal power amplification to signals 1540 and 1542, respectively. In an embodiment, the power amplification level of PAs 1550 and 1560 is set according to the desired power level of output signal r(t).
Amplified output signals 1562 and 1564 are substantially constant envelope signals. Accordingly, when summed together, as shown in
In embodiment 1500A, constant envelope signals 1540 and 1542, output from vector modulators 1520 and 1530, are input into MISO PA 1580. MISO PA 1580 is a two-input single-output power amplifier. In an embodiment, MISO PA 1580 may include various elements, such as pre-drivers, drivers, power amplifiers, and process detectors (not shown in
Operation of the Direct Cartesian 2-Branch VPA embodiment is depicted in the process flowchart 1600 of
Step 1620 includes receiving a clock signal set according to a desired output signal frequency of the desired output signal. In the example of
Step 1630 includes processing the I and Q components to generate in-phase and quadrature amplitude information of first and second constant envelope constituent signals of the desired output signal. In the example of HG. 15, the in-phase and quadrature amplitude information is illustrated by Ux, Uy, Lx, and Ly.
Step 1640 includes processing the amplitude information and the clock signal to generate the first and second constant envelope constituent signals of the desired output signal. In an embodiment, the first and second constant envelope constituent signals are modulated according to the desired output signal frequency. In the example of
Step 1650 includes amplifying the first and second constant envelope constituents, and summing the amplified signals to generate the desired output signal. In an embodiment, the amplification of the first and second constant envelope constituents is according to a desired power level of the desired output signal. In the example of
Referring to
In an embodiment, I and Q Data Transfer Function module 1716 processes information signal 1710 to generate information signals 1720, 1722, 1724, and 1726. The operation of I and Q Data Transfer Function module 1716 is further described below in section 3.4.
Referring to
Still referring to
In the exemplary embodiment of
In other embodiments, information signals 1720, 1722, 1724 and 1726 are generated in analog format and no DACs are required.
Referring to
Still referring to
Referring to
Output signals 1754 and 1756 represent constant envelope signals. A sum of output signals 1754 and 1756 results in a carrier signal having the I and Q characteristics of the original baseband signal. In embodiments, to generate a desired power level at the output of vector power amplifier 1700, signals 1754 and 1756 are amplified and then summed. In the embodiment of
In an embodiment, PAs 1760 and 1762 apply equal or substantially equal power amplification to respective constant envelope signals 1754 and 1756. In an embodiment, the power amplification is set according to the desired output power level. In other embodiments of vector power amplifier 1700, PA drivers are additionally employed to provide additional power amplification capability to the amplifier. In the embodiment of
Respective output signals 1764 and 1766 of PAs 1760 and 1762 are substantially constant envelope signals. In the embodiment of FIG. 17, output signals 1764 and 1766 are coupled together to generate output signal 1770 of vector power amplifier 1700. In embodiments, it is noted that the outputs of PAs 1760 and 1762 are directly coupled. Direct coupling in this manner means that there is minimal or no resistive, inductive, or capacitive isolation between the outputs of PAs 1760 and 1762. In other words, outputs of PAs 1760 and 1762 are coupled together without intervening components. Alternatively, in an embodiment, the outputs of PAs 1760 and 1762 are coupled together indirectly through inductances and/or capacitances that result in low or minimal impedance connections, and/or connections that result in minimal isolation and minimal power loss. Alternatively, outputs of PAs 1760 and 1762 are coupled using well known combining techniques, such as Wilkinson, hybrid couplers, transformers, or known active combiners. In an embodiment, the PAs 1760 and 1762 provide integrated amplification and power combining in a single operation. In an embodiment, one or more of the power amplifiers and/or drivers described herein are implemented using multiple input, single output (MISO) power amplification techniques, examples of which are shown in
Output signal 1770 represents a signal having the desired I and Q characteristics of the baseband signal and the desired output power level and frequency. In embodiments of vector power amplifier 1700, a pull-up impedance 1778 is coupled between the output of vector power amplifier 1700 and a power supply. In other embodiments, an impedance matching network 1780 is coupled at the output of vector power amplifier 1700. Output stage embodiments according to power amplification methods and systems of the present invention will be further described below in section 3.5.
In other embodiments of vector power amplifier 1700, process detectors are employed compensate for any process and/or temperature variations in circuitry of the amplifier. In the exemplary embodiment of
In another embodiment of embodiment 1700, shown as embodiment 1700B of
In the exemplary embodiment of
DAC 1820 receives information signal 1810 from I and Q Data Transfer Function module 1716. Information signal 1810 includes identical information content to signals 1720, 1722, 1724, and 1726 in the embodiment of
DAC 1820 may output a single analog signal at a time. Accordingly, a sample-and-hold architecture may be used as shown in
In the embodiment of
DAC 1820's DAC clock signal 1814, output selector signal 1812, and sample-and-hold clocks 1830A-D, and 1840 are controlled by a control module that can be independent or integrated into transfer function module 1716.
In an embodiment, sample-and-hold circuits 1832, 1834, 1836, and 1838 sample and hold their respective values and, according to a clock signal 1830A-D, release the values to a second set of sample-and-hold circuits 1842, 1844, 1846, and 1848. For example, S/H 1832 release's its value to S/H 1842 according to a received clock signal 1830A. In turn, sample-and-hold circuits 1842, 1844, 1846, and 1848 hold the received analog values, and simultaneously release the values to interpolation filters 1852, 1854, 1856, and 1858 according to a common clock signal 1840.
In another embodiment, a single set of S/H circuitry that includes S/H 1832, 1834, 1836, and 1838 can be employed. Accordingly, S/H circuits 1832, 1834, 1836, and 1838 receive analog values from DAC 1820, and each samples and holds its received value according to independent clocks 1830A-D. For example, S/H 1832 is controlled by clock 1830A, which may not be synchronized with clock 1830B that controls S/H 1834. For example, DAC 1820 outputs signals 1822, 1824, 1826, and 1828 with appropriately selected analog values calculated by transfer function module 1716 to S/H circuits 1832, 1834, 1836, and 1838 in order to compensate for the time differences between clocks 830A-D.
Other aspects of vector power amplifier 1800 correspond substantially to those described above with respect to vector power amplifier 1700.
In the embodiment of
The embodiment of
Other aspects of vector power amplifier 1800A are substantially equivalent to those described above with respect to vector power amplifiers 1700 and 1800.
3.4) I and Q Data to Vector Modulator Transfer Functions
In some of the above described embodiments, I and Q data transfer functions are provided to transform received I and Q data into amplitude information inputs for subsequent stages of vector modulation and amplification. For example, in the embodiment of
According to the present invention, I and Q Data Transfer Function modules may be implemented using digital circuitry, analog circuitry, software, firmware or any combination thereof.
Several factors affect the actual implementation of a transfer function according to the present invention, and vary from embodiment to embodiment. In one aspect, the selected VPA embodiment governs the amplitude information output of the transfer function and associated module. It is apparent, for example, that I and Q Data Transfer Function module 1216 of the CPCP 2-Branch VPA embodiment 1200 differs in output than I and Q Data Transfer Function module 1716 of the Direct Cartesian 2-Branch VPA embodiment 1700.
In another aspect, the complexity of the transfer function varies according to the desired modulation scheme(s) that need to be supported by the VPA implementation. For example, the sample clock, the DAC sample rate, and the DAC resolution are selected in accordance with the appropriate transfer function to construct the desired output waveform(s).
According to the present invention, transfer function embodiments may be designed to support one or more VPA embodiments with the ability to switch between the supported embodiments as desired. Further, transfer function embodiments and associated modules can be designed to accommodate a plurality of modulation schemes. A person skilled in the art will appreciate, for example, that embodiments of the present invention may be designed to support a plurality of modulation schemes (individually or in combination) including, but not limited to, BPSK, QPSK, OQPSK, DPSK, CDMA, WCDMA, W-CDMA, GSM, EDGE, MPSK, MQAM, MSK, CPSK, PM, FM, OFDM, and multi-tone signals. In an embodiment, the modulation scheme(s) may be configurable and/or programmable via the transfer function module.
3.4.1) Cartesian 4-Branch VPA Transfer Function
Step 1920 includes calculating a phase shift angle between first and second substantially equal and constant envelope constituents of the I component. In parallel, step 1920 also includes calculating a phase shift angle between first and second substantially equal and constant envelope constituents of the Q component. As described above, the first and second constant envelope constituents of the I components are appropriately phased relative to the I component. Similarly, the first and second constant envelope constituents of the Q components are appropriately phased relative to the Q component. In the embodiment of
Step 1930 includes calculating in-phase and quadrature amplitude information associated with the first and second constant envelope constituents of the I component. In parallel, step 1930 includes calculating in-phase and quadrature amplitude information associated with the first and second constant envelope constituents of the Q component. In the embodiment of
Step 1940 includes outputting the calculated amplitude information to a subsequent vector modulation stage. In the embodiment of
Referring to
Components 2020 and 2022 output the measured I and Q magnitude information to components 2030 and 2032, respectively, of transfer function module 2000. In an embodiment, the measured I and Q magnitude information is in the form of digital signals. Based on the I magnitude information, component 2030 calculates a phase shift angle φ1 between first and second equal and constant or substantially equal and constant envelope constituents of the sampled I signal. Similarly, based on the Q magnitude information, component 2032 calculates phase shift angle φQ between a first and second equal and constant or substantially equal and constant envelope constituents of the sampled Q signal. This operation shall now be further described.
In the embodiment of
Referring to
Components 2040 and 2042 output the calculated amplitude information to subsequent stages of the vector power amplifier. In embodiments, each of the four calculated values is output separately to a digital-to-analog converter. As shown in the embodiment of
3.4.2) CPCP 2-Branch VPA Transfer Function
Step 2120 includes determining the magnitudes |I| and |Q| of the received I and Q data components.
Step 2130 includes calculating a magnitude |R| of the baseband signal based on the measured |I| and |Q| magnitudes. In an embodiment, |R| is such that |R|2=|I|2+|Q|2. In the embodiment of
Step 2140 includes normalizing the measured III and IQI magnitudes. In an embodiment, |I| and |Q| are normalized to generate an Iclk_phase and Qclk_phase signals (as shown in
Step 2150 includes calculating in-phase and quadrature amplitude information associated with first and second constant envelope constituents. In the embodiment of
Step 2160 includes outputting the generated Iclk_phase and Qclk_phase (from step 2140) and the calculated amplitude information (from step 2150) to appropriate vector modulators. In the embodiment of
In an embodiment, transfer function module 2200 samples I and Q data signal 2210 according to a sampling clock 2212. Sampled I and Q data signals are received by component 2220 of transfer function module 2200. Component 2220 measures the magnitudes |{right arrow over (I)}| and |{right arrow over (Q)}| of the sampled I and Q data signals.
Based on the measured |{right arrow over (I)}| and |{right arrow over (Q)}| magnitudes, component 2230 calculates the magnitude |R| of the baseband signal. In an embodiment, |{right arrow over (R)}| is such that |{right arrow over (R)}|2=|{right arrow over (I)}|2+|{right arrow over (Q)}|2.
In parallel, component 2240 normalizes the measured |{right arrow over (I)}| and |{right arrow over (Q)}|magnitudes. In an embodiment, |{right arrow over (I)}| and |{right arrow over (Q)}| are normalized to generate Iclk_phase and Qclk_phase signals such that |Iclk_phase|2+|Qclk_phase|2=constant, where |Iclk_phase| and |Qclk_phase| represent normalized magnitudes of |{right arrow over (I)}| and |{right arrow over (Q)}|. Typically, given that the constant has a value A, the measured |{right arrow over (I)}| and |{right arrow over (I)}| magnitudes are both divided by the quantity
Component 2250 receives the calculated |{right arrow over (R)}| magnitude from component 2230, and based on it calculates a phase shift angle φ between first and second constant envelope constituents. Using the calculated phase shift angle φ, component 2050 then calculates in-phase and quadrature amplitude information associated with the first and second constant envelope constituents.
In the embodiment of
Referring to
3.4.3) Direct Cartesian 2-Branch Transfer Function
Step 2320 includes determining the magnitudes |I| and |Q| of the received I and Q data components.
Step 2330 includes calculating a magnitude |R| of the baseband signal based on the measured |I| and |Q| magnitudes. In an embodiment, |R| is such that |R|2=|I|2+|Q|2. In the embodiment of
Step 2340 includes calculating a phase shift angle θ of the baseband signal based on the measured |I| and |Q| magnitudes. In an embodiment, θ is such that
and wherein the sign of I and Q determine the quadrant of θ. In the embodiment of
Step 2350 includes calculating in-phase and quadrature amplitude information associated with a first and second constant envelope constituents of the baseband signal. In the embodiment of
Step 2360 includes outputting the calculated amplitude information to DAC's for input into the appropriate vector modulators. In the embodiment of
In an embodiment, transfer function module 2400 samples I and Q data signal 2410 according to a sampling clock 2412. Sampled I and Q data signals are received by component 2420 of transfer function module 2200. Component 2420 measures the magnitudes |{right arrow over (I)}| and |{right arrow over (Q)}| of the sampled I and Q data signals.
Based on the measured |{right arrow over (I)}| and |{right arrow over (Q)}| magnitudes, component 2430 calculates the magnitude |{right arrow over (R)}|. In an embodiment, |{right arrow over (R)}| is such that |{right arrow over (R)}|2=|{right arrow over (I)}|2+|{right arrow over (Q)}|2.
In parallel, component 2240 calculates the phase shift angle θ of the baseband signal. In an embodiment, θ is such that
where the sign of I and Q determine the quadrant of θ.
Component 2450 receives the calculated |{right arrow over (R)}| magnitude from component 2430, and based on it calculates a phase shift angle φ between first and second constant envelope constituent signals. In the embodiment of
In parallel, component 2450 receives the calculated phase shift angle θ from component 2440. As functions of φ and θ, component 2450 then calculates in-phase and quadrature amplitude information for the vector modulator inputs that generate the first and second constant envelope constituents. In an embodiment, the in-phase and quadrature amplitude information supplied to the vector modulators are according to the equations, provided in (18).
Component 2450 outputs the calculated amplitude information to subsequent stages of the vector power amplifier. In embodiments, the output values are separately output to digital-to-analog converters. As shown in the embodiment of
3.4.4) Magnitude to Phase Shift Transform
Embodiments of f(|I|), f(|Q|) of
According to the present invention, any periodic waveform that can be represented by a Fourier series and a Fourier transform can be decomposed into two or more constant envelope signals.
Below are provided two examples for sinusoidal and square waveforms.
3.4.4.1) Magnitude to Phase Shift Transform for Sinusoidal Signals:
Consider a time-varying complex envelope sinusoidal signal r(t). In the time domain, it can be represented as:
r(t)=R(t)sin(ωt+δ(t)) (20)
where R(t) represents the signal's envelope magnitude at time t, δ(t) represents the signal's phase shift angle at time t, and ω represents the signal's frequency in radians per second.
It can be verified that, at any time instant t, signal r(t) can be obtained by the sum of two appropriately phased equal and constant or substantially equal and constant envelope signals. In other words, it can be shown that
R(t)sin(ωt+δ(t))=A sin(ωt)+A sin(ωt+φ(t)) (21)
for an appropriately chosen phase shift angle φ(t) between the two constant envelope signals. The phase shift angle φ(t) will be derived as a function of R(t) in the description below. This is equivalent to the magnitude to phase shift transform for sinusoidal signals.
Using a sine trigonometric identity, equation (21) can be re-written as:
R(t)sin(ωt+δ(t))=A sin(ωt)+A sin(ωt)cos φ(t)+A sin(φ(t))cos ωt; R(t)sin(ωt+δ(t))=A sin(φ(t))cos ωt+A(1+cos φ(t))sin ωt. (22)
Note, from equation (22), that signal r(t) is written as a sum of an in-phase component and a quadrature component. Accordingly, the envelope magnitude R(t) can be written as:
R(t)=√{square root over ((A sin(φ(t)))2+(A(1+cos(φ(t))))2)}{square root over ((A sin(φ(t)))2+(A(1+cos(φ(t))))2)};R(t)=√{square root over (2A(A+cos(φ(t))))} (23)
Equation (23) relates the envelope magnitude R(t) of signal r(t) to the phase shift angle φ(t) between two constant envelope constituents of signal r(t). The constant envelope constituents have equal or substantially equal envelope magnitude A, which is typically normalized to 1.
Inversely, from equation (23), the phase shift angle φ(t) can be written as a function of R(t) as follows:
Equation (24) represents the magnitude to phase shift transform for the case of sinusoidal signals, and is illustrated in
3.4.4.2) Magnitude to Phase Shift Transform for Square Wave Signals:
Signal 2830 results from combining signals 2810 and 2820. According to embodiments of the present invention, signal 2830 will have a magnitude equal or substantially equal to a product of signals 2810 and 2820. In other words, signal 2830 will have a magnitude of zero whenever either of signals 2810 or 2820 has a magnitude of zero, and a non-zero magnitude when both signals 2810 and 2820 have non-zero magnitudes.
Further, signal 2830 represents a pulse-width-modulated signal. In other words, the envelope magnitude of signal 2830 is determined according to the pulse width of signal 2830 over one period of the signal. More specifically, the envelope magnitude of signal 2830 is equal or substantially to the area under the curve of signal 2830.
Referring to
radians.
Still referring to
R=A1×A2×(γT−t′) (25)
Accordingly, it can be deduced that φ is related to R according to:
Note, from equation (26), that R is at a maximum of γA1A2 when φ=0. In other words, the envelope magnitude is at a maximum when the two constant envelope signals are in-phase with each other.
In typical implementations, signals 2810 and 2820 are normalized and have equal or substantially equal envelope magnitude of 1. Further, signals 2810 and 2820 typically have a duty cycle of 0.5. Accordingly, equation (28) reduces to:
Equation (27) illustrates the magnitude to phase shift transform for the case of normalized and equal or substantially equal envelope magnitude square wave signals. Equation (27) is illustrated in
3.4.5) Waveform Distortion Compensation
In certain embodiments, magnitude to phase shift transforms may not be implemented exactly as theoretically or practically derived. In fact, several factors may exist that require adjustment or tuning of the derived transform for optimal operation. Waveform distortion compensation is one facto' that may require adjustment or tuning of the magnitude to phase shift transform. Waveform distortion compensation is now described below.
In practice, several factors may cause waveform distortion of constant envelope constituents of a desired output signal r(t). Furthermore, it is expected that waveform distortion in the constituents translates into waveform distortion in the desired output signal when the constituents are combined.
According to embodiments of the present invention, expected waveform distortion to a desired output signal can be estimated. In some embodiments, the expected waveform distortion is compensated for at the vector modulation stage of the power amplifier. In other embodiments, the expected waveform distortion is compensated for at the transfer function stage of the power amplifier.
In the former approach, compensation is achieved by applying appropriate magnitude and phase shift tuning at the vector modulation stage outputs. Accordingly, waveform distortion is removed by waveshaping of the constituent signals of the desired output signal.
In the latter approach, the transfer function is designed to factor in and cancel or at least reduce the effects of the expected waveform distortion. As can be understood from the above description of transfer functions, waveform distortion compensation can be introduced at different positions within a transfer function stage. In some embodiments, compensation is applied at the output stage, of the transfer function. For example, referring to
3.5) Output Stage
An aspect of embodiments of the present invention lies in summing constituent signals at the output stage of a vector power amplifier (VPA). This is shown, for example, in
In the example of
In another aspect, output stage embodiments of the present invention can be implemented using multiple-input single-output (MISO) power amplifiers.
In another aspect, output stage embodiments of the present invention can be controlled to increase the powe efficiency of the amplifier by controlling the output stage current according to the desired output power level.
In what follows, various output stage embodiments according to VPA embodiments of the present invention are provided in section 3.5.1. In section 3.5.2, embodiments of output stage current shaping functions, for increasing the power efficiency of certain VPA embodiments of the present invention, are presented. Section 3.5.3 describes embodiments of output stage protection techniques that may be utilized for certain output stage embodiments of the present invention.
3.5.1) Output Stage Embodiments
PA branches 3005-{1, . . . , n} apply equal or substantially equal power amplification to respective signals 3010-{1, . . . , n}. In an embodiment, the power amplification level through PA branches 3005-{1, . . . , n} is set according to a power level requirement of the desired output signal.
In the embodiment of
To generate the desired output signal, outputs of PA branches 3005-{1, . . . , n} are coupled directly at summing node 3050. Summing node 3050 provides little or no isolation between the coupled outputs. Further, summing node 3050 represents a relatively lossless summing node. Accordingly, minimal power loss is incurred in summing the outputs of PAs 3040-{1, . . . , n}.
Output signal 3060 represents the desired output signal of output stage 3000. In the embodiment of
In the above-described embodiments of
According to embodiment 3600, PA's 3620-{1, . . . , n} include switching power amplifiers. In the example of
Embodiments are not limited to npn BJT implementations as described herein. A person skilled in the art will appreciate, for example, that embodiments of the present invention may be implemented using pnp BJTs, CMOS, NMOS, PMOS, or other type of transistors. Further, embodiments can be implemented using GaAs and/or SiGe transistors with the desired transistor switching speed being a factor to, consider.
Referring back to
Referring to
Still referring to
It is noted that, in the embodiment of
In another aspect, output stage embodiments can be implemented using multiple-input single-output (MISO) power amplifiers.
It is noted that MISO implementations, similar to the one shown in
Referring to
Still referring to
Embodiment 5110D represents an npn implementation of the two-input single output PA building block. Embodiment 5110D includes two npn transistors coupled together using a common collector node, which provides the output of the PA. A pull-up impedance (not shown) can be coupled between the common collector node and a supply node (not shown).
Embodiment 5130D represents a pnp equivalent of embodiment 5110D. Embodiment 5130D includes two pnp transistors coupled at a common collector node, which provides the output of the PA. A pull-down impedance (not shown) can be coupled between the common collector node and a ground node (not shown).
Embodiment 5140D represents a complementary npn/pnp implementation of the two-input single output PA building block. Embodiment 5140D includes an npn transistor and a pnp transistor coupled at a common collector node, which provides the output of the PA.
Still referring to
Embodiment 5160D represents an PMOS equivalent of embodiment 5120D. Embodiment 5120D includes two PMOS transistors coupled at a common drain node, which provides the output of the PA.
Embodiment 5150D represents a complementary MOS implementation of the two-input single-output PA building block. Embodiment 5150D includes a PMOS transistor and an NMOS transistor coupled at common drain node, which provides the output of the PA.
Two-input single-output embodiments of
Embodiment 5150E represents an npn implementation of a multiple-input single-output A. Embodiment 5150E includes a plurality of npn transistors coupled together using a common collector node, which provides the output of the PA. A pull-up impedance (not shown) can be coupled between the common collector node and a supply voltage (not shown). Note that an n-input single-output PA according to embodiment 5150E can be obtained by coupling additional npn transistors to the two-input single-output PA building block embodiment 5110D.
Embodiment 5170E represents a pnp equivalent of embodiment 5150E. Embodiment 5170E includes a plurality of pnp transistors coupled together using a common collector node, which provides the output of the PA. A pull-down impedance (not shown) may be coupled between the common collector node and a ground node (not shown). Note than an n-input single-output PA according to embodiment 5170E can be obtained by coupling additional pnp transistors to the two-input single-output PA building block embodiment 5130D.
Embodiments 5110E and 5130E represent complementary npn/pnp implementations of a multiple-input single-output PA. Embodiments 5110E and 5130E may include a plurality of npn and/or pnp transistors coupled together using a common collector node, which provides the output of the PA. Note that an n-input single-output PA according to embodiment 5110E can be obtained by coupling additional npn and/or pnp transistors to the two-input single-output PA building block embodiment 5140D. Similarly, an n-input single-output PA according to embodiment 5130E can be obtained by coupling additional npn and/or pnp transistors to the two-input single-output PA building block embodiment 5130D.
Embodiment 5180E represents an PMOS implementation of a multiple-input single-output PA. Embodiment 5180E includes a plurality of PMOS transistors coupled together using a common drain node, which provides the output of the PA. Note that an n-input single-output PA according to embodiment 5180E can be obtained by coupling additional NMOS transistors to the two-input single-output PA building block embodiment 5160D.
Embodiment 5100E represents a NMOS implementation of multiple-input single-output PA. Embodiment 5160E includes a plurality of N: MOS transistors coupled together using a common drain node, which provides the output of the PA. Note that an n-input single-output PA according to embodiment 5160E can be obtained by coupling additional PMOS transistors to the two-input single-output PA building block embodiment 5120D.
Embodiments 5120E and 5140E complementary MOS implementations of a multiple-input single-output PA. Embodiments 5120E and 5140E include a plurality of npn and pnp transistors coupled together using a common drain node, which provides the output of the PA. Note that a n-input single-output PA according to embodiment 5120E can be obtained by coupling additional NMOS and/or PMOS transistors to the two-input single-output PA building block 5150D. Similarly, an n-input single-output PA according to embodiment 5140E can be obtained by coupling additional NMOS and/or PMOS transistors to the two-input single-output PA building block 5160D.
It must be noted that the multiple-input single-output embodiments described above may each correspond to a single or multiple branches of a PA. For example, referring to
It is further noted that the transistors shown in the embodiments of
Still referring to
3.5.2) Output Stage Current Control—Autobias Module
Embodiments of the output stage and optional pre-driver and driver stage bias and current control techniques according to embodiments of the present invention are described below. In certain embodiments, output stage current control functions are employed to increase the output stage efficiency of a vector power amplifier (VPA) embodiment In other embodiments, output stage current control is used to provide output stage protection from excessive voltages and currents which is further describe in section 3.5.3. In embodiments, output stage current control functions are performed using the Autobias module described above with reference to
According to embodiments of the present invention, power efficiency of the output stage of a VPA can be increased by controlling the output stage current of the VPA as a function of the output power and the envelope of the output waveform.
Note that Iout signal 4020 varies as a function of envelope signal 3920. Accordingly, Iout signal 4020 is at the maximum when a maximum output power is required, but decreases as the required output power goes down. Particularly, Iout signal 4020 approaches zero as the associated output power goes to zero. Accordingly, a person skilled in the art will appreciate that output stage current control, according to embodiments of the present invention, results in significant power savings and increases the power efficiency of the power amplifier.
According to embodiments of the present invention, output stage current control may be implemented according to a variety of functions. In an embodiment, the output stage current can be shaped to correspond to the desired output power of the amplifier. In, such an embodiment, the output stage current is a function that is derived from the envelope of the desired output signal, and the power efficiency will increase.
In implementation, several approaches exist for performing output stage current control. In some embodiments, output stage current shaping is performed using the Autobias module. The Autobias module is illustrated as autobias circuitry 714 and 716 in the embodiments of
Output stage current control using Autobias is depicted in process flowchart 4800 of the embodiment of
Step 4820 includes calculating a signal according to the output power and output envelope signal information. In embodiments, an Autobias signal is calculated as a function of some measure of the desired output power. For example, the Autobias signal may be calculated as a function of the envelope magnitude of the desired output signal. Referring to the embodiments of
Step 4830 includes applying the calculated signal at an output stage of the VPA, thereby controlling a current of the output stage according to the output power of the desired output signal. In embodiments, step 4830 includes coupling the Autobias signal at the PA stage input of the VPA. This is illustrated, for example, in the embodiments of
In other embodiments, step 4830 includes coupling the Autobias signal using pull-up impedances at the PA stage input and optionally the inputs of the driver and pre-driver stages of the VPA.
Embodiments for implementing the Autobias circuitry described above will now be provided.
In embodiment 2700A, Autobias circuitry 2700A includes an Autobias Transfer Function module 2712, a DAC 2714, and an optional interpolation filter 2718. Autobias circuitry 2700A receives an I and Q Data signal 2710. Autobias Transfer Function module 2712 processes the received I and Q Data signal 2710 to generate an appropriate bias signal 2713. Autobias Transfer Function module 2712 outputs bias signal 2713 to DAC 2714. DAC 2714 is controlled by a DAC clock 2716 which may be generated in Autobias transfer module 2712. DAC 2714 converts bias signal 2713 into an analog signal, and outputs the analog signal to interpolation filter 2718. Interpolation filter 2718, which also serves as an anti-aliasing filter, shapes the DAC's output to generate Autobias signal 2720, illustrated as Bias A in embodiment 5112G. Autobias signal 2720 may be used to bias the PA stage and/or the driver stage, and/or the pre-driver stage of the amplifier. In an embodiment, Autobias signal 2720 may have several other Autobias signals derived therefrom to bias different stages within the PA stage. This can be done using additional circuitry not included in embodiment 2700A.
In contrast, embodiment 2700B illustrates an Autobias circuitry embodiment in which multiple Autobias signals are derived within the Autobias circuitry. As shown in embodiment 2700B, circuit networks 2722, 2726, and 2730, illustrated as circuit networks A, B, and C in embodiment 2700B, are used to derive Autobias signals 2724 and 2728 from Autobias signal 2720. Autobias signals 2720, 2724, and 2728 are used to bias different amplification stages.
Embodiment 2700C illustrates another Autobias circuitry embodiment in which multiple Autobias signals are generated independently within the Autobias Transfer Function module 2712. In embodiment 2700C, Autobias Transfer Function module 2712 generates multiple bias signals according to the received I and Q Data signal 2710. The bias signals may or may not be related. Autobias Transfer Function module 2712 outputs the generated bias signals to subsequent DACs 2732, 2734, and 2736. DACs 2732, 2734, and 2736 are controlled by DAC clock signals 2733, 2735, and 2737, respectively. DACs 2732, 2734, and 2736 convert the received bias signals into analog signals, and output the analog signals to optional interpolation filters 2742, 2744, and 2746. Interpolation filters 2742, 2744, and 2746, which also serve as anti-aliasing filters, shape the DACs outputs to generate Autobias signals 2720, 2724, and 2728. Similar to embodiment 2700B, Autobias signals 2720, 2724, and 2728 are used to bias different amplification stages such as the pre-driver, driver, and PA.
As noted above, Autobias circuitry embodiments according to the present invention are not limited to the ones described in embodiments 2700A, 2700B, and 2700C. A person skilled in the art will appreciate, for example, that Autobias circuitry can be extended to generate any number of bias control signals as required to control the bias of various stages of amplification, and not just three as shown in embodiments 5200B and 5200C, for example.
3.5.3) Output Stage Protection
As described above, output stage embodiments according to embodiments of the present invention are highly power efficient as a result of being able, to directly couple, outputs at the PA stage using no combining or isolating elements. Certain output stage embodiments in certain circumstances and/or applications, however, may require additional special output stage protection measures in order to withstand such direct coupling approach. This may be the case for example for output stage embodiments such as 5110D, 5120D, 5130D, 5160D, 5150E, 5160E, 5170E, and 5180E illustrated in
In one aspect, transistors of distinct branches of a PA stage should generally not simultaneously be in opposite states of operation for extended periods of time. Following a restart or power on with no inputs being supplied to the final PA stages, transients within the PA branches may cause this mode to occur resulting in the PA stage transistors potentially damaging one another or circuit elements connected to the output. Accordingly, embodiments of the present invention further constrain the Autobias module to limit the output current in the PA stage.
In another aspect, it may be desired to ensure that the Autobias module limits the output voltages below the breakdown voltage specification of the PA stage transistors. Accordingly, in embodiments of the present invention, such as the one illustrated in
A person skilled in the art will appreciate that other output stage protection techniques may also be implemented. Furthermore, output stage protection techniques may be implementation specific. For example, depending on the type of PA stage transistors (npn, pnp, NMOS, PMOS, npn/pnp, NMOS/PMOS), different protection functions may be required.
3.6) Harmonic Control
According to embodiments of the present invention, an underlying principle for each branch PA is to maximize the transfer of power to a fundamental harmonic of the output spectrum. Typically, each branch PA may be multi-stage giving rise to a harmonically rich output spectrum. In one aspect, transfer of real power is maximized for the fundamental harmonic. In another aspect, for non-fundamental harmonics, real power transfer is minimized while imaginary power transfer may be tolerated. Harmonic control, according to embodiments of the present invention, may be performed in a variety of ways.
In one embodiment, real power transfer onto the fundamental harmonic is maximized by means of wave-shaping of the PA stage input signals. In practice, several factors play a role in determining the optimal wave shape that results in a maximum real power transfer onto the fundamental harmonic. Embodiment 3400 of the present invention, described above, represents one embodiment that employs waveshaping of PA stage input signals. In embodiment 3400, a plurality of harmonic control circuitry (HCC) networks 3410-{1, . . . , n} are coupled at the PA stage input of each PA branch {1, . . . , n}. HCC networks 3410-{1, . . . , n} have the effect of waveshaping the PA stage inputs, and are typically selected so as to maximize real power transfer to the fundamental harmonic of the summed output spectrum. According to embodiments of the present invention, waveshaping can be used to generate variations of harmonically diverse waveforms. In other embodiments, as can be apparent to a person skilled in the art, waveshaping can be performed at the pre-driver and/or the driver stage.
In another embodiment, harmonic control is achieved by means of waveshaping of the PA stage output.
Harmonic control using staggered bias is depicted in process flowchart 4900 of the embodiment of
Step 4920 includes coupling a plurality of impedances between the first ports of the plurality of transistors and a bias signal. In the example embodiment of
According to embodiments of the present invention, a variety of multi-stage staggered bias approaches may be designed. Bias impedance values may be fixed or variable. Furthermore, bias impedance values may be equal or substantially equal, distinct, or set according to a variety of permutations. For example, referring to the example of
3.7) Power Control
Vector power amplification embodiments of the present invention intrinsically provide a mechanism for performing output power control.
Phasors {right arrow over (U2)} and {right arrow over (L2)} represent upper and lower constituents of a second phasor {right arrow over (R2)}. {right arrow over (U2)} and {right arrow over (L2)} are constant magnitude and are symmetrically shifted in phase relative to {right arrow over (R2)} by a phase shift angle
It is noted, from
According to the above observation, output power control can be performed by imposing constraints on the phase shift angle of the constituent signals of a desired output signal. Referring to
can take, magnitude constraints can be imposed on phasor {right arrow over (R1)}.
According to embodiments of the present invention, a maximum output power level can be achieved by imposing a minimum phase shift angle condition. For example, referring to
the magnitude of phasor {right arrow over (R1)} is constrained not to exceed a certain maximum level. Similarly, a maximum phase shift angle condition imposes a minimum magnitude level requirement.
In another aspect of power control, output power resolution is defined in terms at a minimum power increment or decrement step size. According to an embodiment of the present invention, output power resolution may be implemented by defining a minimum phase shift angle step size. Accordingly, phase shift angle values are set according to a discrete value range having a pre-determined step size.
is set according to a pre-determined value range having a minimum step φstep.
A person skilled in the art will appreciate that a variety of power control schemes may be implemented in a fashion similar to the techniques described above. In other words, various power control algorithms can be designed, according to the present invention, by setting corresponding constraints on phase shift angle values. It is also apparent, based on the description above of data transfer functions, that power control schemes can be naturally incorporated into a transfer function implementation.
3.8) Exemplary Vector Power Amplifier Embodiment
Referring to
Further illustrated in
4. Summary
Mathematical basis for a new concept related to processing signals to provide power amplification and up-conversion is provided herein. These new concepts permit arbitrary waveforms to be constructed from sums of waveforms which are substantially constant envelope in nature. Desired output signals and waveforms may be constructed from substantially constant envelope constituent signals which can be created from the knowledge of the complex envelope of the desired output signal. Constituent signals are summed using new, unique, and novel techniques not available commercially, not taught or found in literature or related art. Furthermore, the blend of various techniques and circuits provided in the disclosure provide unique aspects of the invention which permits superior linearity, power added efficiency, monolithic implementation and low cost when compared to current offerings. In addition, embodiments of the invention are inherently less sensitive to process and temperature variations. Certain embodiments include the use of multiple input single output amplifiers described herein.
Embodiments of the invention can be implemented by a blend of hardware, software and firmware. Both digital and analog techniques can be used with or without microprocessors and DSP's.
Embodiments of the invention can be implemented for communications systems and electronics in general. In addition, and without limitation, mechanics, electro mechanics, electro optics, and fluid mechanics can make use of the same principles for efficiently amplifying and transducing signals.
5. Conclusion
The present invention has been described above with the aid of functional building blocks illustrating the performance of specified functions and relationships thereof. The boundaries of these functional building blocks have been arbitrarily defined herein for the convenience of the description. Alternate boundaries can be defined so long as the specified functions and relationships thereof are appropriately performed. Any such alternate boundaries are thus within the scope and spirit of the claimed invention. One skilled in the art will recognize that these functional building blocks can be implemented by discrete components, application specific integrated circuits, processors executing appropriate software and the like and combinations thereof.
While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.
The present application is a continuation of U.S. patent application Ser. No. 11/512,146, which was filed on Aug. 30, 2006, as a continuation of U.S. patent application Ser. No. 11/256,172, which was filed on Oct. 24, 2005 and claims the benefit of U.S. Provisional Patent Application No. 60/620,972 filed on Oct. 22, 2004, U.S. Provisional Patent Application No. 60/671,542 filed on Apr. 15, 2005, U.S. Provisional Patent Application No. 60/671,536 filed on Apr. 15, 2005, U.S. Provisional Patent Application No. 60/673,397 filed on Apr. 21, 2005, U.S. Provisional Patent Application No. 60/706,003 filed on Aug. 8, 2005, U.S. Provisional Patent Application No. 60/709,092 filed on Aug. 18, 2005, U.S. Provisional Patent Application No. 60/717,244 filed on Sep. 16, 2005, and U.S. Provisional Patent Application No. 60/721,114 filed on Sep. 28, 2005, all of which are incorporated herein by reference in their entirety.
| Number | Name | Date | Kind |
|---|---|---|---|
| 1882119 | Chireix | Oct 1932 | A |
| 1946308 | Chireix | Feb 1934 | A |
| 2116667 | Chireix | May 1938 | A |
| 2210028 | Doherty | Aug 1940 | A |
| 2220201 | Bliss | Nov 1940 | A |
| 2269518 | Chireix et al. | Jan 1942 | A |
| 2282706 | Chireix et al. | May 1942 | A |
| 2282714 | Fagot | May 1942 | A |
| 2294800 | Price | Sep 1942 | A |
| 2508524 | Lang | May 1950 | A |
| 2529073 | Chireix | Nov 1950 | A |
| 2555039 | Bissonette | May 1951 | A |
| 2591749 | Villemagne | Apr 1952 | A |
| 2670404 | Chireix | Feb 1954 | A |
| 2677806 | Chireix | May 1954 | A |
| 2714634 | Hall | Aug 1955 | A |
| 2734100 | Kendall | Feb 1956 | A |
| 2857591 | Nagel | Oct 1958 | A |
| 2890280 | Feyzeau | Jun 1959 | A |
| 2908753 | Ernyei et al. | Oct 1959 | A |
| 2938945 | France | May 1960 | A |
| 2963933 | Bereskin | Dec 1960 | A |
| 2964622 | Fire | Dec 1960 | A |
| 2968697 | Rager, Jr. | Jan 1961 | A |
| 3056017 | Peras | Sep 1962 | A |
| 3078456 | Alpers | Feb 1963 | A |
| 3121198 | Potter | Feb 1964 | A |
| 3154782 | Kagawa et al. | Oct 1964 | A |
| 3170127 | Cramer | Feb 1965 | A |
| 3176060 | Bissonette et al. | Mar 1965 | A |
| 3212008 | Kahn | Oct 1965 | A |
| 3219862 | Kieffert | Nov 1965 | A |
| 3263019 | Hurvitz | Jul 1966 | A |
| 3341697 | Kaufman et al. | Sep 1967 | A |
| 3413570 | Bruene et al. | Nov 1968 | A |
| 3418595 | Loewenstern, Jr. | Dec 1968 | A |
| 3436686 | Vackar | Apr 1969 | A |
| 3437945 | Duncan | Apr 1969 | A |
| 3458816 | O'Brien | Jul 1969 | A |
| 3493718 | Kestner et al. | Feb 1970 | A |
| 3513352 | Souillard | May 1970 | A |
| 3525941 | Smith | Aug 1970 | A |
| 3544697 | Munch, Jr. | Dec 1970 | A |
| 3651429 | Ruthroff | Mar 1972 | A |
| 3697692 | Hafler | Oct 1972 | A |
| 3716730 | Cerny, Jr. | Feb 1973 | A |
| 3777275 | Cox | Dec 1973 | A |
| 3789314 | Beurrier | Jan 1974 | A |
| 3815040 | Seidel | Jun 1974 | A |
| 3852530 | Shen | Dec 1974 | A |
| 3852669 | Bowman et al. | Dec 1974 | A |
| 3895304 | Klein | Jul 1975 | A |
| 3896395 | Cox | Jul 1975 | A |
| 3906390 | Rollett | Sep 1975 | A |
| 3909742 | Cox et al. | Sep 1975 | A |
| 3927379 | Cox et al. | Dec 1975 | A |
| 3936819 | Angelle et al. | Feb 1976 | A |
| 3991343 | Delpy | Nov 1976 | A |
| 4090147 | Seidel | May 1978 | A |
| 4095196 | Seidel | Jun 1978 | A |
| 4104946 | Peterson | Aug 1978 | A |
| 4151517 | Kelley | Apr 1979 | A |
| 4178557 | Henry | Dec 1979 | A |
| 4229715 | Henry | Oct 1980 | A |
| 4301490 | Nagel et al. | Nov 1981 | A |
| 4346354 | Hanna | Aug 1982 | A |
| 4378530 | Garde | Mar 1983 | A |
| 4433312 | Kahn | Feb 1984 | A |
| 4439744 | Kumar et al. | Mar 1984 | A |
| 4441080 | Saari | Apr 1984 | A |
| 4446440 | Bell | May 1984 | A |
| 4485357 | Voorman | Nov 1984 | A |
| 4509017 | Andren et al. | Apr 1985 | A |
| 4511813 | Pan | Apr 1985 | A |
| 4580111 | Swanson | Apr 1986 | A |
| 4584541 | Nossen | Apr 1986 | A |
| 4605902 | Harrington | Aug 1986 | A |
| 4628286 | Nossen | Dec 1986 | A |
| 4682119 | Michel | Jul 1987 | A |
| 4682149 | Larson | Jul 1987 | A |
| 4686448 | Jones et al. | Aug 1987 | A |
| 4687999 | Desperben et al. | Aug 1987 | A |
| 4701716 | Poole | Oct 1987 | A |
| 4717894 | Edwards et al. | Jan 1988 | A |
| 4743858 | Everard | May 1988 | A |
| 4780803 | Dede Garcia-Santamaria | Oct 1988 | A |
| 4816783 | Leitch | Mar 1989 | A |
| 4817116 | Akaiwa et al. | Mar 1989 | A |
| 4873492 | Myer | Oct 1989 | A |
| 4951303 | Larson | Aug 1990 | A |
| 4974236 | Gurcan et al. | Nov 1990 | A |
| 4995055 | Weinberger et al. | Feb 1991 | A |
| 5005419 | O'Donnell et al. | Apr 1991 | A |
| 5012200 | Meinzer | Apr 1991 | A |
| 5017888 | Meinzer | May 1991 | A |
| 5077539 | Howatt | Dec 1991 | A |
| 5081673 | Engelke et al. | Jan 1992 | A |
| 5093636 | Higgins, Jr. et al. | Mar 1992 | A |
| 5115203 | Krett et al. | May 1992 | A |
| 5124665 | McGann | Jun 1992 | A |
| 5164678 | Puri et al. | Nov 1992 | A |
| 5214670 | Ballatore | May 1993 | A |
| 5229735 | Quan | Jul 1993 | A |
| 5239275 | Leitch | Aug 1993 | A |
| 5239686 | Downey | Aug 1993 | A |
| 5264807 | Okubo et al. | Nov 1993 | A |
| 5287069 | Okubo et al. | Feb 1994 | A |
| 5302914 | Arntz et al. | Apr 1994 | A |
| 5304943 | Koontz | Apr 1994 | A |
| 5307069 | Evans | Apr 1994 | A |
| 5345189 | Hornak et al. | Sep 1994 | A |
| 5351288 | Engelke et al. | Sep 1994 | A |
| 5365187 | Hornak et al. | Nov 1994 | A |
| 5365190 | Yu et al. | Nov 1994 | A |
| 5404114 | Sager | Apr 1995 | A |
| 5410280 | Linguet et al. | Apr 1995 | A |
| 5420541 | Upton et al. | May 1995 | A |
| 5426641 | Afrashteh et al. | Jun 1995 | A |
| 5432473 | Mattila et al. | Jul 1995 | A |
| 5438591 | Oie et al. | Aug 1995 | A |
| 5485120 | Anvari | Jan 1996 | A |
| 5490172 | Komara | Feb 1996 | A |
| 5495500 | Jovanovich et al. | Feb 1996 | A |
| 5508657 | Behan | Apr 1996 | A |
| 5515068 | Uragami et al. | May 1996 | A |
| 5530722 | Dent | Jun 1996 | A |
| 5541554 | Stengel et al. | Jul 1996 | A |
| 5554865 | Larson | Sep 1996 | A |
| 5559471 | Black | Sep 1996 | A |
| 5568088 | Dent et al. | Oct 1996 | A |
| 5574967 | Dent et al. | Nov 1996 | A |
| 5574992 | Cygan et al. | Nov 1996 | A |
| 5612651 | Chethik | Mar 1997 | A |
| 5621351 | Puri et al. | Apr 1997 | A |
| 5631604 | Dent et al. | May 1997 | A |
| RE35536 | Irissou et al. | Jun 1997 | E |
| 5638024 | Dent et al. | Jun 1997 | A |
| 5694433 | Dent | Dec 1997 | A |
| 5697074 | Makikallio et al. | Dec 1997 | A |
| 5710520 | Frey | Jan 1998 | A |
| 5719527 | Bateman et al. | Feb 1998 | A |
| 5724005 | Chen et al. | Mar 1998 | A |
| 5732334 | Miyake | Mar 1998 | A |
| 5739723 | Sigmon et al. | Apr 1998 | A |
| 5757229 | Mitzlaff | May 1998 | A |
| 5764704 | Shenoi | Jun 1998 | A |
| 5767750 | Yamaji | Jun 1998 | A |
| 5770971 | McNicol | Jun 1998 | A |
| 5784412 | Ichihara | Jul 1998 | A |
| 5784689 | Kobayashi | Jul 1998 | A |
| 5786727 | Sigmon | Jul 1998 | A |
| 5792956 | Li | Aug 1998 | A |
| 5805640 | O'Dea et al. | Sep 1998 | A |
| 5815531 | Dent | Sep 1998 | A |
| 5835128 | Macdonald et al. | Nov 1998 | A |
| 5841876 | Gifford et al. | Nov 1998 | A |
| 5854571 | Pinckley et al. | Dec 1998 | A |
| 5862460 | Rich | Jan 1999 | A |
| 5872481 | Sevic et al. | Feb 1999 | A |
| 5877643 | Drogi | Mar 1999 | A |
| 5880633 | Leizerovich et al. | Mar 1999 | A |
| 5886573 | Kolanek | Mar 1999 | A |
| 5886575 | Long | Mar 1999 | A |
| 5890051 | Schlang et al. | Mar 1999 | A |
| 5892394 | Wu | Apr 1999 | A |
| 5892395 | Stengel et al. | Apr 1999 | A |
| 5901346 | Stengel et al. | May 1999 | A |
| 5903854 | Abe et al. | May 1999 | A |
| 5933766 | Dent | Aug 1999 | A |
| 5949283 | Proctor et al. | Sep 1999 | A |
| 5952947 | Nussbaum et al. | Sep 1999 | A |
| 5956097 | Nguyen et al. | Sep 1999 | A |
| 5963091 | Chen et al. | Oct 1999 | A |
| 5973559 | Alberty | Oct 1999 | A |
| 5973568 | Shapiro et al. | Oct 1999 | A |
| 5974041 | Kornfeld et al. | Oct 1999 | A |
| 5990734 | Wright et al. | Nov 1999 | A |
| 5990738 | Wright et al. | Nov 1999 | A |
| 5999046 | Kotzamanis | Dec 1999 | A |
| 6011830 | Sasin et al. | Jan 2000 | A |
| 6026286 | Long | Feb 2000 | A |
| 6028485 | Sigmon et al. | Feb 2000 | A |
| 6043707 | Budnik | Mar 2000 | A |
| 6054894 | Wright et al. | Apr 2000 | A |
| 6054896 | Wright et al. | Apr 2000 | A |
| 6057798 | Burrier et al. | May 2000 | A |
| 6069525 | Sevic et al. | May 2000 | A |
| 6085074 | Cygan | Jul 2000 | A |
| 6097252 | Sigmon et al. | Aug 2000 | A |
| 6104991 | Newland et al. | Aug 2000 | A |
| 6111461 | Matsuno | Aug 2000 | A |
| 6111462 | Mucenieks et al. | Aug 2000 | A |
| 6115368 | Schilling | Sep 2000 | A |
| 6125266 | Matero et al. | Sep 2000 | A |
| 6130910 | Anderson et al. | Oct 2000 | A |
| 6130916 | Thomson | Oct 2000 | A |
| 6133788 | Dent | Oct 2000 | A |
| 6133789 | Braithwaite | Oct 2000 | A |
| 6137355 | Sevic et al. | Oct 2000 | A |
| 6147553 | Kolanek | Nov 2000 | A |
| 6154093 | Chen et al. | Nov 2000 | A |
| 6157253 | Sigmon et al. | Dec 2000 | A |
| 6169455 | Yamaguchi | Jan 2001 | B1 |
| 6175747 | Tanishima et al. | Jan 2001 | B1 |
| 6181199 | Camp, Jr. et al. | Jan 2001 | B1 |
| 6188277 | Borodulin et al. | Feb 2001 | B1 |
| 6198416 | Velazquez | Mar 2001 | B1 |
| 6201452 | Dent et al. | Mar 2001 | B1 |
| 6204735 | Cairns | Mar 2001 | B1 |
| 6215354 | Kolanek et al. | Apr 2001 | B1 |
| 6232838 | Sugimoto | May 2001 | B1 |
| 6236688 | Ohta et al. | May 2001 | B1 |
| 6242975 | Eidson et al. | Jun 2001 | B1 |
| 6246286 | Persson | Jun 2001 | B1 |
| 6246599 | Jang et al. | Jun 2001 | B1 |
| 6252461 | Raab | Jun 2001 | B1 |
| 6256482 | Raab | Jul 2001 | B1 |
| 6259320 | Valk et al. | Jul 2001 | B1 |
| 6285251 | Dent et al. | Sep 2001 | B1 |
| 6292054 | Ma et al. | Sep 2001 | B1 |
| 6295442 | Camp, Jr. et al. | Sep 2001 | B1 |
| 6300828 | McInnis | Oct 2001 | B1 |
| 6304545 | Armbruster et al. | Oct 2001 | B1 |
| 6307894 | Eidson et al. | Oct 2001 | B2 |
| 6311045 | Domokos | Oct 2001 | B1 |
| 6311046 | Dent | Oct 2001 | B1 |
| 6313703 | Wright et al. | Nov 2001 | B1 |
| 6337599 | Lee | Jan 2002 | B2 |
| 6342812 | Abdollahian et al. | Jan 2002 | B1 |
| 6349216 | Alberth, Jr. et al. | Feb 2002 | B1 |
| 6351189 | Hirvilampi | Feb 2002 | B1 |
| 6359506 | Camp, Jr. et al. | Mar 2002 | B1 |
| 6359508 | Mucenieks | Mar 2002 | B1 |
| 6359513 | Kuo et al. | Mar 2002 | B1 |
| 6366177 | McCune et al. | Apr 2002 | B1 |
| 6369651 | Dent | Apr 2002 | B1 |
| 6373901 | O'Dea et al. | Apr 2002 | B1 |
| 6373902 | Park et al. | Apr 2002 | B1 |
| 6374092 | Leizerovich et al. | Apr 2002 | B1 |
| 6380802 | Pehike et al. | Apr 2002 | B1 |
| 6384680 | Takei et al. | May 2002 | B1 |
| 6384681 | Bonds | May 2002 | B1 |
| 6385439 | Hellberg | May 2002 | B1 |
| 6388513 | Wright et al. | May 2002 | B1 |
| 6392483 | Suzuki et al. | May 2002 | B2 |
| 6396341 | Pehlke | May 2002 | B1 |
| 6396347 | Lie et al. | May 2002 | B1 |
| 6404823 | Grange et al. | Jun 2002 | B1 |
| 6407635 | Mucenieks et al. | Jun 2002 | B2 |
| 6411655 | Holden et al. | Jun 2002 | B1 |
| 6421389 | Jett et al. | Jul 2002 | B1 |
| 6424216 | Mu et al. | Jul 2002 | B2 |
| 6434122 | Barabash et al. | Aug 2002 | B2 |
| 6437644 | Kenington | Aug 2002 | B1 |
| 6449465 | Gailus et al. | Sep 2002 | B1 |
| 6452446 | Eisenberg et al. | Sep 2002 | B1 |
| 6459334 | Wright et al. | Oct 2002 | B2 |
| 6459337 | Goren et al. | Oct 2002 | B1 |
| 6462617 | Kim | Oct 2002 | B1 |
| 6469581 | Kobayashi | Oct 2002 | B1 |
| 6470431 | Nicosia et al. | Oct 2002 | B2 |
| 6472934 | Pehlke | Oct 2002 | B1 |
| 6472937 | Gerard et al. | Oct 2002 | B1 |
| 6476670 | Wright et al. | Nov 2002 | B1 |
| 6496062 | Nitz et al. | Dec 2002 | B1 |
| 6501331 | Adar | Dec 2002 | B2 |
| 6504428 | Cova et al. | Jan 2003 | B2 |
| 6504447 | Laney et al. | Jan 2003 | B1 |
| 6507731 | Hasegawa | Jan 2003 | B1 |
| 6510309 | Thompson et al. | Jan 2003 | B1 |
| 6510310 | Muralidharan | Jan 2003 | B1 |
| 6522194 | Pehlke | Feb 2003 | B1 |
| 6522198 | Ahn | Feb 2003 | B2 |
| 6522201 | Hsiao et al. | Feb 2003 | B1 |
| 6525605 | Hu et al. | Feb 2003 | B2 |
| 6529773 | Dewan | Mar 2003 | B1 |
| 6531935 | Russat et al. | Mar 2003 | B1 |
| 6535060 | Goren et al. | Mar 2003 | B2 |
| 6538509 | Ren | Mar 2003 | B2 |
| 6538793 | Rosenberg et al. | Mar 2003 | B2 |
| 6545535 | Andre | Apr 2003 | B2 |
| 6552634 | Raab | Apr 2003 | B1 |
| 6566944 | Pehlke et al. | May 2003 | B1 |
| 6577199 | Dent | Jun 2003 | B2 |
| 6577691 | Richards et al. | Jun 2003 | B2 |
| 6583679 | Cox et al. | Jun 2003 | B1 |
| 6583739 | Kenington | Jun 2003 | B1 |
| 6586995 | Tachibana | Jul 2003 | B1 |
| 6587010 | Wagh et al. | Jul 2003 | B2 |
| 6587511 | Barak et al. | Jul 2003 | B2 |
| 6587514 | Wright et al. | Jul 2003 | B1 |
| 6587913 | Campanale et al. | Jul 2003 | B2 |
| 6593806 | Melanson | Jul 2003 | B1 |
| 6600368 | Kim | Jul 2003 | B2 |
| 6603352 | Wight | Aug 2003 | B2 |
| 6606483 | Baker et al. | Aug 2003 | B1 |
| 6614854 | Chow et al. | Sep 2003 | B1 |
| 6622198 | Jones, Jr. | Sep 2003 | B2 |
| 6624694 | Ma et al. | Sep 2003 | B2 |
| 6633200 | Kolanek | Oct 2003 | B2 |
| 6636112 | McCune | Oct 2003 | B1 |
| 6637030 | Klein | Oct 2003 | B1 |
| 6646505 | Anderson | Nov 2003 | B2 |
| 6647073 | Tapio | Nov 2003 | B2 |
| 6653896 | Sevic et al. | Nov 2003 | B2 |
| 6672167 | Buell et al. | Jan 2004 | B2 |
| 6674326 | Hiramoto et al. | Jan 2004 | B1 |
| 6678041 | Kimura et al. | Jan 2004 | B2 |
| 6681101 | Eidson et al. | Jan 2004 | B1 |
| 6683918 | Jackson et al. | Jan 2004 | B2 |
| 6690232 | Ueno et al. | Feb 2004 | B2 |
| 6690233 | Sander | Feb 2004 | B2 |
| 6697436 | Wright et al. | Feb 2004 | B1 |
| 6697603 | Lovinggood et al. | Feb 2004 | B1 |
| 6700440 | Hareyama | Mar 2004 | B2 |
| 6700441 | Zhang et al. | Mar 2004 | B1 |
| 6700453 | Heiskala et al. | Mar 2004 | B2 |
| 6701419 | Tomaiuolo et al. | Mar 2004 | B2 |
| 6707338 | Kenington et al. | Mar 2004 | B2 |
| 6714776 | Birleson | Mar 2004 | B1 |
| 6724252 | Ngo et al. | Apr 2004 | B2 |
| 6735424 | Larson et al. | May 2004 | B1 |
| 6737914 | Gu | May 2004 | B2 |
| 6737916 | Luu | May 2004 | B2 |
| 6741840 | Nagode et al. | May 2004 | B2 |
| 6741867 | Tetsuya | May 2004 | B1 |
| 6750707 | Takei et al. | Jun 2004 | B2 |
| 6751265 | Schell et al. | Jun 2004 | B1 |
| 6765519 | Karlquist | Jul 2004 | B2 |
| 6781534 | Karlquist | Aug 2004 | B2 |
| 6784732 | Hajimiri et al. | Aug 2004 | B2 |
| 6784837 | Revankar et al. | Aug 2004 | B2 |
| 6785342 | Isaksen et al. | Aug 2004 | B1 |
| 6791408 | Goren et al. | Sep 2004 | B2 |
| 6791410 | Kim et al. | Sep 2004 | B2 |
| 6794934 | Betti-Berutto et al. | Sep 2004 | B2 |
| 6794938 | Weldon | Sep 2004 | B2 |
| 6798843 | Wright et al. | Sep 2004 | B1 |
| 6801086 | Chandrasekaran | Oct 2004 | B1 |
| 6801567 | Schmidl et al. | Oct 2004 | B1 |
| 6806767 | Dow | Oct 2004 | B2 |
| 6806789 | Bawell et al. | Oct 2004 | B2 |
| 6819171 | Kenington | Nov 2004 | B2 |
| 6819176 | Lee | Nov 2004 | B1 |
| 6819720 | Willetts | Nov 2004 | B1 |
| 6825719 | Barak et al. | Nov 2004 | B1 |
| 6829471 | White et al. | Dec 2004 | B2 |
| 6831491 | Karlquist | Dec 2004 | B2 |
| 6834183 | Black et al. | Dec 2004 | B2 |
| 6836183 | Wight | Dec 2004 | B2 |
| 6838942 | Somerville et al. | Jan 2005 | B1 |
| 6842070 | Nilsson | Jan 2005 | B2 |
| 6847266 | Laney et al. | Jan 2005 | B2 |
| 6853244 | Robinson et al. | Feb 2005 | B2 |
| 6853247 | Weldon | Feb 2005 | B2 |
| 6853248 | Weldon | Feb 2005 | B2 |
| 6859098 | Husseini | Feb 2005 | B2 |
| 6864742 | Kobayashi | Mar 2005 | B2 |
| 6867647 | Wouters | Mar 2005 | B2 |
| 6873211 | Thompson et al. | Mar 2005 | B1 |
| 6879209 | Grundlingh | Apr 2005 | B2 |
| 6882217 | Mueller | Apr 2005 | B1 |
| 6882711 | Nicol | Apr 2005 | B1 |
| 6882829 | Mostov et al. | Apr 2005 | B2 |
| 6889034 | Dent | May 2005 | B1 |
| 6891432 | Nagle et al. | May 2005 | B2 |
| 6900694 | Suzuki et al. | May 2005 | B2 |
| 6906585 | Weldon | Jun 2005 | B2 |
| 6914487 | Doyle et al. | Jul 2005 | B1 |
| 6917244 | Rosnell et al. | Jul 2005 | B2 |
| 6917389 | Lee | Jul 2005 | B2 |
| 6924699 | Ahmed | Aug 2005 | B2 |
| 6928272 | Doi | Aug 2005 | B2 |
| 6930547 | Chandrasekaran et al. | Aug 2005 | B2 |
| 6937096 | Wight et al. | Aug 2005 | B2 |
| 6937102 | Lopez et al. | Aug 2005 | B2 |
| 6940349 | Hellberg | Sep 2005 | B2 |
| 6943624 | Ohnishi et al. | Sep 2005 | B2 |
| 6947713 | Checoury et al. | Sep 2005 | B2 |
| 6960956 | Pehlke et al. | Nov 2005 | B2 |
| 6970040 | Dening | Nov 2005 | B1 |
| 6975177 | Varis et al. | Dec 2005 | B2 |
| 6980780 | Chen et al. | Dec 2005 | B2 |
| 6987954 | Nielsen | Jan 2006 | B2 |
| 6990323 | Prikhodko et al. | Jan 2006 | B2 |
| 6993301 | Kenington et al. | Jan 2006 | B1 |
| 7010276 | Sander et al. | Mar 2006 | B2 |
| 7015752 | Saed | Mar 2006 | B2 |
| 7023272 | Hung et al. | Apr 2006 | B2 |
| 7026871 | Saèd | Apr 2006 | B2 |
| 7030714 | Korol | Apr 2006 | B2 |
| 7031382 | Hessel et al. | Apr 2006 | B2 |
| 7034613 | Saèd | Apr 2006 | B2 |
| 7035607 | Lim et al. | Apr 2006 | B2 |
| 7042283 | Suzuki et al. | May 2006 | B2 |
| 7042286 | Meade et al. | May 2006 | B2 |
| 7043208 | Nigra | May 2006 | B2 |
| 7043213 | Robinson et al. | May 2006 | B2 |
| 7054296 | Sorrells et al. | May 2006 | B1 |
| 7054597 | Rosnell | May 2006 | B2 |
| 7057461 | Canilao et al. | Jun 2006 | B1 |
| 7064607 | Maclean et al. | Jun 2006 | B2 |
| 7068099 | Versteegen | Jun 2006 | B2 |
| 7068101 | Saèd et al. | Jun 2006 | B2 |
| 7068103 | Lind | Jun 2006 | B2 |
| 7071774 | Hellberg | Jul 2006 | B2 |
| 7071777 | McBeath et al. | Jul 2006 | B2 |
| 7078976 | Blednov | Jul 2006 | B2 |
| 7081795 | Matsuura et al. | Jul 2006 | B2 |
| 7084702 | Ichitsubo et al. | Aug 2006 | B1 |
| 7088970 | Williams | Aug 2006 | B2 |
| 7091775 | Ichitsubo et al. | Aug 2006 | B2 |
| 7091777 | Lynch | Aug 2006 | B2 |
| 7092675 | Lim et al. | Aug 2006 | B2 |
| 7092676 | Abdelgany et al. | Aug 2006 | B2 |
| 7099382 | Aronson et al. | Aug 2006 | B2 |
| 7103328 | Zelley | Sep 2006 | B2 |
| 7139535 | Zschunke | Nov 2006 | B2 |
| 7145397 | Yamamoto et al. | Dec 2006 | B2 |
| 7173980 | Masenten et al. | Feb 2007 | B2 |
| 7177418 | Maclean et al. | Feb 2007 | B2 |
| 7184723 | Sorrells et al. | Feb 2007 | B2 |
| 7193459 | Epperson et al. | Mar 2007 | B1 |
| 7197284 | Brandt et al. | Mar 2007 | B2 |
| 7200369 | Kim et al. | Apr 2007 | B2 |
| 7230996 | Matsuura et al. | Jun 2007 | B2 |
| 7242245 | Burns et al. | Jul 2007 | B2 |
| 7260368 | Blumer | Aug 2007 | B1 |
| 7260369 | Feher | Aug 2007 | B2 |
| 7292189 | Orr et al. | Nov 2007 | B2 |
| 7327803 | Sorrells et al. | Feb 2008 | B2 |
| 7345534 | Grebennikov | Mar 2008 | B2 |
| 7349673 | Moloudi et al. | Mar 2008 | B2 |
| 7355470 | Sorrells et al. | Apr 2008 | B2 |
| 7378902 | Sorrells et al. | May 2008 | B2 |
| 7403579 | Jaffe et al. | Jul 2008 | B2 |
| 7414469 | Sorrells et al. | Aug 2008 | B2 |
| 7421036 | Sorrells et al. | Sep 2008 | B2 |
| 7423477 | Sorrells et al. | Sep 2008 | B2 |
| 7428230 | Park | Sep 2008 | B2 |
| 7436894 | Norris | Oct 2008 | B2 |
| 7440733 | Maslennikov et al. | Oct 2008 | B2 |
| 7459893 | Jacobs | Dec 2008 | B2 |
| 7460612 | Eliezer et al. | Dec 2008 | B2 |
| 7466760 | Sorrells et al. | Dec 2008 | B2 |
| 7474695 | Liu et al. | Jan 2009 | B2 |
| 7486894 | Aronson et al. | Feb 2009 | B2 |
| 7502599 | Ben-Ayun et al. | Mar 2009 | B2 |
| 7509102 | Rofougaran et al. | Mar 2009 | B2 |
| 7526261 | Sorrells et al. | Apr 2009 | B2 |
| 7560984 | Akizuki et al. | Jul 2009 | B2 |
| 7616057 | Sutardja | Nov 2009 | B2 |
| 7620129 | Sorrells et al. | Nov 2009 | B2 |
| 7639072 | Sorrells et al. | Dec 2009 | B2 |
| 7647030 | Sorrells et al. | Jan 2010 | B2 |
| 7672648 | Groe et al. | Mar 2010 | B1 |
| 7672650 | Sorrells et al. | Mar 2010 | B2 |
| 7738853 | Eddy et al. | Jun 2010 | B2 |
| 7750733 | Sorrells et al. | Jul 2010 | B2 |
| RE41582 | Larson et al. | Aug 2010 | E |
| 7778320 | Agazzi et al. | Aug 2010 | B2 |
| 7835709 | Sorrells et al. | Nov 2010 | B2 |
| 7844235 | Sorrells et al. | Nov 2010 | B2 |
| 7885682 | Sorrells et al. | Feb 2011 | B2 |
| 7907671 | Klomsdorf et al. | Mar 2011 | B2 |
| 7911272 | Sorrells et al. | Mar 2011 | B2 |
| 7929989 | Sorrells et al. | Apr 2011 | B2 |
| 7932776 | Sorrells et al. | Apr 2011 | B2 |
| 7937106 | Sorrells et al. | May 2011 | B2 |
| 7945224 | Sorrells et al. | May 2011 | B2 |
| 7949365 | Sorrells et al. | May 2011 | B2 |
| 7978390 | Kikuchi | Jul 2011 | B2 |
| 8013675 | Sorrells et al. | Sep 2011 | B2 |
| 8026764 | Sorrells et al. | Sep 2011 | B2 |
| 8031804 | Sorrells et al. | Oct 2011 | B2 |
| 8036306 | Sorrells et al. | Oct 2011 | B2 |
| 8050353 | Sorrells et al. | Nov 2011 | B2 |
| 8059749 | Sorrells et al. | Nov 2011 | B2 |
| 8073078 | Kaczman et al. | Dec 2011 | B2 |
| 8170081 | Forenza et al. | May 2012 | B2 |
| 8233858 | Sorrells et al. | Jul 2012 | B2 |
| 8280321 | Sorrells et al. | Oct 2012 | B2 |
| 8315336 | Sorrells et al. | Nov 2012 | B2 |
| 8334722 | Sorrells et al. | Dec 2012 | B2 |
| 8351870 | Sorrells et al. | Jan 2013 | B2 |
| 8355466 | Kleider et al. | Jan 2013 | B2 |
| 8369807 | Mikhemar et al. | Feb 2013 | B2 |
| 8384484 | Winslow et al. | Feb 2013 | B2 |
| 8406711 | Sorrells et al. | Mar 2013 | B2 |
| 8410849 | Sorrells et al. | Apr 2013 | B2 |
| 8428527 | Sorrells et al. | Apr 2013 | B2 |
| 8433264 | Sorrells et al. | Apr 2013 | B2 |
| 8433745 | Roger | Apr 2013 | B2 |
| 8447248 | Sorrells et al. | May 2013 | B2 |
| 8461924 | Rawlins et al. | Jun 2013 | B2 |
| 8502600 | Rawlins et al. | Aug 2013 | B2 |
| 8548093 | Sorrells et al. | Oct 2013 | B2 |
| 8577313 | Sorrells et al. | Nov 2013 | B2 |
| 8626093 | Sorrells et al. | Jan 2014 | B2 |
| 8639196 | Sorrells et al. | Jan 2014 | B2 |
| 8755454 | Sorrells et al. | Jun 2014 | B2 |
| 8766717 | Sorrells et al. | Jul 2014 | B2 |
| 20010001008 | Dent | May 2001 | A1 |
| 20010004373 | Hirata | Jun 2001 | A1 |
| 20010006354 | Lee | Jul 2001 | A1 |
| 20010006359 | Suzuki et al. | Jul 2001 | A1 |
| 20010030581 | Dent | Oct 2001 | A1 |
| 20010052816 | Ahn | Dec 2001 | A1 |
| 20020008577 | Cova et al. | Jan 2002 | A1 |
| 20020027958 | Kolanek | Mar 2002 | A1 |
| 20020042253 | Dartois | Apr 2002 | A1 |
| 20020047745 | Kolanek | Apr 2002 | A1 |
| 20020053973 | Ward, Jr. | May 2002 | A1 |
| 20020058486 | Persson | May 2002 | A1 |
| 20020071497 | Bengtsson et al. | Jun 2002 | A1 |
| 20020079962 | Sander | Jun 2002 | A1 |
| 20020084845 | Eisenberg et al. | Jul 2002 | A1 |
| 20020086707 | Struhsaker et al. | Jul 2002 | A1 |
| 20020094034 | Moriyama | Jul 2002 | A1 |
| 20020101907 | Dent et al. | Aug 2002 | A1 |
| 20020105378 | Tapio | Aug 2002 | A1 |
| 20020105384 | Dent | Aug 2002 | A1 |
| 20020125947 | Ren | Sep 2002 | A1 |
| 20020126769 | Jett et al. | Sep 2002 | A1 |
| 20020127986 | White et al. | Sep 2002 | A1 |
| 20020130716 | Larson et al. | Sep 2002 | A1 |
| 20020130727 | Nagasaka | Sep 2002 | A1 |
| 20020130729 | Larson et al. | Sep 2002 | A1 |
| 20020136275 | Wight | Sep 2002 | A1 |
| 20020136325 | Pehlke et al. | Sep 2002 | A1 |
| 20020146996 | Bachman, II et al. | Oct 2002 | A1 |
| 20020153950 | Kusunoki et al. | Oct 2002 | A1 |
| 20020159532 | Wight | Oct 2002 | A1 |
| 20020164965 | Chominski et al. | Nov 2002 | A1 |
| 20020168025 | Schwent et al. | Nov 2002 | A1 |
| 20020171478 | Wouters | Nov 2002 | A1 |
| 20020171485 | Cova | Nov 2002 | A1 |
| 20020172376 | Bizjak | Nov 2002 | A1 |
| 20020180547 | Staszewski et al. | Dec 2002 | A1 |
| 20020183021 | Brandt | Dec 2002 | A1 |
| 20020186079 | Kobayashi | Dec 2002 | A1 |
| 20020191638 | Wang et al. | Dec 2002 | A1 |
| 20020196864 | Booth et al. | Dec 2002 | A1 |
| 20030006845 | Lopez et al. | Jan 2003 | A1 |
| 20030031268 | Wight | Feb 2003 | A1 |
| 20030041667 | White | Mar 2003 | A1 |
| 20030083026 | Liu | May 2003 | A1 |
| 20030087625 | Conti | May 2003 | A1 |
| 20030098753 | Wagh et al. | May 2003 | A1 |
| 20030102910 | Sevic et al. | Jun 2003 | A1 |
| 20030102914 | Kenington et al. | Jun 2003 | A1 |
| 20030107435 | Gu | Jun 2003 | A1 |
| 20030114124 | Higuchi | Jun 2003 | A1 |
| 20030118121 | Makinen | Jun 2003 | A1 |
| 20030119526 | Edge | Jun 2003 | A1 |
| 20030123566 | Hasson | Jul 2003 | A1 |
| 20030125065 | Barak et al. | Jul 2003 | A1 |
| 20030132800 | Kenington | Jul 2003 | A1 |
| 20030143967 | Ciccarelli et al. | Jul 2003 | A1 |
| 20030179041 | Weldon | Sep 2003 | A1 |
| 20030190895 | Mostov et al. | Oct 2003 | A1 |
| 20030201835 | Dening et al. | Oct 2003 | A1 |
| 20030210096 | Pengelly et al. | Nov 2003 | A1 |
| 20030210746 | Asbeck et al. | Nov 2003 | A1 |
| 20030219067 | Birkett et al. | Nov 2003 | A1 |
| 20030220086 | Birkett | Nov 2003 | A1 |
| 20030228856 | Orihashi et al. | Dec 2003 | A1 |
| 20030231057 | Hiramoto et al. | Dec 2003 | A1 |
| 20040008081 | Friedel et al. | Jan 2004 | A1 |
| 20040021517 | Irvine et al. | Feb 2004 | A1 |
| 20040025104 | Amer | Feb 2004 | A1 |
| 20040027198 | Chandrasekaran et al. | Feb 2004 | A1 |
| 20040037363 | Norsworthy et al. | Feb 2004 | A1 |
| 20040037378 | Komori et al. | Feb 2004 | A1 |
| 20040046524 | Zschunke | Mar 2004 | A1 |
| 20040052312 | Matero | Mar 2004 | A1 |
| 20040056723 | Gotou | Mar 2004 | A1 |
| 20040062397 | Amer | Apr 2004 | A1 |
| 20040075492 | Wight | Apr 2004 | A1 |
| 20040076238 | Parker et al. | Apr 2004 | A1 |
| 20040085134 | Griffith et al. | May 2004 | A1 |
| 20040092281 | Burchfiel | May 2004 | A1 |
| 20040095192 | Krvavac | May 2004 | A1 |
| 20040101065 | Hagh et al. | May 2004 | A1 |
| 20040108896 | Midtgaard | Jun 2004 | A1 |
| 20040113698 | Kim et al. | Jun 2004 | A1 |
| 20040119514 | Karlquist | Jun 2004 | A1 |
| 20040119622 | Karlquist | Jun 2004 | A1 |
| 20040119624 | Karlquist | Jun 2004 | A1 |
| 20040125006 | Tani et al. | Jul 2004 | A1 |
| 20040131131 | Peach et al. | Jul 2004 | A1 |
| 20040135630 | Hellberg | Jul 2004 | A1 |
| 20040142667 | Lochhead et al. | Jul 2004 | A1 |
| 20040146116 | Kang et al. | Jul 2004 | A1 |
| 20040166813 | Mann et al. | Aug 2004 | A1 |
| 20040169559 | Weldon | Sep 2004 | A1 |
| 20040172583 | Amer | Sep 2004 | A1 |
| 20040174213 | Thompson | Sep 2004 | A1 |
| 20040181745 | Amer | Sep 2004 | A1 |
| 20040184559 | Ballantyne | Sep 2004 | A1 |
| 20040185805 | Kim et al. | Sep 2004 | A1 |
| 20040189380 | Myer et al. | Sep 2004 | A1 |
| 20040189381 | Louis | Sep 2004 | A1 |
| 20040196899 | Zhou et al. | Oct 2004 | A1 |
| 20040198263 | Ode et al. | Oct 2004 | A1 |
| 20040222851 | Weldon | Nov 2004 | A1 |
| 20040224715 | Rosenlof et al. | Nov 2004 | A1 |
| 20040227570 | Jackson et al. | Nov 2004 | A1 |
| 20040233599 | Busking | Nov 2004 | A1 |
| 20040246060 | Varis et al. | Dec 2004 | A1 |
| 20040251962 | Rosnell et al. | Dec 2004 | A1 |
| 20040263242 | Hellberg | Dec 2004 | A1 |
| 20040263245 | Winter et al. | Dec 2004 | A1 |
| 20040263246 | Robinson et al. | Dec 2004 | A1 |
| 20040266059 | Wight et al. | Dec 2004 | A1 |
| 20040266365 | Hasson et al. | Dec 2004 | A1 |
| 20040266368 | Rosnell | Dec 2004 | A1 |
| 20040266374 | Saed et al. | Dec 2004 | A1 |
| 20040267399 | Funk | Dec 2004 | A1 |
| 20050001674 | Saed et al. | Jan 2005 | A1 |
| 20050001675 | Saed | Jan 2005 | A1 |
| 20050001676 | Saed | Jan 2005 | A1 |
| 20050001677 | Saed | Jan 2005 | A1 |
| 20050001678 | Saed | Jan 2005 | A1 |
| 20050001679 | Saed | Jan 2005 | A1 |
| 20050002470 | Saed et al. | Jan 2005 | A1 |
| 20050003770 | Saed | Jan 2005 | A1 |
| 20050007194 | Grundlingh | Jan 2005 | A1 |
| 20050012547 | Kwon et al. | Jan 2005 | A1 |
| 20050018787 | Saed | Jan 2005 | A1 |
| 20050024262 | Cantrell et al. | Feb 2005 | A1 |
| 20050025181 | Nazari | Feb 2005 | A1 |
| 20050047038 | Nakajima et al. | Mar 2005 | A1 |
| 20050058059 | Amer | Mar 2005 | A1 |
| 20050058193 | Saed | Mar 2005 | A1 |
| 20050058209 | Magrath | Mar 2005 | A1 |
| 20050058227 | Birkett et al. | Mar 2005 | A1 |
| 20050058228 | Birkett | Mar 2005 | A1 |
| 20050073360 | Johnson et al. | Apr 2005 | A1 |
| 20050073374 | Korol | Apr 2005 | A1 |
| 20050088226 | Robinson et al. | Apr 2005 | A1 |
| 20050110590 | Korol | May 2005 | A1 |
| 20050111574 | Muller et al. | May 2005 | A1 |
| 20050118973 | Khlat | Jun 2005 | A1 |
| 20050129140 | Robinson | Jun 2005 | A1 |
| 20050129141 | Lee | Jun 2005 | A1 |
| 20050136864 | Zipper | Jun 2005 | A1 |
| 20050181746 | Wight | Aug 2005 | A1 |
| 20050191976 | Shakeshaft et al. | Sep 2005 | A1 |
| 20050195031 | Grundlingh | Sep 2005 | A1 |
| 20050201483 | Coersmeier | Sep 2005 | A1 |
| 20050215206 | Granstrom et al. | Sep 2005 | A1 |
| 20050227646 | Yamazaki et al. | Oct 2005 | A1 |
| 20050242879 | Muller | Nov 2005 | A1 |
| 20050253652 | Song et al. | Nov 2005 | A1 |
| 20050253745 | Song et al. | Nov 2005 | A1 |
| 20050260956 | Loraine et al. | Nov 2005 | A1 |
| 20060006946 | Burns et al. | Jan 2006 | A1 |
| 20060017500 | Hellberg | Jan 2006 | A1 |
| 20060035618 | Pleasant | Feb 2006 | A1 |
| 20060052068 | Sander et al. | Mar 2006 | A1 |
| 20060052124 | Pottenger et al. | Mar 2006 | A1 |
| 20060055458 | Shiikuma et al. | Mar 2006 | A1 |
| 20060066396 | Brandt | Mar 2006 | A1 |
| 20060068707 | Greeley | Mar 2006 | A1 |
| 20060088081 | Withington et al. | Apr 2006 | A1 |
| 20060142821 | Bange et al. | Jun 2006 | A1 |
| 20060160502 | Kintis | Jul 2006 | A1 |
| 20060220625 | Chapuis | Oct 2006 | A1 |
| 20060238245 | Carichner et al. | Oct 2006 | A1 |
| 20060262889 | Kalvaitis et al. | Nov 2006 | A1 |
| 20060264190 | Aleiner | Nov 2006 | A1 |
| 20060291589 | Eliezer et al. | Dec 2006 | A1 |
| 20060292999 | Sorrells et al. | Dec 2006 | A1 |
| 20060293000 | Sorrells et al. | Dec 2006 | A1 |
| 20070019757 | Matero | Jan 2007 | A1 |
| 20070021080 | Kuriyama et al. | Jan 2007 | A1 |
| 20070030063 | Izumi et al. | Feb 2007 | A1 |
| 20070050758 | Arevalo et al. | Mar 2007 | A1 |
| 20070071114 | Sanderford et al. | Mar 2007 | A1 |
| 20070076814 | Ikeda et al. | Apr 2007 | A1 |
| 20070082630 | Aridas et al. | Apr 2007 | A1 |
| 20070087708 | Sorrells et al. | Apr 2007 | A1 |
| 20070087709 | Sorrells et al. | Apr 2007 | A1 |
| 20070090874 | Sorrells et al. | Apr 2007 | A1 |
| 20070096806 | Sorrells et al. | May 2007 | A1 |
| 20070111686 | Lee | May 2007 | A1 |
| 20070127563 | Wu et al. | Jun 2007 | A1 |
| 20070155344 | Wiessner et al. | Jul 2007 | A1 |
| 20070184790 | Gilberton et al. | Aug 2007 | A1 |
| 20070190952 | Waheed et al. | Aug 2007 | A1 |
| 20070194986 | Dulmovits, Jr. et al. | Aug 2007 | A1 |
| 20070218852 | Huynh | Sep 2007 | A1 |
| 20070247217 | Sorrells et al. | Oct 2007 | A1 |
| 20070247220 | Sorrells et al. | Oct 2007 | A1 |
| 20070247221 | Sorrells et al. | Oct 2007 | A1 |
| 20070248156 | Sorrells et al. | Oct 2007 | A1 |
| 20070248185 | Sorrells et al. | Oct 2007 | A1 |
| 20070248186 | Sorrells et al. | Oct 2007 | A1 |
| 20070249299 | Sorrells et al. | Oct 2007 | A1 |
| 20070249300 | Sorrells et al. | Oct 2007 | A1 |
| 20070249301 | Sorrells et al. | Oct 2007 | A1 |
| 20070249302 | Sorrells et al. | Oct 2007 | A1 |
| 20070249304 | Snelgrove et al. | Oct 2007 | A1 |
| 20070291668 | Duan | Dec 2007 | A1 |
| 20080019459 | Chen et al. | Jan 2008 | A1 |
| 20080072025 | Staszewski et al. | Mar 2008 | A1 |
| 20080089252 | Choi | Apr 2008 | A1 |
| 20080133982 | Rawlins et al. | Jun 2008 | A1 |
| 20080225929 | Proctor et al. | Sep 2008 | A1 |
| 20080225935 | Reddy | Sep 2008 | A1 |
| 20080259846 | Gonikberg et al. | Oct 2008 | A1 |
| 20080272841 | Sorrells et al. | Nov 2008 | A1 |
| 20080299913 | Han et al. | Dec 2008 | A1 |
| 20080311860 | Tanaka et al. | Dec 2008 | A1 |
| 20090004981 | Eliezer et al. | Jan 2009 | A1 |
| 20090070568 | Shi et al. | Mar 2009 | A1 |
| 20090091384 | Sorrells et al. | Apr 2009 | A1 |
| 20090134947 | Tarng | May 2009 | A1 |
| 20090201084 | See et al. | Aug 2009 | A1 |
| 20090227214 | Georgantas et al. | Sep 2009 | A1 |
| 20090238249 | Van Waasen et al. | Sep 2009 | A1 |
| 20090262861 | Nielsen | Oct 2009 | A1 |
| 20090262877 | Shi et al. | Oct 2009 | A1 |
| 20100103052 | Ying | Apr 2010 | A1 |
| 20100311353 | Teillet et al. | Dec 2010 | A1 |
| 20110300885 | Darabi et al. | Dec 2011 | A1 |
| 20120025624 | Lee et al. | Feb 2012 | A1 |
| 20120153731 | Kirby et al. | Jun 2012 | A9 |
| 20120256684 | Sorrells et al. | Oct 2012 | A1 |
| 20120263215 | Peng | Oct 2012 | A1 |
| 20130101074 | Hickling | Apr 2013 | A1 |
| Number | Date | Country |
|---|---|---|
| 0 011 464 | May 1980 | EP |
| 0 471 346 | Aug 1990 | EP |
| 0 630 104 | Dec 1994 | EP |
| 0 708 546 | Apr 1996 | EP |
| 0 471 346 | Nov 1996 | EP |
| 0 639 307 | Dec 1997 | EP |
| 0 821 304 | Jan 1998 | EP |
| 0 725 478 | Aug 1998 | EP |
| 0 892 529 | Jan 1999 | EP |
| 0 897 213 | Feb 1999 | EP |
| 0 598 585 | Mar 1999 | EP |
| 0 630 104 | Aug 2000 | EP |
| 0 821 304 | Feb 2002 | EP |
| 1 068 666 | May 2003 | EP |
| 1 381 154 | Jan 2004 | EP |
| 0 897 213 | Mar 2004 | EP |
| 1 487 100 | Dec 2004 | EP |
| 1 332 550 | Mar 2005 | EP |
| 1 142 250 | Apr 2005 | EP |
| 1 521 359 | Apr 2005 | EP |
| 1 583 228 | Oct 2005 | EP |
| 2159374 | Nov 1985 | GB |
| 2 267 402 | Dec 1993 | GB |
| 54-022749 | Feb 1979 | JP |
| 60-63517 | Apr 1985 | JP |
| 1-284106 | Nov 1989 | JP |
| 2-87708 | Mar 1990 | JP |
| 3-232307 | Oct 1991 | JP |
| 4-095409 | Mar 1992 | JP |
| 4-104604 | Apr 1992 | JP |
| 5-22046 | Jan 1993 | JP |
| 5-037263 | Feb 1993 | JP |
| 6-338728 | Dec 1994 | JP |
| 9-018536 | Jan 1997 | JP |
| 9-074320 | Mar 1997 | JP |
| 10-70451 | Mar 1998 | JP |
| 2000-209291 | Jul 2000 | JP |
| 2000-244261 | Sep 2000 | JP |
| 2001-136057 | May 2001 | JP |
| 2001-217659 | Aug 2001 | JP |
| 2001-308650 | Nov 2001 | JP |
| 2002-543729 | Dec 2002 | JP |
| 2003-298357 | Oct 2003 | JP |
| 2003-298361 | Oct 2003 | JP |
| H08-163189 | Oct 2003 | JP |
| 2004-260707 | Sep 2004 | JP |
| 2005-101940 | Apr 2005 | JP |
| 2005-151543 | Jun 2005 | JP |
| 102824 | Nov 1991 | RO |
| 100466 | Aug 1992 | RO |
| 1322183 | Jul 1987 | SU |
| WO 9421035 | Sep 1994 | WO |
| WO 9610310 | Apr 1996 | WO |
| WO 9619063 | Jun 1996 | WO |
| WO 9741642 | Nov 1997 | WO |
| WO 9748219 | Dec 1997 | WO |
| WO 9923755 | May 1999 | WO |
| WO 9952206 | Oct 1999 | WO |
| WO 0041371 | Jul 2000 | WO |
| WO 0067370 | Nov 2000 | WO |
| WO 0103292 | Jan 2001 | WO |
| WO 0145205 | Jun 2001 | WO |
| WO 0191282 | Nov 2001 | WO |
| WO 0239577 | May 2002 | WO |
| WO 02082633 | Oct 2002 | WO |
| WO 03047093 | Jun 2003 | WO |
| WO 03061115 | Jul 2003 | WO |
| WO 2004036736 | Mar 2004 | WO |
| WO 2004036736 | Apr 2004 | WO |
| WO 2004057755 | Jul 2004 | WO |
| WO 2005031966 | Apr 2005 | WO |
| WO 2005036732 | Apr 2005 | WO |
| WO 2005055413 | Jun 2005 | WO |
| Entry |
|---|
| “Ampliphase AM transmission system,” ABU Technical Review, No. 33, pp. 10-18 (Jul. 1974). |
| “Designing an SSB Outphaser,” Electronics World, pp. 306-310 (Apr. 1996). |
| “New 50 KW Ampliphase AM Transmitter,” RCA in Broadcast News, No. 111, pp. 36-39 (Jun. 1961). |
| ***The Ampliphase Page***: Ampliphase—A quick description . . . , Reproduction of text from http://rossrevenge.co.uk/tx/ampli.htm, 13 pages (visited Jan. 18, 2006). |
| Ajluni, C., “Chip Set Withstands WLAN's Future Blows,” at http://www.wsdmag.com/Articles/Print.cfm?ArticleID=6792, 5 pages (Oct. 2003). |
| Ampen-Darko, S. and Al-Raweshidy, H.S., “Gain/phase imbalance cancellation technique in LINC transmitters,” Electronics Letters, vol. 34, No. 22, pp. 2093-2094 (Oct. 29, 1988). |
| Ampen-Darko, S.O. and Al-Raweshidy, H.S., “A Novel Technique for Gain/Phase Cancellation in LINC Transmitters,” IEEE VTS—50th Vehicular Technology Conference, Amsterdam, pp. 2034-2038 (Sep. 19-22, 1999). |
| Andreani, P., Linear PA architectures (Chapter 13), available at http://server.oersted.dtu.dk/personal/pa/31636/pdf/paL.in.pdf, 10 pages (Jun. 14, 2007). |
| Ariyavisitakul, S. and Lie, T.P., “Characterizing the Effects of Nonlinear Amplifiers on Linear Modulation for Digital Portable Radio Communications,” IEEE Transactions on Vehicular Technology, vol. 39, No. 4, pp. 383-389 (Nov. 1990). |
| ARMMS—The RF and Microwave Society—Last Meeting, at http://www.armms.org/last.html, 4 pages (printed Apr. 14, 2005). |
| Asbeck, P.M. et al., “Power Amplifier Approaches for High Efficiency and Linearity,” in Itoh, T. et al. (eds.), RF Technologies for Low Power Wireless Communications, ISBN No. 0-471-38267-1, pp. 189-227 (2001). |
| Asbeck, P.M. et al., “Synergistic Design of DSP and Power Amplifiers for Wireless Communications,” IEEE Transactions on Microwave Theory and Techniques, vol. 49, No. 11, pp. 2163-2169 (Nov. 2001). |
| Banelli, P., “Error Sensitivity in Adaptive Predistortion Systems,” Global Telecommunications Conference—Globecom '99, pp. 883-888 (1999). |
| Bateman, A., et al., “The Application of Digital Signal Processing to Transmitter Linearisation,” EUROCON 88: 8th European Conference on Electrotechnics, pp. 64-67 (Jun. 13-17, 1988). |
| Bespalov, V.B. and Aslamazyan, A.S., “Broadband Strip-Line SHF Ampliphasemeter,” Measurement Techniques (Translated from Russian), vol. 25, No. 8, pp. 712-715 (Aug. 1982). |
| Birafane, A. and Kouki, A., “An Analytical Approach to LINC Power Combining Efficiency Estimation and Optimization,” 33rd European Microwave Conference—Munich, pp. 1227-1229 (2003). |
| Birafane, A. and Kouki, A., “Distortion Free LINC Amplifier with Chireix-Outphasing Combiner Using Phase-Only Predistortion,” 34th European Microwave Conference—Amsterdam, pp. 1069-1072 (2004). |
| Birafane, A. and Kouki, A., “On the Linearity and Efficiency of Outphasing Microwave Amplifiers,” IEEE Transactions on Microwave Theory and Techniques, vol. 52, No. 7, pp. 1702-1708 (Jul. 2004). |
| Birafane, A. and Kouki, A., “Sources of Linearity Degradation in LINC Transmitters for Hybrid and Outphasing Combiners,” Canadian Conference on Electrical and Computer Engineering—Niagara Falls, pp. 547-550 (May 2004). |
| Birafane, A. and Kouki, A.B., “Phase-Only Predistortion for LINC Amplifiers With Chireix-Outphasing Combiners,” IEEE Transactions on Microwave Theory and Techniques, vol. 53, No. 6, pp. 2240-2250 (Jun. 2005). |
| Breed, G., “Intermodulation Distortion Performance and Measurement Issues,” High Frequency Electronics, p. 56(2) (May 2003). |
| Bruckmann, H., “Modulation Arrangements and Operating Costs of Broadcasting and Radio-Telephony Transmitters,” Telegraphen-Fernsprech-Funk-und Fernsehtechnik, vol. 24, pp. 83-91 (Apr. 1935). |
| Burnill, J., “Transmitting AM,” Electronics World + Wireless World, pp. 58-60 (Jan. 1995). |
| Casadevall, F. and Olmos, J.J., “On the Behavior of the LINC Transmitter,” 40th IEEE Vehicular Technology Conference, pp. 29-34 (May 6-9, 1990). |
| Casadevall, F.J. and Valdovinos, A., “Performance Analysis of QAM Modulations Applied to the LINC Transmitter,” IEEE Transactions on Vehicular Technology, vol. 42, No. 4, pp. 399-406 (Nov. 1993). |
| Casadevall, F.J., “The LINC Transmitter”, RF Design, pp. 41-48 (Feb. 1990). |
| Cha, J. et al., “Highly Efficient Power Amplifier for CDMA Base Stations Using Doherty Configuration,” IEEE MTT-S International Microwave Symposium Digest, pp. 533-536 (2004). |
| Chan, K.Y. et al., “Analysis and Realisation of the LINC Transmitter using the Combined Analogue Locked Loop Universal Modulator (CALLUM),”IEEE 44th Vehicular Technology Conference, vol. 1, pp. 484-488 (Jun. 8-10, 1994). |
| Chen, J.-T. et al., “The Optimal RLS Parameter Tracking Algorithm for a Power Amplifier Feedforward Linearizer,” IEEE Transactions on Circuits and Systems-II: Analog and Digital Signal Processing, vol. 46, No. 4, pp. 464-468 (Apr. 1999). |
| Chireix, H., “High Power Outphasing Modulation” Proceedings of the Institute of Radio Engineers, vol. 23, No. 11, pp. 1370-1392 (Nov. 1935). |
| Choi, L.U., Multi-user MISO and MIMO Transmit Signal Processing for Wireless Communication, PhD Thesis submitted to the Hong Kong University of Science and Technology, 191 pages, Mar. 2003. |
| Clark, G. “A Comparison of AM Techniques,” ABU Technical Review, No. 44, p. 33-42, (May 1976). |
| Clark, G., “A Comparison of Current Broadcast Amplitude Modulation Techniques”, IEEE Transactions on Broadcasting, vol. BC-21, No. 2, pp. 25-31 (Jun. 1975). |
| Clifton, J.C. et al., “Novel Multimode J-pHEMT Front-End Architecture With Power-Control Scheme for Maximum Efficiency,” IEEE Transactions on Microwave Theory and Techniques, vol. 53, No. 6, pp. 2251-2258 (Jun. 2005). |
| Colantonio, P., “High Linearity and Efficiency Microwave PAs,” 12th GAAS Symposium—Amsterdam, pp. 183-186 (2004). |
| Computational Science Research Center Colloquium—Time Reversal Bases Communications in Complex Environments, Friday, Apr. 9, 2004, 2 pages, printed Jul. 14, 2006 from http://www.sdsunivers.info/info—content—event.asp?id=15044. |
| Conradi, C.P. et al., “Evaluation of a Lossless Combiner in a LINC Transmitter,” Proceedings of the 1999 IEEE Canadian Conference on Electrical Computer Engineering, pp. 105-110 (May 9-12, 1999). |
| Couch, L. and Walker, J.L., “A VHF LINC Amplifier,” Proceedings of IEEE Southeastcon, pp. 122-125 (1982). |
| Course #08: Advanced RF Power Amplifier Techniques for Modern Wireless and Microwave Systems, from http://www.cei.se/008.htm, 6 pages (printed Apr. 14, 2005). |
| Course #114: Advanced RF Power Amplifier Techniques, from http://www.bessercourse.com/outlinesOnly.asp?CTID=114, 3 pages (printed Jun. 22, 2005). |
| Cox, “Component Signal Separation and Recombination for Linear Amplification with Nonlinear Components,” IEEE Transactions on Communications, vol. COM-23, No. 11, pp. 1281-1287 (Nov. 1975). |
| Cox, D.C. and Leck, R.P., “A VHF Implementation of a LINC Amplifier,” IEEE Transactions on Communications, pp. 1018-1022 (Sep. 1976). |
| Cox, D.C., “Linear Amplification with Nonlinear Components,”IEEE Transactions on Communications, vol. COM-22, pp. 1942-1945 (Dec. 1974). |
| Cripps, S.C., Advanced Techniques in RF Power Amplifier Design, Section—“Doherty and Chireix,” pp. 33-72, Artech House (2002). |
| Cripps, Steve C., PA Linearisation in RFICs . . . ignoring the obvious?, available at http://www.cei.se/pa—milan.ppt, Hywave Associates, 24 pages (Created Aug. 2, 2001). |
| Cripps, Steve C., RF Power Amplifiers for Wireless Communications, Artech House, ISBN No. 0890069891, pp. 240-250 (Apr. 1999). |
| Deltimple, N. et al., “A Reconfigurable RF Power Amplifier Biasing Scheme”, Proceedings of the 2nd Annual IEEE Northeast Workshop on Circuits and Systems (NEWCAS2004), pp. 365-368, (Jun. 20-23, 2004). |
| Dennis, A., “A Novel Digital Transmitter Architecture for Multimode/Multiband Applications: DTX, A Technology of MACOM,” Tyco Electronics, 32 pages (Aug. 17, 2004). |
| Dinis, R. et al., “Performance Trade-Offs with Quasi-Linearly Amplified OFDM Through a Two-Branch Combining Technique,” IEEE 46th Vehicular Technology Conference, pp. 899-903 (Apr. 28-May 1, 1996). |
| Ellinger, F. et al., “Calibratable Adaptive Antenna Combiner at 5.2 GHz with High Yield for Laptop Interface Card,” IEEE Transactions on Microwave Theory and Techniques, vol. 48, No. 12, pp. 2714-2720 (Dec. 2000). |
| Faust, H.H. et al., “A Spectrally Clean Transmitting System for Solid-State Phased-Array Radars,” Proceedings of the 2004 IEEE Radar Conference, pp. 140-144 (Apr. 26-29, 2004). |
| Fisher, S.T., “A New Method of Amplifying with High Efficiency a Carrier Wave Modulated in Amplitude by a Voice Wave,” Proceedings of the Institute of Radio Engineers, vol. 34, pp. 3-13P (Jan. 1946). |
| Garcia, P. et al., “An Adaptive Digital Method of Imbalances Cancellation in LINC Transmitters,” IEEE Transactions on Vehicular Technology, vol. 54, No. 3, pp. 879-888 (May 2005). |
| Gaudernack, L.F., “A Phase-Opposition System of Amplitude Modulation,” IRE Proceedings, vol. 26, No. 8, pp. 983-1008 (Aug. 1938). |
| Genztler, C.G. and Leong, S.K., “Broadband VHF/UHF Amplifier Design Using Coaxial Transformers,” High Frequency Electronics, pp. 42, 44, 46, 48, 50, and 51 (May 2003). |
| Gerhard, W. and Knöchel, R., “Digital Component Separator for future W-CDMA-LINC Transmitters implemented on an FPGA,” Advances in Radio Science, 3, pp. 239-246 (2005). |
| Gründlingh, J. et al., “A High Efficiency Chireix Out-phasing Power Amplifier for 5GHz WLAN Applications,” IEEE MTT-S International Microwave Symposium Digest, vol. 3, pp. 1535-1538 (2004). |
| Hakala, I. et al., “A 2.14-GHz Chireix Outphasing Transmitter,” IEEE Transactions on Microwave Theory and Techniques, vol. 53, No. 6, pp. 2129-2138 (Jun. 2005). |
| Hakala, I. et al., “Chireix Power Combining with Saturated Class-B Power Amplifiers,” Conference Proceedings, 34th European Microwave Conference, pp. 379-382 (2004). |
| Hamedi-Hagh, S. and Salama, A.T., “CMOS Wireless Phase-Shifted Transmitter,” IEEE Journal of Solid-State Circuits, vol. 39, No. 8, pp. 1241-1252 (Aug. 2004). |
| Hammond, R. and Henry, J., “High Power Vector Summation Switching Power Amplifier Development,” IEEE Power Electronics Specialists Conference (PESC), pp. 267-272 (Jun. 29-Jul. 3, 1981). |
| Heiden, D., “Principle of a phase constant and low distortion amplitude modulation system for transistor transmitters,” Nachrichtentechnische Zeitschrift, vol. 23, No. 12, pp. 608-612 (Dec. 1970). |
| Hetzel, S.A. et al., “LINC Transmitter,” Electronics Letters, vol. 27, No. 10, pp. 844-846 (May 9, 1991). |
| Internet Postings at “Class E-AM Forum”:: View Topic—What exactly is class D?, at http://classe.monkeypuppet.com/viewtopic.php?t=220, 6 pages (Dec. 14-17, 2003). |
| Iwamoto, M. et al., “An Extended Doherty Amplifier with High Efficiency Over a Wide Power Range,” IEEE Transactions on Microwave Theory and Techniques, vol. 49, No. 12, pp. 2472-2479 (Dec. 2001). |
| Jeong, Y.-C., Linearizing Principles on High Power Amplifier, Chonbuk National University School of Electronics & Information Engineering, 41 pages (Oct. 26, 2004). |
| Karn, P., Re: [amsat-bb] AO-40 Satellite RF Architecture Question, at http://www.uk/amsat.org/ListArchives/amsat-bb/2002/msg01409.html, 2 pages (Feb. 25, 2002). |
| Katz, A., Linearization: Reducing Distortion in Power Amplifiers, The College of New Jersey, 52 pages (Apr. 16, 2004). |
| Kaunisto, R., “A Vector-Locked Loop for Power Amplifier Linearization,” IEEE MTT-S International Microwave Symposium Digest, 4 pages (Jun. 6-11, 2004). |
| Kelly, W.M. et al., “Vector Modulator, Output Amplifier, and Multiplier Chain Assemblies for a Vector Signal Generator,” Hewlett-Packard Journal, vol. 38, No. 11, pp. 48-52 (Dec. 1987). |
| Kenington, P.B. et al., “Broadband Linearisation of High-Efficiency Power Amplifiers,” Proceedings of the Third International Mobile Satellite Conference, pp. 59-64 (1993). |
| Kim, I. et al., “The linearity and efficiency enhancement using 3-way Doherty amplifier with uneven power drive,” International Technical Conference on Circuits/Systems, Computers and Communications, Jeju, Korea, pp. 369-370 (Jul. 2005). |
| Kim, J. et al., “Optimum Operation of Asymmetrical-Cells-Based Linear Doherty Power Amplifiers—Uneven Power Drive and Power Matching,”IEEE Transactions on Microwave Theory and Techniques, vol. 53, No. 5, pp. 1802-1809 (May 2005). |
| Kosugi, H. et al., “A High-Efficiency Linear Power Amplifier Using an Envelope Feedback Method,” Electronics and Communications in Japan, Part 2, vol. 77, No. 3, pp. 50-57 (1994). |
| Kurzok, R., “Simple Lab-Built Test Accessories for RF, IF, Baseband and Audio,” High Frequency Electronics, pp. 60 and 62-64 (May 2003). |
| Langridge, R. et al., “A Power Re-Use Technique for Improved Efficiency of Outphasing Microwave Power Amplifiers,” IEEE Transactions on Microwave Theory and Techniques, vol. 47, No. 8, pp. 1467-1470 (Aug. 1999). |
| Li. C. et al., “Optimal IDM-MISO Transmit Strategy with Partial CSI at Transmitter,” 6 pages, downloaded Jun. 2006 from http://www288.pair.com/ciss/ciss/numbered/36.pdf. |
| Love, D.J. et al., “Grassmannian Beamforming for Multiple-Input Multiple-Output Wireless Systems,” pp. 1-29, downloaded Jun. 2006 from http://www.math.uedavis.edu/˜strohmer/papers/2003/grassbeam.ps.gz, Jun. 3, 2003. |
| Lyles, J.T., [Amps] Amplifuzz [TSPA], at http://lists.contesting.com/pipermail/amps/2005-January/042303.html, 2 pages (Jan. 28, 2005). |
| Manual and Schematics, at http://www.lks.net/˜radio/Pages/manuals.htm, 8 pages (last update Aug. 23, 2005). |
| Masse, D., “Advanced Techniques in RF Power Amplifier Design,” Microwave Journal (International Edition), vol. 45, Issue 9, p. 216 (Sep. 2002). |
| Masse, D., “Design of Linear RF Outphasing Power Amplifiers,” Microwave Journal (International Edition), vol. 47, Issue 7, p. 152 (Jul. 2004). |
| McCune, E., “High-Efficiency, Multi-Mode Multi-Band Terminal Power Amplifiers,” IEEE Microwave Magazine, vol. 6, No. 1, pp. 44-55 (Mar. 2005). |
| McPherson, D.S. et al., “A 28 GHz HBT Vector Modulator and Its Application to an LMCS Feedforward Power Amplifier,” 28th European Microwave Conference—Amsterdam, vol. 1, pp. 523-528 (1998). |
| Mead Education: Information Registration: RF Transceivers and Power Amplifiers, at http://www.mead.ch/htm/ch/bios—texte/RF-PA—05—text.html, 3 pages (printed Sep. 1, 2005). |
| Morais, D.H. and Feher, K., “NLA-QAM: A Method for Generating High-Power QAM Signals Through Nonlinear Amplification,” IEEE Transections on Communications, vol. COM-30, No. 3, pp. 517-522 (Mar. 1982). |
| Moustakas, A.L. and Simon, S.H., “Optimizing multiple-input single-output (MISO) communication systems with general Gaussian channels; nontrivial covariance and nonzero mean,” IEEE Trans. on Information Theory, vol. 49, Issue 10, pp. 2770-2780, Oct. 2003. |
| Musson, D.R., “Ampliphase . . . for Economical Super-Power AM Transmitters”, Broadcast News, vol. No. 119, pp. 24-29 (Feb. 1964). |
| Norris, G.B. et al., “A Fully Monolithic 4-18 GHZ Digital Vector Modulator,” IEEE MIT-S International Microwave Symposium Diges, pp. 789-792 (1990). |
| Olson, S.A. and Stengel, R.E., “LINC Imbalance Correction using Baseband Preconditioning,” Proceedings IEEE Radio Wireless Conference, pp. 179-182 (Aug. 1-4, 1999). |
| Pereyra, L. A., “Modulation techniques for radiodiffusion transmitters,” Revista Telegrafica Electronica, vol. 67, No. 801, pp. 1132-1138 and 1148 (Oct. 1979). |
| Pigeon, M., “A CBC Engineering Report: Montreal Antenna Replacement Project,” Broadcast Technology, vol. 15, No. 4, pp. 25-27 (Jan. 1990). |
| Poitau, G. et al., “Experimental Characterization of LINC Outphasing Combiners' Efficiency and Linearity,” Proceedings IEEE Radio and Wireless Conference, pp. 87-90 (2004). |
| Price, T.H., “The Circuit Development of the Ampliphase Broadcasting Transmitter,” The Proceedings of The Institution of Electrical Engineers, vol. 101, pp. 391-399 (1954). |
| Qiu, R.C. et al., “Time Reversal with MISO for Ultra-Wideband Communications: Experimental Results (invited paper),” 4 pages, downloaded Jun. 2006 from http://iweb.tntech.edu/roiu/paper/conference/RWS06Qiu—TH2B1.pdf. |
| Raab, F.H. et al., “Power Amplifiers and Transmitters for RF and Microwave,” IEEE Transactions on Microwave Theory and Techniques, vol. 50, No. 3, pp. 814-826 (Mar. 2002). |
| Raab, F.H. et al., “RF and Microwave Power Amplifier and Transmitter Technologies—Part 1,” High Frequency Electronics, pp. 22, 24, 26, 28, 29, 30, 32, 34, and 36 (May 2003). |
| Raab, F.H. et al., “RF and Microwave Power Amplifier and Transmitter Technologies—Part 3,” High Frequency Electronics, pp. 34, 36, 38, 40, 42-44, 46, and 48 (2003). |
| Raab, F.H. et al., “RF and Microwave Power Amplifier and Transmitter Technologies—Part 5,” High Frequency Electronics, pp. 46, 48-50, 52, and 54 (2004). |
| Raab, F.H., “Efficiency of Doherty RF-Power Amplifier Systems,” IEEE Transactions on Broadcasting, vol. BC-33, No. 3, pp. 77-83 (Sep. 1987). |
| Raab, F.H., “Efficiency of Outphasing RF Power-Amplifier Systems,” IEEE Transactions on Communications, vol. COM-33, No. 10, pp. 1094-1099 (Oct. 1985). |
| Rabjohn, G. and Wight, J., “Improving Efficiency, Output Power with 802.11a Out-Phasing PAs,” at http://www.us.design-reuse.com/articles/article6937.html, 8 pages (Jan. 9, 2004). |
| Rustako, A.J. and Yeh, Y.S., “A Wide-Band Phase-Feedback Inverse-Sine Phase Modulator with Application Toward a LINC Amplifier,” IEEE Transactions on Communications, vol. COM-24, No. 10, pp. 1139-1143 (Oct. 1976). |
| Saleh, A.A.M. and Cox, D.C., “Improving the Power-Added Efficiency of FET Amplifiers Operating with Varying-Envelope Signals,” IEEE Transactions on Microwave Theory and Techniques, vol. 31, No. 1, pp. 51-56 (Jan. 1983). |
| Saraga, W., “A new version of the out-phasing (quadrature-modulation) method for frequency translation (SSB generation and detection),” Transmission Aspects of Communications Networks, pp. 131-134 (1964). |
| Shi, B. and Sundström, L., “A 200-MHz IF BiCMOS Signal Component Separator for Linear LINC Transmitters,” IEEE Journal of Solid-State Circuits, vol. 35, No. 7, pp. 987-993 (Jul. 2000). |
| Shi, B. and Sundström, L., “A Voltage-Translinear Based CMOS Signal Component Separator Chip for Linear LINC Transmitters,” Analog Integrated Circuits and Signal Processing, 30, pp. 31-39 (2002). |
| Shi, B. and Sundström, L., “Investigation of a Highly Efficient LINC Amplifier Topology,” Proceedings IEEE 45th Vehicular Technology Conference, vol. 2, pp. 1215-1219 (Oct. 7-11, 2001). |
| Shin, B. et al., “Linear Power Amplifier based on 3-Way Doherty Amplifier with Predistorter,” IEEE MTT-S International Microwave Symposium Digest, pp. 2027-2030 (2004). |
| Simon, M. and Weigel, R., “A Low Noise Vector Modulator with integrated Basebandfilter in 120 nm CMOS Technology,” 2003 IEEE Radio Frequency Integrated Circuits Symposium, pp. 409-412 (2003). |
| Skarbek, I. “New High-Efficiency 5-KW AM Transmitter ‘Unique Class C Amplifier Operates with 90% Efficiency’,” RCE Broadcast News # 107, pp. 8-13 (Mar. 1960). |
| Sokal, N. O., “RF Power Amplifiers, Classes A though S—How they Operate, and When to Use Each,” Electronics Industries Forum of New England, Professional Program Proceedings, Boston, MA, pp. 179-252 (1997). |
| Staudinger, J. et al, “High Efficiency CDMA RF Power Amplifier Using Dynamic Envelope Tracking Technique,” IEEE MTT-S International Microwave Symposium Digest, vol. 2, pp. 873-876 (Jun. 11-16, 2000). |
| Stengel, B. and Eisenstadt, W.R., “LINC Power Amplifier Combiner Method Efficiency Optimzation,” IEEE Transactions on Vehicular Technology, vol. 49, No. 1, pp. 229-234 (Jan. 2000). |
| Sundström, L. “Spectral Sensitivity of LINC Transmitters to Quadrature Modulator Misalignments,” IEEE Transactions on Vehicular Technology, vol. 49, No. 4, pp. 1474-1487 (Jul. 2000). |
| Sundström, L., “Automatic adjustment of gain and phase imbalances in LINC transmitters,” Electronics Letters, vol. 31, No. 3, pp. 155-156 (Feb. 2, 1995). |
| Sundström, L., “Effect of modulation scheme on LINC transmitter power efficiency,” Electronics Letters, vol. 30, No. 20, pp. 1643-1645 (Sep. 29, 1994). |
| Sundström, L., “Effects of reconstruction filters and sampling rate for a digital signal component separator on LINC transmitter performance,” Electronics Letters, vol. 31, No. 14, pp. 1124-1125 (Jul. 6, 1995). |
| Sundström, L., “The Effect of Quantization in a Digital Signal Component Separator for LINC Transmitters,” IEEE Transactions on Vehicular Technology, vol. 45, No. 2, pp. 346-352 (May 1996). |
| Sundström, L., Digital RF Power Amplifier Linearisers Analysis and Design, Department of Applied Electronics, Lund University, pp. i-x and 1-64 (1995). |
| Tan, J. S. and Gardner, P., “A LINC Demonstrator Based on Switchable Phase Shifters,” Microwave and Optical Technology Letters, vol. 35, No. 4, pp. 262-264 (Nov. 20, 2002). |
| Tchamov, N. T., Power Amplifiers, Tampere University of Technology, Institute of Communications Engineering, RF-ASIC Laboratory, 26 pages (May 17, 2004). |
| TDP: RCA BHF-100A, at http://www.transmitter.be/rca-bhf100a.html, 8 pages (printed Jun. 15, 2005). |
| The Ampliphase Ancestry, at http://www.rossrevenge.co.uk/tx/ancest.htm, 8 pages, (latest update Aug. 2002). |
| Tomisato, S. et al., “Phase Error Free LINC Modulator,” Electronics Letters, vol. 25, No. 9, pp. 576-577 (Apr. 27, 1989). |
| Ullah, I., “Exciter Modulator For An Ampliphase Type Broadcast Transmitter,” ABU Technical Review, No. 62, pp. 21-27 (May 1979). |
| Ullah, I., “Output Circuit Of An Ampliphase Broadcast Transmitter,” ABU Technical Review, No. 63, pp. 17-24 (Jul. 1979). |
| Vasyukov, V.V. et al., “The Effect of Channel Phase Asymmetry on Nonlinear Distortions in Modulation by Dephasing,” Radioelectronics and Communications Systems, vol. 28, No. 4, pp. 86-87 (1985). |
| Venkataramani, M., Efficiency Improvement of WCDMA Base Station Transmitters using Class-F power amplifiers, Thesis, Virginia Polytechnic Institute, Blacksburg, Virginia, pp. i-xi and 1-55 (Feb. 13, 2004). |
| Virmani, B.D., “Phase-to-amplitude modulation,” Wireless World, vol. 61, No. 4, pp. 183-187 (Apr. 1955). |
| Wang, F. et al., “Envelope Tracking Power Amplifier with Pre-Distortion Linearization for WLAN 802.11g,” 2004 IEEE MTT-S International Microwave Symposium Digest, vol. 3, pp. 1543-1546 (Jun. 6-11, 2004). |
| Whitaker, Jerry C., Power Vacuum Tubes Handbook (Electronics Handbook Series), CRC Publishing, ISBN No. 0849313457, pp. 236-238 (May 1999). |
| Wight, J., “Computational Microwave circuits arrive,” at http://www.eetimes.com/showArticle.ihtml?articleID=18900752, EE Times, 3 pages (Apr. 12, 2004). |
| Wilds, R.B., “An S-Band Two-Phase Demodulator,” pp. 48-53 (Aug. 1958). |
| Woo, Y.Y. et al., “SDR Transmitter Based on LINC Amplifier with Bias Control,” IEEE MTT-S International Microwave Symposium Digest, pp. 1703-1706 (2003). |
| Ya, S. et al., “A C-Band Monolithic Vector Modulator,” Research & Progress of SSE, vol. 14, No. 4, pp. 302-306 (Nov. 1994). |
| Yang, Y. et al., “A Fully Matched N-Way Doherty Amplifier With Optimized Linearity,” IEEE Transactions on Microwave Theory and Techniques, vol. 51, No. 3. pp. 986-993 (Mar. 2003). |
| Yang, Y. et al., “A Microwave Doherty Amplifier Employing Envelope Tracking Technique for High Efficiency and Linearity,” IEEE Microwave and Wireless Components Letters, vol. 13, No. 9, pp. 370-372 (Sep. 2003). |
| Yang, Y. et al., “Experimental Investigation on Efficiency and Linearity of Microwave Doherty Amplifier,” IEEE, 4 pages (2001). |
| Yang, Y. et al., “Optimum Design for Linearity and Efficiency of a Microwave Doherty Amplifier Using a New Load Matching Technique,” Microwave Journal, 8 pages (Dec. 1, 2001). |
| Yankin, V. A., “Effect of quantization, amplifier noise and the parameters of the calibration elements on the accuracy of measurement using a six-port microwave ampliphasemeter,” Radioelectronics and Communication Systems, vol. 32, No. 8, pp. 110-112 (1989). |
| Yao, J. and Long, S.I., “High Efficiency Switching-Mode Amplifier for Mobile and Base Station Applications,” Final Report Mar. 2002 for MICRO Project 02-044, 4 pages (2002-2003). |
| Yao, J. et al., “Efficiency Switch Mode Amplifiers for Mobile and Base Station Applications,” Final Report 2000-2001 for MICRO Project 00-061, 4 pages (2000-2001). |
| Yi, J. et al., “Effect of Effciency optimization on linearity of LINC amplifiers with CDMA signal,” IEEE MTT-S International Microwave Symposium Digest, vol. 2, pp. 1359-1362 (May 2001). |
| Zhang, X., An Improved Outphasing Power Amplifier System for Wireless Communications, Dissertation, University of California, San Diego, pp. i-xvii and 1-201 (2001). |
| Zhang, X. and Larson, L.E., “Gain and Phase Error-Free LINC Transmitters,” IEEE Transactions on Vehicular Technology, vol. 49, No. 5, pp. 1986-1994 (Sep. 2000). |
| Zhang, X. et al. “Gain/Phase Imbalance-Minimization Techniques for LINC Transmitters,” IEEE Transactions on Microwave Theory and Techniques, vol. 49, No. 12, pp. 2507-2516 (Dec. 2001). |
| Zhang, X. et al., “A Gain/Phase Imbalance Minimization Technique for LINC Transmitter,” IEEE MTT-S International Microwave Symposium Digest, pp. 801-804 (2001). |
| Zhang, X. et al., “Analysis of Power Recycling Techniques for RF and Microwave Outphasing Power Amplifiers,” IEEE Transactions on Circuits and Systems II: Analog and Digital Signal Processing, vol. 49, No. 5, p. 312-320 (May 2002). |
| Zhang, X. et al., “Calibration scheme for LINC transmitter,” Electronics Letters, vol. 37, No. 5, pp. 317-318 (Mar. 1, 2001). |
| Zhang, X. et al., Design of Linear RF Outphasing Power Amplifiers, entire book, Artech House, ISBN No. 1-58053-374-4 (2003). |
| Zhong, S.S. and Cui, J.H., “A New Dual Polarized Aperture-Coupled Printer Array for SAR Applications,” Journal of Shanghai University (English Edition), vol. 5, No. 4, pp. 295-298. |
| English Abstract for European Patent Publication No. EP 0 639 307 B1, published Feb. 22, 1995, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for European Patent Publication No. EP 0 708 546 A2, published Apr. 24, 1996, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for European Patent Publication No. EP 0 892 529 A2, published Jan. 20, 1999, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 60-63517 A, published Apr. 11, 1985, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2-87708 A, published Feb. 28, 1990, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 3-232307 A, published Oct. 16, 1991, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 5-22046 A, published Jan. 29, 1993, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 6-338728 A, published Dec. 6, 1994, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 10-70451 A, published Mar. 19, 1998, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2001-136057 A, published May 18, 2001, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2004-260707 A, published Sep. 16, 2004, downloaded from http://v3.espacenet.com, 1 page. |
| English Translation for Romanian Patent Publication No. RO 100466, published Aug. 20, 1992, obatined from Transperfect Translations, 4 pages. |
| English Abstract for Romanian Patent Publication No. RO 102824, published Nov. 19, 2001, downloaded from http://v3.espacenet.com, 1 page. |
| English Translation for Russian Patent Publication No. SU 1322183 A1, published Jul. 7, 1987, obtained from Transperfect Translations, 2 pages. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Mar. 4, 2008, for PCT Application No. PCT/US07/06197, 8 pages. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Aug. 15, 2008, for PCT Application No. PCT/US08/06360, 6 pages. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Sep. 3, 2008, for PCT Application No. PCT/US2008/008118, 6 pages. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Sep. 8, 2008, for PCT Application No. PCT/US2008/007623, 6 pages. |
| Silverman, L. and Del Plato, C., “Vector Modulator Enhances Feedforward Cancellation,” Microwaves & RF, pp. 1-4 (Mar. 1998). |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Jul. 7, 2009, for PCT Application No. PCT/US09/03212, 6 pages. |
| Jang, M. et al., “Linearity Improvement of Power Amplifier Using Modulation of Low Frequency IMD Signals,” Asia-Pacific Microwave Conference Proceedings, vol. 2, pp. 1156-1159, Dec. 4-7, 2005. |
| Woo, W. et al., “A Hybrid Digital/RF Envelope Predistortion Linearization System for Power Amplifiers,” IEEE Transactions on Microwave Theory and Techniques, vol. 53, No. 1, pp. 229-237, Jan. 2005. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Apr. 27, 2010, for PCT Application No. PCT/US2009/057306, 11 pages. |
| English Abstract for Japanese Patent Publication No. JP 2005-151543 A, published Jun. 9, 2005, downloaded from http://v3.espaceret.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 1-284106 A, published Nov. 15, 1989, downloaded from http://v3.cspacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 4-095409 A, published Mar. 27, 1992, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 4-104604 A, published Apr. 7, 1992, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 9-018536 A, published Jan. 17, 1997, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 9-074320 A, published Mar. 18, 1997, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2000-209291 A, published Jul. 28, 2000, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2003-298357 A, published Oct. 17, 2003, downloaded from http://v3.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2000-244261 A, published Sep. 8, 2000, downloaded from http://worldwide.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2001-217659 A, published Aug. 10, 2001, downloaded from http://worldwide.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2001-308650 A, published Nov. 2, 2001, downloaded from http://worldwide.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2002-543729 A, published Dec. 17, 2002, downloaded from http://worldwide.espacenet.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 5-037263 A, published Feb. 12, 1993, downloaded from http://worldwide.espace.com, 1 page. |
| English Abstract for Japanese Patent Publication No. JP 2005-101940 A, published Apr. 14, 2005, downloaded from http://worldwide.espacenet.com, 1 page. |
| Office Communication, dated May 29, 2008, for U.S. Appl. No. 11/512,359, filed Aug. 30, 2006, 19 pages. |
| Office Communication, dated May 7, 2009, for U.S. Patent Appl. No. 11/599,390, filed Nov. 15, 2006, 6 pages. |
| Office Communication, dated Nov. 20, 2010, for U.S. Appl. No. 11/599,390, filed Nov. 15, 2006, 14 pages. |
| Office Communication, dated May 25, 2010, for U.S. Appl. No. 11/599,390, filed Nov. 15, 2006, 17 pages. |
| Office Communication, dated Oct. 22, 2010, for U.S. Appl. No. 11/599,390, filed Nov. 15, 2006, 17 pages. |
| Office Communication, dated Mar. 7, 2011, for U.S. Appl. No. 11/599,390, filed Nov. 15, 2006, 18 pages. |
| Office Communication, dated Jul. 18, 2011, for U.S. Appl. No. 11/599,390, filed Nov. 15, 2006, 17 pages. |
| Office Communication, dated Oct. 31, 2007, for U.S. Appl. No. 11/599,344, filed Nov. 15, 2006, 11 pages. |
| Office Communication, dated Aug. 12, 2008, for U.S. Appl. No. 11/599,344, filed Nov. 15, 2006, 8 pages. |
| Office Communication, dated Feb. 24, 2009, for U.S. Appl. No. 11/599,344, filed Nov. 15, 2006, 7 pages. |
| Office Communication, dated Nov. 23, 2009, for U.S. Appl. No. 11/599,344, filed Nov. 15, 2006, 9 pages. |
| Office Communication, dated Jul. 7, 2010, for U.S. Appl. No. 11/599,344, filed Nov. 15, 2006, 11 pages. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Aug. 14, 2012, for PCT Appl. No. PCT/US2012/032791, 7 pages. |
| Harlan, G. et al, “Dynamically-Configurable Multimode Transmitter Systems for Wireless Handsets, Cognitive Radio and SDR Applications,” IEEE International Conference on Microwaves, Communications, Antennas and Electronics Systems, Nov. 9, 2009, pp. 1-5. |
| Rawlins, G. and Sorrells, D., “A Thermodynamic Theory of RF Power Transmitters with an Example,” IEEE 10th Annual Wireless and Microwave Technology Conference, Apr. 20, 2009, pp. 1-5. |
| Rawlins, G. et al., “Using an IQ Data to RF Power Transmitter to Realize a Highly-Efficient Transmit Chain for Current and Next-Generation Mobile Handsets,” Proceedings of the 38th European Microwave Conference, Oct. 27, 2008, pp. 579-582. |
| Notification of Transmittal of the International Search Report and Written Opinion, dated Aug. 14, 2012, for PCT Appl. No. PCT/US2012/040500, 9 pages. |
| Complaint, filed Dec. 28, 2011, in the United States District Court, District of New Jersey, Maxtak Capital Advisors LLC et al. v. ParkerVision, Inc. et al., Case No. 2:11-cv-07549-CCC-JAD, 63 pages. |
| English Abstract for Japanese Patent Publication No. JP H08-163189 A, published Jun. 21, 1996, downloaded from http://worldwide.espacenet.com, 2 pages. |
| English Abstract for Japanese Patent Publication No. JP 2003-298361 A, published Oct. 17, 2003, downloaded from http://worldwide.espacenet.com, 2 pages. |
| Number | Date | Country | |
|---|---|---|---|
| 20130129019 A1 | May 2013 | US |
| Number | Date | Country | |
|---|---|---|---|
| 60620972 | Oct 2004 | US | |
| 60671542 | Apr 2005 | US | |
| 60671536 | Apr 2005 | US | |
| 60673397 | Apr 2005 | US | |
| 60706003 | Aug 2005 | US | |
| 60709092 | Aug 2005 | US | |
| 60717244 | Sep 2005 | US | |
| 60721114 | Sep 2005 | US |
| Number | Date | Country | |
|---|---|---|---|
| Parent | 11512146 | Aug 2006 | US |
| Child | 13747998 | US | |
| Parent | 11256172 | Oct 2005 | US |
| Child | 11512146 | US |
