SEMICONDUCTOR INTEGRATED CIRCUIT FOR RF COMMUNICATIONS

Abstract

An AFC-control D/A converter which controls a reference frequency oscillator is a voltage-potentiometer-type D/A converter containing three voltage followers. At least in the latter-stage voltage follower, an NMOS differential input circuit, a CMOS output circuit, and a bias circuit are supplied with an external power voltage. However, PMOS differential input circuit is supplied with an internal regulated power supply voltage generated by a reference voltage generator. Even if there is a shift in the pair nature of MP1 and MP2 of the differential PMOS, an increase of current of MP3 of a PMOS current source due to the increase of the external power voltage is suppressed. Also an input offset voltage of the differential PMOS does not increase, and a change of an AFC control analog output signal can be reduced.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating an entire constitution of a mobile terminal device which mounts an RF IC and a baseband LSI with a digital interface examined by the present inventors in advance of the present invention, and also a block diagram illustrating an entire constitution of a mobile terminal device according to one embodiment of the present invention;

FIG. 2 is a circuit diagram illustrating constitution of an AFC-control D/A converter arranged in the RF analog signal processing integrated circuit of the mobile terminal device illustrated in FIG. 1;

FIG. 3 is a circuit diagram illustrating constitution of a CMOS rail-to-rail amplifier which constitutes three voltage followers of the AFC-control D/A converter illustrated in FIG. 2;

FIG. 4 is a drawing of simulation result illustrating change of an AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 2 due to the change of an external power voltage when 100% of pair nature of the electrical property of differential pair MOS is attained;

FIG. 5 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 2 due to the change of the external power voltage when the pair nature of the electrical property of the differential pair NMOS of the NMOS differential input circuit of FIG. 3 shifts 10%;

FIG. 6 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 2 due to the change of the external power voltage when the pair nature of the electrical property of the differential pair PMOS of the PMOS differential input circuit of FIG. 3 shifts 10%;

FIG. 7 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 2 due to the change of the external power voltage when the pair nature of the electrical property of the differential pair NMOS of an NMOS differential input circuit of FIG. 3 shifts 10% and the pair nature of the electrical property of differential pair PMOS of the PMOS differential input circuit of FIG. 3 shifts 10%;

FIG. 8 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 2 due to the change of the external power voltage when the pair nature of the electrical property of the differential pair NMOS of the NMOS differential input circuit of FIG. 3 shifts 10% and the pair nature of the electrical property of the differential pair PMOS of the PMOS differential input circuit of FIG. 3 shifts 10%;

FIG. 9 is a drawing of an experimental result illustrating change of an AFC control analog output signal measured on an semiconductor chip containing an AFC-control D/A converter, the semiconductor chip having been actually made for trial;

FIG. 10 is a circuit diagram illustrating constitution of a CMOS rail-to-rail amplifier which constitutes three voltage followers of the AFC-control D/A converter of the mobile terminal device of FIG. 1 as one embodiment of the present invention;

FIG. 11 is a circuit diagram illustrating constitution of an AFC-control D/A converter arranged in the RF analog signal processing integrated circuit of the mobile terminal device illustrated in FIG. 1 as one embodiment of the present invention;

FIG. 12 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 11 when the source of the current source transistor of the PMOS differential input circuit of the CMOS rail-to-rail amplifier illustrated in FIG. 10 is connected to an external power voltage as conventional instead of the internal regulated power supply voltage;

FIG. 13 is a drawing of simulation result illustrating change of the AFC control analog output signal of the AFC-control D/A converter illustrated in FIG. 11 when the source of the current source transistor of the PMOS differential input circuit of the CMOS rail-to-rail amplifier illustrated in FIG. 10 is connected to the internal regulated power supply voltage according to the embodiment of the present invention, instead of an external power voltage;

FIG. 14 is a drawing illustrating constitution of a fractional synthesizer of a fractional-N PLL arranged in the RF analog signal processing integrated circuit of the mobile terminal device illustrated in FIG. 1 as one embodiment of the present invention;

FIG. 15 is a drawing illustrating constitution of the RF analog signal processing integrated circuit RF IC according to a more concrete embodiment of the present invention;

FIG. 16 is a drawing illustrating constitution of the RF analog signal processing integrated circuit RF IC according to a still more concrete embodiment of the present invention;

FIG. 17 is a drawing illustrating constitution of the RF analog signal processing integrated circuit RF IC according to another concrete embodiment of the present invention;

FIG. 18 is a drawing illustrating frequency spectra of an RF transmission signal of a mobile phone terminal apparatus specified by GMSK specification; and

FIG. 19 is a circuit diagram illustrating another constitution of a CMOS rail-to-rail amplifier which constitutes three voltage followers of the AFC-control D/A converter of the mobile terminal device of FIG. 1 as one embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

<<Entire Constitution of the Mobile Terminal Device>>

FIG. 1 is a drawing illustrating an entire constitution of a mobile terminal device according to one embodiment of the present invention. Although the mobile terminal device in the present description assumes a mobile phone terminal device, alternatively, it may be a mobile communication device for a notebook-size personal computer or for a PDA (Personal Digital Assist) device. In the mobile terminal device illustrated in FIG. 1, A/D converters 303, 304 and D/A converters 307, 308, 315 are arranged in the RF analog signal processing integrated circuit (RF_IC) 300. Namely, the A/D converters 303 and 304 converts analog baseband signals RxABI and RxABQ into digital baseband signals RxDBI and RxDBQ and supplies the digital baseband signals RxDBI and RxDBQ to a baseband signal processing LSI (BB_LSI) 400, wherein the analog baseband signals RxABI and RxABQ are an output of an RF-reception-signal analog signal processing subunit (RX SPU) 301 in the RF analog signal processing integrated circuit (RF_IC) 300. The D/A converters 307 and 308 convert quadrature components TxDBI and TxDBQ of a digital baseband transmitting signal of an output of the baseband signal processing LSI (BB_LSI) 400 into analog baseband transmitting signals TxABI and TxABQ. The D/A converters 307 and 308 supply an RF-transmission-signal analog signal processing subunit (TX SPU) 302 in the RF analog signal processing integrated circuit (RF_IC) 300 with the analog baseband transmitting signals TxABI and TxABQ. Furthermore, the AFC-control D/A converter (AFCDAC) 315 converts an AFC control digital signal into an AFC control analog signal, and supplies it to a system reference clock oscillator (VCXO) 314. Here, the AFC control digital signal is an output of a baseband processor core 401, obtained by a digital signal path L3 of an RF digital interface 402 of the baseband signal processing LSI 400.

In order that the mobile phone terminal device may perform receiving operation from a base station and a send operation to the base station, an antenna (ANT) 100 receives a reception signal of radio frequency (called RF hereinafter) from the base station, and outputs an RF transmission signal to the base station. The antenna 100 is connected to a front-end module (FEM) 200. The front-end module 200 has an antenna switch (ANT_SW) 201. When the antenna switch 201 is connected to the upper side, the RF reception signal received with the antenna 100 is supplied to a receive filter (SAW) 202 made of a surface acoustic wave device (passing a desired frequency signal, and decreasing a disturbing frequency signal). On the other hand, when the antenna switch 201 is connected to the down side, the antenna switch 201 is connected to an output of a transmission RF power amplifier (RF_PA) 203. Therefore, an RF transmission signal is outputted from the antenna 100 to the base station by RF power output of the transmission RF power amplifier 203. The antenna switch 201 of the front-end module 200 is connected to the upper side in a time slot of receiving operation of the TDMA (Time Sharing Multiple Access) system, and it is connected to the down side in a time slot of send operation.

An RF reception signal of the output of the receive filter 202 of the front-end module 200 is supplied to an input of the RF-reception-signal analog signal processing subunit (RX SPU) 301 arranged in the RF analog signal processing integrated circuit (RF_IC) 300 as an RF analog signal processing unit. On the other hand, an RF input of the transmission RF power amplifier 203 of the front-end module 200 is connected to an output of the RF-transmission-signal analog signal processing subunit (TX SPU) 302 in the RF analog signal processing integrated circuit 300.

<<Entire Constitution of the RF Analog Signal Processing Integrated Circuit>>

Next, the RF analog signal processing integrated circuit 300, which performs signal transfer bidirectionally with the baseband digital LSI 400 via the digital interface, is explained in detail.

First, the RF analog signal processing integrated circuit 300 includes the RF-reception-signal analog signal processing subunit 301 and the RF-transmission-signal analog signal processing subunit 302. The RF-reception-signal analog signal processing subunit 301 performs frequency down-conversion of the RF reception signal into the analog baseband reception signals RxABI and RxABQ, and the RF-transmission-signal analog signal processing subunit 302 performs frequency up-conversion of the analog baseband transmitting signals TxABI and TxABQ into the RF transmission signal. The RF analog signal processing integrated circuit 300 includes the reference frequency oscillator 314 which generates the reference frequency signal for generating a high frequency signal, which is used for the frequency down-conversion in the RF-reception-signal analog signal processing subunit 301, and the frequency up-conversion in the RF-transmission-signal analog signal processing subunit 302. The RF analog signal processing integrated circuit 300 includes further the AFC-control D/A converter 315 operable to control frequency of the reference frequency signal, which is generated by the reference frequency oscillator 314, by converting an AFC control digital input signal supplied from the baseband LSI 400 into an AFC control analog output signal.

The RF-reception-signal analog signal processing subunit 301 in the RF analog signal processing integrated circuit 300 creates quadrature components RxABI and RxABQ of the analog baseband reception signal from the RF reception signal supplied by the receive filter 202. These quadrature components RxABI and RxABQ are supplied to an input of the A/D converter (I_ADC) 303 for analog-baseband reception signal I, and an input of the A/D converter (Q_ADC) 304 for analog-baseband reception signal Q. The A/D converters 303 and 304 for the analog baseband reception signals I and Q convert the supplied analog baseband reception signals RxABI and RxABQ into digital baseband reception signals RxDBI and RxDBQ. These digital baseband reception signals RxDBI and RxDBQ are supplied to two inputs of a multiplexer (MPX) 305. The multiplexer 305 is connected to the baseband signal processing LSI (BB_LSI) 400, which is a baseband digital signal processing unit, via a bidirectional digital signal path L5. The bidirectional digital signal path L5 possesses one signal line (1 bit), therefore, in receiving operation, two digital baseband reception signals RxDBI and RxDBQ are supplied to the baseband signal processing LSI 400 by time sharing.

In send operation, the multiplexer 305 outputs a digital baseband transmitting signal TxDB fed by the baseband signal processing LSI 400 via the bidirectional digital signal path L5, which possesses one signal line (1 bit), to a digital baseband modulator (Dig_MOD) 306. The digital baseband modulator 306 creates quadrature components TxDBI and TxDBQ of the digital baseband transmitting signals from the digital baseband transmitting signal TxDB supplied by the multiplexer 305. These quadrature components TxDBI and TxDBQ are supplied to an input of the D/A converter (I_DAC) 307 for the digital baseband transmitting signal I, and to an input of the D/A converter (Q_DAC) 308 for the digital baseband transmitting signal Q, respectively. The D/A converters 307 and 308 for the digital baseband transmitting signals I and Q convert the supplied digital baseband transmitting signals TxDBI and TxDBQ into analog baseband transmitting signals TxABI and TxABQ. These analog baseband transmitting signals TxABI and TxABQ are supplied to the input of the RF-transmission-signal analog signal processing subunit (TX SPU) 302 in the RF analog signal processing integrated circuit 300. The RF-transmission-signal analog signal processing subunit 302 creates an RF transmission signal from the analog baseband transmitting signal TxABI and TxABQ, and supplies the RF transmission signal to the transmission RF power amplifier 203 as an RF power input. The transmission RF power amplifier 203 amplifies the RF power input to create an RF power output from the RF-amplified output. An amplification gain of the transmission RF power amplifier 203 is set up by an automatic power control voltage Vapc created by a ramp-signal D/A converter (Ramp DAC) 309 in the RF analog signal processing integrated circuit 300. Not only the operating condition of the ramp-signal D/A converter 309 but the operating conditions of the RF-reception-signal analog signal processing subunit 301 and of the RF-transmission-signal analog signal processing subunit 302 are controlled by a transmitting/receiving control subunit (Rx/Tx_CTRL) 310 in the RF analog signal processing integrated circuit 300. The transmitting/receiving control subunit 310 is connected with the baseband signal processing LSI 400 via a first interface (INT_1) 311, a second interface (INT_2) 312 and digital signal paths L1, L2, L3, and L4.

<<A Digital-Interface of the RF Analog Signal Processing Integrated Circuit>>

The digital signal of the digital signal path L1 is control data (Ctrl Data) supplied from the baseband signal processing LSI 400, and the control data includes an instruction code for setting-operation, and control information for executing the instruction. The digital signal of the digital signal path L2 is a control clock (Ctrl CLk) supplied from the baseband signal processing LSI 400, and the control clock is a synchronous control signal for the setting-operation. The digital signal of the digital signal path L3 is a control enable signal (Ctrl En) supplied from the baseband signal processing LSI 400. The control enable signal (Ctrl En) is driven to a level which makes enabling possible by the baseband signal processing LSI 400, when the baseband signal processing LSI 400 sets up an operating condition of the transmission/reception operation of an internal circuitry in the RF analog signal processing integrated circuit 300 and the transmission/reception operation of the front-end module 200. On the other hand, the digital signal of the digital signal path L4 is a strobe signal (Strb) used by a special operation mode of operation setting which makes plural time slots one setting unit. In this special operation mode, before outputting the strobe signal (Strb) to the digital signal path L4, reservation of operation setting which makes plural time slots one setting unit is performed. After the completion of reservation of operation setting in this special operation mode, a strobe signal (Strb) is supplied to the second interface (INT_2) 312 of the RF analog signal processing integrated circuit 300. This strobe signal (Strb) determines to which timing of time slots the instruction code for the reserved operation setting and the control information for executing the instruction are supplied to the RF analog signal processing subunits 301 and 302 and to the front-end module 200 from the transmitting/receiving control subunit 310.

<<A System Reference Clock Oscillator of the RF Analog Signal Processing Integrated Circuit>>

The RF analog signal processing integrated circuit 300 has a system reference clock oscillator (VCXO) 314. The oscillating frequency of a system reference clock signal SysCLk created in terms of an output of the system reference clock oscillator 314 can be maintained stable by a crystal oscillator (Xtal) 501 outside the integrated-circuit 300 and an AFC control analog signal for automatic frequency control supplied by the AFC-control D/A converter (AFCDAC) 315. The AFC control digital signal supplied to the AFC-control D/A converter (AFCDAC) 315 is low speed data at several 10 to several 100 kHz, which is a kind of control data (Ctrl Data) supplied to the first interface (INT_1) 311 from the baseband processor core 401 of the baseband signal processing LSI 400 via the digital signal path L1. The baseband processor core 401 of the baseband signal processing LSI 400 generates, by means of digital signal processing of the digital baseband signal, the AFC control digital signal which serves as to compensate the deviation from the desired value of 26 MHz of the oscillating frequency of the system reference clock signal SysCLk of the system reference clock oscillator (VCXO) 314. The AFC control digital signal is converted into the AFC control analog signal by the D/A converter (AFCDAC) 315, and then the capacity of a variable capacitance element of the system reference clock oscillator (VCXO) 314 is controlled by the AFC control analog signal. Consequently, the oscillating frequency of the system reference clock signal SysCLk of the system reference clock oscillator (VCXO) 314 comes to be in agreement with the desired value of 26 MHz.

<<Transmission and Reception Operations of the Mobile Terminal Device>>

Next, transmission and reception operations of the mobile terminal device are explained. The baseband signal processing LSI 400 establishes communication of the GSM system or the EDGE system using the RF analog signal processing integrated circuit 300 and the front-end module 200. In that case, a GSM timer (GSM Timer) 403 in the baseband signal processing LSI 400 supplies a system-reference-clock-signal enabling signal SysCLkEn to the RF analog signal processing integrated circuit 300. Then, the system reference clock signal SysCLk created in terms of the output of the system reference clock oscillator 314 of the RF analog signal processing integrated circuit 300 is supplied to the GSM timer (GSM Timer) 403 in the baseband signal processing LSI 400, via a wave-shaping circuit 3103 of the transmitting/receiving control subunit 310. This information is supplied also to the baseband processor core (BB_Pr_Core) 401 in the baseband signal processing LSI 400. Then, a CPU in the baseband processor core 401 starts operation setting of a time slot in the time sharing multiple access system, via the RF digital interface (Dig_RF_INT) 402 and the digital signal paths L1, L2, L3, L4. A digital signal processor (DSP) in the baseband processor core 401 performs signal processing about a receiving baseband signal processed by the RF-reception-signal analog signal processing subunit 301 of the RF analog signal processing integrated circuit 300. By this signal processing, when communication established in advance is the GSM system, phase demodulation is performed by generating a phase modulation component. By the result of the phase demodulation, an audio signal of conversation of a communications partner is obtained with a D/A converter (DAC) 502 and a speaker (SP) 503 outside the baseband signal processing LSI 400. On the other hand, an analog audio signal uttered by a user who uses the mobile terminal device of FIG. 1 is converted into a digital audio signal by a microphone (MIC) 504 and an A/D converter (ADC) 505. The digital signal processor (DSP) in the baseband processor core 401 performs signal processing about this digital audio signal. By this signal processing, when communication established in advance is the GSM system, phase modulation is performed. Consequently, it becomes possible to include the phase modulation component in the transmitting baseband signal which should be processed by the RF-transmission-signal analog signal processing subunit 302 of the RF analog signal processing integrated circuit 300. When communication established in advance is the EDGE system, since not only the phase modulation component but the amplitude modulation component is included in the communicative transmitting/receiving information, a communicative data transfer rate can be improved. The baseband signal processing LSI 400 has an SRAM 404 as an internal memory, which can be used as a work memory in the case of communication of the GSM system or the EDGE system.

The baseband signal processing LSI 400 is connectable with an external nonvolatile memory (not shown) and an application processor (not shown). The application processor is connected to a liquid crystal display device (not shown) and a key input device (not shown), and is able to execute various application programs containing a general program and a game. A boot program (a start-up initializing program) of mobile devices including a mobile phone, an operating system program (OS), a program for phase demodulation about a receiving baseband signal and phase modulation about a transmitting baseband signal of the GSM system by the digital signal processor (DSP) in the baseband signal processing LSI 400, and various application programs can be stored in an external nonvolatile memory.

<<CMOS Rail-to-Rail-Amplifier>>

FIG. 10 is a circuit diagram illustrating constitution of a CMOS rail-to-rail amplifier which constitutes three voltage followers (AMP1, AMP2, AMP3) of the AFC-control D/A converter (AFCDAC) 315 of the mobile terminal device of FIG. 1 as one embodiment of the present invention. The point that the CMOS rail-to-rail amplifier of FIG. 10 is different from the CMOS rail-to-rail amplifier illustrated in FIG. 3 is, for the first place, that a reference voltage generator RVG, which generates internal regulated power supply voltage Vdd_int maintained stable to about 2.45 V from the external power voltage Vdd_ext of a standard value of 2.8 V with the range of fluctuation from 2.67 V (minimum) to 3.0 V (maximum), is included in the CMOS rail-to-rail amplifier of FIG. 10. By supplying the external power voltage Vdd_ext to a band-gap reference circuit BGR of the reference voltage generator RVG, a band-gap reference voltage Vref of about 1.23 V is created from the band-gap reference circuit BGR. From this reference voltage Vref, the internal regulated power supply voltage V_REF and the internal regulated power supply voltage Vdd_int which are maintained stable to about 2.45 V are created.

In the CMOS rail-to-rail amplifier which constitutes the first voltage follower AMP1, the second voltage follower AMP2, and the third voltage follower AMP3 of the AFC-control D/A converter 315 of the mobile terminal device illustrated in FIG. 1 as one embodiment of the present invention, the external power voltage Vdd_ext is supplied to an NMOS differential input circuit NMOS_DA, a bias circuit BIAS_CKT, and a CMOS output circuit OUT_CKT, as illustrated in FIG. 10. In the CMOS rail-to-rail amplifier which constitutes at least the third voltage follower AMP3 of the AFC-control D/A converter (AFCDAC) 315, an internal regulated power supply voltage Vdd_int of about 2.45 V generated from the reference voltage generator RVG is supplied to a PMOS differential input circuit PMOS_DA, as illustrated in FIG. 10. Even in the CMOS rail-to-rail amplifier which constitutes the first voltage follower AMP1 and the second voltage follower AMP2 of the AFC-control D/A converter (AFCDAC) 315, the PMOS differential input circuit PMOS_DA is recommended to be supplied with the internal regulated power supply voltage Vdd_int of about 2.45 V generated from the reference voltage generator RVG, as illustrated in FIG. 10.

<<AFC-Control D/A Converter>>

FIG. 11 is a circuit diagram illustrating the constitution of the AFC-control D/A converter (AFCDAC) 315 arranged in the RF analog signal processing integrated circuit (RF_IC) 300 of the mobile terminal device illustrated in FIG. 1 as one embodiment of the present invention. The point that the AFC-control D/A converter (AFCDAC) 315 of FIG. 11 is different from the AFC-control D/A converter 315 illustrated in FIG. 2 is that the internal regulated power supply voltage Vdd_int maintained stable to about 2.45 V as well as the external power voltage Vdd_ext is supplied to three voltage followers AMP1, AMP2, AMP3 of the AFC-control D/A converter (AFCDAC) 315. As a result, inside the three voltage followers AMP1, AMP2, AMP3, as illustrated in FIG. 10, the external power voltage Vdd_ext is supplied to the NMOS differential input circuit NMOS_DA, the bias circuit BIAS_CKT, and the CMOS output circuit OUT_CKT, and the internal regulated power supply voltage Vdd_int of about 2.45 V generated from the reference voltage generator RVG is supplied to the PMOS differential input circuit PMOS_DA. The internal regulated power supply voltage V_REF of about 2.45 V generated together with the internal regulated power supply voltage Vdd_int from the reference voltage generator RVG is supplied to the top-stage resistance R of the first variable voltage divider VDIV1 of the AFC-control D/A converter (AFCDAC) 315 as a reference voltage.

In this way, in the CMOS rail-to-rail amplifier which constitutes at least the third voltage follower AMP3 of the AFC-control D/A converter (AFCDAC) 315, the internal regulated power supply voltage Vdd_int, which is generated from the reference voltage generator RVG and maintained almost stable, is supplied to the PMOS differential input circuit PMOS_DA. Therefore, even if the external power voltage Vdd_ext changes, the level of the internal regulated power supply voltage Vdd_int is suppressed to a very small change. As a result, increase of the current value of MP3 as a current source transistor of the PMOS differential input circuit PMOS_DA due to the change of the external power voltage Vdd_ext is also suppressed to a small increase. At this time, even if there is some shifting in the pair nature of the electrical property of MP1, MP2 of the PMOS differential input circuit PMOS_DA, the input offset voltage Vinoffset (P) of the PMOS differential input circuit PMOS_DA does not increase, and finally, the change ΔVout of the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter (AFCDAC) 315 can be reduced.

<<Simulation Result>>

FIG. 12 is a drawing of the simulation result illustrating change ΔVout of the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter (AFCDAC) 315 illustrated in FIG. 11, when the source of MP3, as a current source transistor of the PMOS differential input circuit PMOS_DA of the CMOS rail-to-rail amplifier illustrated in FIG. 10, is connected to the external power voltage Vdd_ext as usual, instead of the internal regulated power supply voltage Vdd_int. In this case, the change ΔVout of the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter (AFCDAC) 315 becomes as a large level of about from −5.0 mV to +5.0 mV, almost same as in FIG. 8.

FIG. 13 is a drawing of the simulation result illustrating change ΔVout of the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter 315 illustrated in FIG. 11, when the source of MP3, as a current source transistor of the PMOS differential input circuit PMOS_DA of the CMOS rail-to-rail amplifier illustrated in FIG. 10, is connected to the internal regulated power supply voltage Vdd_int according to the embodiment of the present invention, instead of the external power voltage Vdd_ext. The change ΔVout of the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter (AFCDAC) 315 in the case of the embodiment of the present invention is recognized to be suppressed to the almost same small level of from about −2.0 mV to +11.0 mV, when compared with the characteristics illustrated from FIG. 5 to FIG. 9, and in FIG. 12.

In addition, the change ΔVout of the AFC control analog output signal (V_TUNE) of the very small level of about −2.0 mV to +1.0 mV is thought to be due to the fact that the band-gap reference voltage Vref of about 1.23 V generated from the band-gap reference circuit BGR of the reference voltage generator RVG of the CMOS rail-to-rail amplifier of FIG. 10 changes to some extent due to the change of the external power voltage Vdd_ext.

<<Fractional-N PLL>>

FIG. 14 is a drawing illustrating the constitution of a fractional synthesizer Frct_Synth of the fractional-N PLL arranged in the RF analog signal processing integrated circuit (RF_IC) 300 of the mobile terminal device illustrated in FIG. 1 as one embodiment of the present invention.

As illustrated in FIG. 14, the fractional synthesizer Frct_Synth includes the reference frequency oscillator (VCXO) 314 which is set so as to generate a stable-and-accurate reference oscillating frequency f_REF by the crystal oscillator Xtal and the AFC control analog output signal (V_TUNE) of the AFC-control D/A converter (AFCDAC) 315. The reference oscillating frequency f_REF is set, for example, as the frequency of 26 MHz. The reference frequency signal of the reference oscillating frequency f_REF from the reference frequency oscillator (VCXO) 314 is supplied to one input terminal of a phase comparator PDC of the fractional PLL circuit. The output of the phase comparator PDC is supplied to an RF voltage controlled oscillator RFVCO via a charge pump circuit CPC and a low pass filter LFC. The output of the RF voltage controlled oscillator RFVCO is supplied to the input of a divider DIV. The dividing output signal of the divider DIV is supplied to another input terminal of the phase comparator PDC. The control input terminal of the divider DIV, which controls the dividing ratio, is connected to a division ratio setting logic DRSL. Channel selection information Channel_inf for RF communication from the baseband LSI (not shown) is supplied to the division ratio setting logic DRSL. The divider DIV is constructed by a counter. The divider DIV counts up, from zero for example, the change from a low level to a high level of the output of the RF voltage controlled oscillator RFVCO, and changes the dividing output signal of the divider DIV from a low level to a high level, at the frequency of a certain value minus one, the certain value being set at the control input terminal to control a dividing ratio. When the dividing output signal of the divider DIV becomes the high level, the count value of the counter is set to zero by the next change from the low level to the high level of the output of the RF voltage controlled oscillator RFVCO, the dividing output signal of divider DIV is returned to the low level, and the following dividing operation is performed. The division ratio setting logic DRSL is comprised of a division ratio arithmetic unit DRALU, a ΣΔ-modulator ΔΣMod, and an adder ADD. First, an integer unit Int and a fraction unit Fra of the division ratio arithmetic unit DRALU calculate the integral value information I and the fraction information F based on the inputted channel selection information Channel_inf. The integral value information I from the integer unit Int of the division ratio arithmetic unit DRALU is supplied to one input terminal of the adder ADD, the fraction information F from the fraction unit Fra of the division ratio arithmetic unit DRALU is supplied to the ΣΔ-modulator ΣΔMod. The reference frequency signal of the reference oscillating frequency f_REF from the reference frequency oscillator (VCXO) 314 is further supplied to the ΣΔ-modulator ΣΔMod as an operation clock signal. On the other hand, the ΣΔ-modulator ΣΔMod holds, as inside information, the denominator information G which sets up a dividing ratio. As an example, the denominator information G is set as 1625. From the fraction information F and the denominator information G, the ΣΔ-modulator ΣΔMod generates an output signal F/G which has information of the fraction information F divided by the denominator information G, a fraction of 403/1625 for example, and supplies the output signal F/G to the other input terminal of the adder ADD. From the integral value information I (I=137 as an example) and the output signal F/G, the adder ADD supplies the divider DIV with the output information of I+F/G (137+(403/1625)=137.248 for example), as an average division ratio N. As a result, the average division ratio of the divider DIV is set as the value of 137.248, containing an integer and a fraction (decimal). Therefore, the fractional synthesizer Frct_Synth generates the RF oscillation output signal of oscillating frequency f_RFVCO of 3568.448 MHz which is the product of the reference oscillating frequency f_REF (26 MHz) from the reference frequency oscillator (VCXO) 314 and the average division ratio N (137.248). If the average division ratio N is described in detail, a dividing ratio n of the divider DIV will be changed from n (=I=137) to n+1 (=I+1=138) in response to the integral value information I (I=137) from the integer unit Int of the division ratio arithmetic unit DRALU and the overflowed 1-bit output which occurs at a frequency of (403/1625) corresponding to the output signal F/G from the ΣΔ-modulator ΣΔMod. Therefore, the frequency at which the dividing ratio of the divider DIV is set to n (=I=137) is 1222/1625=75.2%, and the frequency at which the dividing ratio of divider DIV is set to n+1 (=I+1=138) is 403/1625=24.8%. Therefore, the average division ratio N is set to 137×0.752+138×0.248=137.248.

Frequency control of the transmission-system signal processing subunit of the communication integrated circuit RF IC is performed using the fractional-synthesizer Frct_Synth including the reference frequency oscillator (VCXO) 314. The closed-loop band of the fractional-N PLL circuit, which constitutes this fractional synthesizer Frct_Synth, is set as the order of several tens kHz, far lower than 100 kHz. To be specific, this closed-loop band is set as 30 kHz, for example. The transmission-system signal processing subunit includes a transmission-system offset PLL circuit TX_Offset_PLL. By supplying the RF oscillation output signal of the oscillating frequency f_RFVCO (3568.448 MHz), which is the output of the RF voltage controlled oscillator RFVCO of the fractional synthesizer Frct_Synth, to an intermediate frequency divider IF DIV with a dividing ratio of 26, a twofold intermediate frequency signal (137.248 MHz) is created from the output of the intermediate frequency divider IF DIV. By supplying this twofold intermediate frequency signal (137.248 MHz) to the input of a 90-degree phase shifter 90degShift, two intermediate frequency signals (68.624 MHz) which are mutually different in phase by 90 degrees are created. By supplying to transmission-mixers TX-MIX_I and TX-MIX_Q the two intermediate frequency signals (68.624 MHz) which are different in phase by 90 degrees from the baseband transmitting signals TxABI and TxABQ, the intermediate frequency transmitting signal (68.624 MHz) after vector composition is created in the output of an adder connected to the output of the transmission mixers TX-MIX_I and TX-MIX_Q. This intermediate frequency transmitting signal (68.624 MHz) is supplied to one input terminal of a phase comparator PC. By supplying the output of the phase comparator PC to an RF transmission voltage controlled oscillator TXVCO via a low pass filter LF1, the frequency of the RF transmission voltage controlled oscillator TXVCO is controlled to about 1715.6 MHz. The oscillation output signal of the RF transmission voltage controlled oscillator TXVCO is supplied to one input terminal of a phase-controlled-feedback frequency down mixer DWN_MIX_PM via a buffer amplifier BF. A down mixer RF signal (1784. 224 MHz) from an RF divider RF DIV with a dividing ratio of 2 is supplied to another input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM. In the phase-controlled-feedback frequency down mixer DWN_MIX_PM, mixing with the oscillation signal (about 1715.6 MHz) from the RF transmission voltage controlled oscillator TXVCO and the down mixer RF signal (1784.224 MHz) from the RF divider RF DIV is performed. Therefore, from the output of the phase-controlled-feedback frequency down mixer DWN_MIX_PM, the feedback signal or the frequency of a difference (1784.224 MHz-1715.6 MHz=68.624 MHz) is created, and supplied to another input terminal of the phase comparator PC. The transmission-system offset PLL circuit TX_Offset_PLL performs negative feedback control so that the phase and frequency of the two input signals of the phase comparator PC may be in agreement, consequently, the signal of accurate RF transmission frequency f_TXVCO of 1715.6 MHz can be obtained from the RF transmission voltage controlled oscillator TXVCO. One input terminal of the phase comparator PC is supplied with an intermediate frequency transmitting signal f_IF (68.624 MHz) which is the vector composition at the output of the adder connected to the output of the transmission-mixers TX-MIX_I and TX-MIX_Q. The other input terminal of the phase comparator PC is supplied with a difference frequency signal (f_RFVCO/2−f_TXVCO) which possesses a frequency after subtracting a frequency f_TXVCO of the RF transmission frequency signal of the RF transmission voltage controlled oscillator TXVCO from a dividing RF oscillating frequency f_RFVCO/2, which is the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO divided by the dividing ratio of 2. By the negative feedback control of the transmission-system offset PLL circuit TX_Offset_PLL, the reference frequency supplied to one input terminal of the phase comparator PC and the negative feedback frequency supplied to the other input terminal of the phase comparator PC becomes in agreement. Therefore, the following relation holds.

f _IF =f _RFVCO/2−f_TXVCO  (Eq. 5)

When the above equation is modified, the following equation is obtained.

$\begin{array}{cc}\begin{array}{c}{f}_{\mathrm{TXVCO}}=f_{\mathrm{RFVCO}}/2-f_{\mathrm{IF}}\\ =\left(3568.448\mathrm{MHz}/2\right)-68.624\mathrm{MHz}\\ =1784.224\mathrm{MHz}-68.624\mathrm{MHz}\\ =1715.6\mathrm{MHz}\end{array}& \left(\mathrm{Eq}.6\right)\end{array}$

Therefore, the RF transmission frequency f_TXVCO generated from the RF transmission voltage controlled oscillator TXVCO in the transmission-system offset PLL circuit TX_Offset_PLL is set up correctly, in response to the oscillating frequency f_RFVCO of the RF oscillation output signal generated from the RF voltage controlled oscillator RFVCO in the fractional synthesizer Frct_Synth and in response to the intermediate frequency transmitting signal f_IF of the output of the adder connected to the output of the transmission mixers. This intermediate frequency transmitting signal f_IF is also correctly set up by the RF transmission frequency f_TXVCO generated from the RF transmission voltage controlled oscillator TXVCO in the transmission-system offset PLL circuit TX_Offset_PLL.

On the other hand, the RF transmission signal generated from the RF transmission voltage controlled oscillator TXVCO in the transmission-system offset PLL circuit TX_Offset_PLL is transmitted to a base station from an antenna via an RF power amplifier and an antenna switch.

In the fractional-N PLL circuit of fractional synthesizer Frct_Synth illustrated in FIG. 14, it is possible to reduce the external-power-voltage-change-induced change of the AFC control analog output signal of the AFC-control D/A converter for controlling the oscillating frequency of the reference frequency oscillator, which is very important when obtaining sufficient frequency band and a precise frequency resolution to choose a desired channel and to catch modulation. Consequently, it is possible to improve the data precision in transmission and reception in the EDGE system with a great communication data rate, and also to reduce the level of the neighboring disturbance signal outside the regular frequency spectrum of the RF transmission signal at the time of transmission.

The neighboring disturbance signal outside the regular frequency spectrum of the RF transmission signal undergoes electric power amplification by the RF power amplifier connected to the output of the transmission-system signal processing subunit of RF IC, and is eventually transmitted as a neighboring disturbance signal from the antenna of the mobile phone terminal apparatus. The leaking signal component in the neighborhood (400 kHz) of the RF transmission frequency f_TXVCO (1715.6 MHz), which is the output signal of RF transmission voltage controlled oscillator TXVCO, is severely prescribed to be not more than the predetermined value (−60 dBm) by the standard of GMSK (Gaussian Minimum Shift Keying).

FIG. 18 is a drawing illustrating frequency spectra of an RF transmission signal of a mobile phone terminal apparatus specified by the standard of GMSK. In the figure, the thick solid line PSD is a level specified by the standard of GMSK. The amount of attenuation at ±200 kHz from the center frequency (RF transmission frequency) is specified to be less than −30 dBm, and the amount of attenuation at ±400 kHz from the center frequency (RF transmission frequency) is specified to be less than −60 dBm. The thin solid line illustrates the example which satisfies this standard. It becomes possible to satisfy the severe standard of GMSK by using the combination of the AFC-control D/A converter (AFCDAC) 315 of which the change ΔVout of the AFC control analog output signal (V_TUNE) illustrated in FIG. 10 is very small, and the fractional-N PLL circuit of fractional synthesizer Frct_Synth which has a sufficient frequency band and a precise frequency resolution illustrated in FIG. 14.

<<More Concrete Embodiment of the Present Invention>>

FIG. 15 is a drawing illustrating constitution of the RF analog signal processing integrated circuit RF IC according to a more concrete embodiment of the present invention. The RF IC illustrated in FIG. 15 is constituted so that it may support four bands or the quad bands of GSM 850 MHz, GSM 900 MHz, DCS 1800 MHz, and PCS 1900 MHz in both the receiving operation from a base station, and the send operation to the base station. DCS is the abbreviated name for Digital Cellular System, and PCS is the abbreviated name for Personal Communication System. In FIG. 15, Frct_Synth is an RF carrier synchronization subunit which is comprised of the fractional PLL circuit or the fractional synthesizer explained using FIG. 14.

The RF IC which supports the quad bands is comprised of the fractional synthesizer Frct_Synth explained in FIG. 14, the RF-reception-signal analog signal processing subunit RX SPU, and the RF-transmission-signal analog signal processing subunit TX SPU. An RF reception signal received with an antenna ANT of the mobile phone terminal apparatus is supplied to the RF-reception-signal analog signal processing subunit RX SPU via an antenna switch ANTSW and a surface acoustic wave filter SAW. By demodulating the inputted RF reception signal, the RF-reception-signal analog signal processing subunit RX SPU generates receiving baseband signals RxABI and RxABQ, and supplies the receiving baseband signals RxABI and RxABQ to the baseband LSI (BB_LSI). Transmitting baseband signals TxABI and TxABQ are supplied to the RF-transmission-signal analog signal processing subunit TX SPU from the baseband LSI (BB_LSI). By modulating the inputted transmitting baseband signals, the RF-transmission-signal analog signal processing subunit TXSPU creates an RF transmission signal, and supplies it to the antenna ANT of the mobile phone terminal apparatus via RF power amplifiers RF_PA1 and RD_PA2, and the antenna switch ANTSW.

First, the receiving operation of the RF-reception-signal analog signal processing subunit RX SPU is explained. The RF reception signal received with the antenna of the mobile phone terminal apparatus is supplied to four low-noise amplifiers via the antenna switch ANTSW and the surface acoustic wave filter SAW. The frequency band of the RF reception signal Rx_GSM850 of the band of GSM 850 MHz is 869 MHz-894 MHz, and is amplified by a first low-noise amplifier LNA1. The frequency band of RF reception signal Rx_GSM900 of the band of GSM 900 MHz is 925 MHz-960 MHz, and is amplified by a second low-noise amplifier LNA2. The frequency band of RF reception signal Rx_DCS1800 of the band of DCS 1800 MHz is 1805 MHz-1880 MHz, and is amplified by a third low-noise amplifier LNA3. The frequency band of RF reception signal Rx_PCS1900 of the band of PCS 1900 MHz is 1930 MHz-1990 MHz, and is amplified by a fourth low-noise amplifier LNA4. The RF-amplified reception output signal of the four low-noise amplifiers LNA1-LNA4 is supplied to one input terminal of two mixing circuits RX-MIX_I and RX-MIX_Q which serve as a receive mixer. Two RF carrier signals which are created by a 90-degree phase shifter 90degShift (½) and have about 90-degree phases are supplied to another input terminal of the two mixing circuits RX-MIX_I and RX-MIX_Q. In the receive mode of GSM 850 MHz or GSM 900 MHz, the output of an RF voltage controlled oscillator RFVCO is supplied to the 90-degree phase shifter 90degShift (½) via a ½ divider DIV1 with a dividing ratio of 2. In the receive mode of DCS 1800 MHz or PCS 1900 MHz, the output of the RF voltage controlled oscillator RFVCO is directly supplied to the 90-degree phase shifter 90degShift (½). From the output of the mixing circuit RX-MIX_I, and the output of the mixing circuit RX-MIX_Q, a receiving baseband signal RxABI and a receiving baseband signal RxABQ are generated, respectively. The receiving baseband signal RxABI and the receiving baseband signal RxABQ are supplied to the baseband LSI (BB_LSI) via variable gain amplifiers PGAI1, PGAI2, PGAI3, filter circuits FCI1, FCI2, FCI3 and a buffer amplifier BAI, and via variable gain amplifiers PGAQ1, PGAQ2, PGAQ3, filter circuits FCQ1, FCQ2, FCQ3 and a buffer amplifier BAQ, respectively.

In order to support the receiving operation in 869 MHz-894 MHz of the frequency band of the RF reception signal Rx_GSM850 of the band of GSM 850 MHz, the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is set as 3476 MHz-3576 MHz. The oscillating frequency f_RFVCO of this RF voltage controlled oscillator RFVCO undergoes (¼)-dividing by the divider DIV1 (½) with the dividing ratio set as 2 and the 90-degree phase shifter 90degShift (½). The RF divided frequency signal of 869 MHz-894 MHz after the (¼)-dividing is supplied to the two mixing circuits RX-MIX_I and RX-MIX_Q which constitute a receive mixer. Thereby, analog baseband reception signals RxABI and RxABQ after the reception of the RF reception signal Rx_GSM850 of the band of GSM 850 MHz are created from the output of two mixing circuits RX-MIX_I and RX-MIX_Q. In order to support the receiving operation in 925 MHz-960 MHz of the frequency band of RF reception signal Rx_GSM900 of the band of GSM 900 MHz, the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is set as 3700 MHz-3840 MHz. The oscillating frequency f_RFVCO of this RF voltage controlled oscillator RFVCO undergoes (¼)-dividing by the divider DIV1 (½) with the dividing ratio set as 2 and the 90-degree phase shifter 90degShift (½). The RF divided frequency signal of 925 MHz-960 MHz after the (¼)-dividing is supplied to the two mixing circuits RX-MIX_I and RX-MIX_Q which constitute the receive mixer. Thereby, analog baseband reception signals RxABI and RxABQ after the reception of the RF reception signal Rx_GSM900 of the band of GSM 900 MHz are created from the output of two mixing circuits RX-MIX_I and RX-MIX_Q. In order to support the receiving operation in 1805 MHz-1880 MHz of the frequency band of RF reception signal Rx_DCS1800 of the band of DCS 1800 MHz, the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is set as 3610 MHz-3760 MHz. The oscillating frequency f_RFVCO of this RF voltage controlled oscillator RFVCO undergoes (½)-dividing by the 90-degree phase shifter 90degShift (½). The RF divided frequency signal of 1805 MHz-1880 MHz after the (½)-dividing is supplied to the two mixing circuits RX-MIX_I and RX-MIX_Q which constitute the receive mixer. Thereby, analog baseband reception signals RxABI and RxABQ after the reception of the RF reception signal Rx_DCS1800 of the band of DCS 1800 MHz are created from the output of two mixing circuits RX-MIX_I and RX-MIX_Q. In order to support the receiving operation in 1930 MHz-1990 MHz of the frequency band of RF reception signal Rx_PCS1900 of the band of PCS 1900 MHz, the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is set as 3860 MHz-3980 MHz. The oscillating frequency f_RFVCO of this RF voltage controlled oscillator RFVCO undergoes (½)-dividing by the 90-degree phase shifter 90degShift (½). The RF divided frequency signal of 1930 MHz-1990 MHz after the (½)-dividing is supplied to two mixing circuits RX-MIX_I and RX-MIX_Q which constitute the receive mixer. Thereby, analog baseband reception signal RxABI and RxABQ after the reception of the RF reception signal Rx_PCS1900 of the band of PCS 1900 MHz are created from the output of two mixing circuits RX-MIX_I and RX-MIX_Q.

Next, the send operation of the RF-transmission-signal analog signal processing subunit TX SPU is explained. By supplying the RF oscillation output signal of the output of the RF voltage controlled oscillator RFVCO of the fractional synthesizer Frct_Synth to an intermediate frequency divider DIV2 (1/N_IF) with the predetermined dividing ratio, a twofold intermediate frequency signal is created from the output of the intermediate frequency divider DIV2 (1/N_IF). By supplying this twofold intermediate frequency signal to the input of a 90-degree phase shifter 90degShift, two intermediate frequency signals of 68.624 MHz with different phases by about 90 degrees are created. By supplying the transmission mixers TX-MIX_I and TX-MIX_Q with the baseband transmitting signals TxABI and TxABQ from the baseband LSI (BB_LSI) and the two intermediate frequency signals of 68.624 MHz with different phases by about 90 degrees, the 68.624-MHz intermediate frequency transmitting signal after vector composition is created at the output of the adder connected to the output of the transmission mixers TX-MIX_I and TX-MIX_Q. This 68.624 MHz intermediate frequency transmitting signal is supplied to one input terminal of the phase comparator PC. By supplying the output of the phase comparator PC to the RF-transmission voltage controlled oscillator TXVCO via the low pass filter LPF1, the oscillating frequency of the RF-transmission voltage controlled oscillator TXVCO is controlled to about 3431.2 MHz. The frequency bands of the RF transmission signal Tx_GSM850 of the band of GSM 850 MHz are 824 MHz-849 MHz. The oscillation output signals of 3296 MHz-3396 MHz of the RF-transmission voltage controlled oscillator TXVCO are supplied to the input of the first RF power amplifier RF_PA1 via two dividers DIV5 (½) and DIV3 (½), each with the dividing ratio of 2. The frequency bands of the RF transmission signal Tx_GSM900 of the band of GSM 900 MHz are 880 MHz-915 MHz. The oscillation output signals of 3520 MHz-3660 MHz of the RF-transmission voltage controlled oscillator TXVCO are supplied to the input of the first RF power amplifier RF_PA1 via two dividers DIV5 (½) and DIV3 (½), each with the dividing ratio of 2. The frequency bands of the RF transmission signal Tx_DCS1800 of the band of DCS 1800 MHz are 1710 MHz-1785 MHz. The oscillation output signals of 3420 MHz-3570 MHz of the RF-transmission voltage controlled oscillator TXVCO are supplied to the input of the second RF power amplifier RF_PA2 via one divider DIV5 (½) with the dividing ratio of 2. The frequency bands of the RF transmission signal Tx_PCS1900 of the band of PCS 1900 MHz are 1850 MHz-1910 MHz. The oscillation output signals of 3700 MHz-3820 MHz of the RF-transmission voltage controlled oscillator TXVCO are supplied to the input of the second RF power amplifier RF PA2 via one divider DIV5 (½) with the dividing ratio of 2.

It is necessary to support the send operation with the frequency band 824 MHz-848 MHz of the RF transmission signal Tx_GSM850 of the band of GSM 850 MHZ and the frequency band 880 MHZ-915 MHz of the RF transmission signal Tx_GSM900 of the band of GSM 900 MHz. Therefore, the oscillating frequency f_RFVCO Of the RF voltage controlled oscillator RFVCO is supplied to one input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM of the transmission-system offset PLL circuit TX_Offset_PLL via two dividers DIV1 (½) and DIV4 (½), each with the dividing ratio of 2. Dividing ratio N_IF is set as 26 for the intermediate frequency divider DIV2 (1/N_IF) which is connected to the 90-degree phase shifter 90degShift (½) connected to two mixing circuits TX-MIX_I and TX-MIX_Q constituting the transmission mixer of the transmission-system offset PLL circuit TX_Offset PLL. Therefore, the oscillation output signal of the oscillating frequency f_TXVCO of the RF transmission voltage controlled oscillator TXVCO is supplied to one input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM, via two dividers DIV5 (½) and DIV3 (½), each with the dividing ratio of 2. A (¼)-dividing signal of the oscillating frequency f_RFVCO Of the RF voltage controlled oscillator RFVCO is supplied to the other input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM via two dividers DIV1 (½) and DIV4 (½). In the phase-controlled-feedback frequency down mixer DWN_MIX_PM, mixing with a (¼)-dividing signal of the oscillating frequency f_RFVCO and a (¼)-dividing signal of the oscillation output signal of the oscillating frequency f_TXVCO of the RF transmission voltage controlled oscillator TXVCO is performed. Therefore, from the output of the phase-controlled-feedback frequency down mixer DWN_MIX_PM, the feedback signal of the frequency of difference of ((¼)×f_RFVCO−(¼)×f_TXVCO) is created, and supplied to the other input terminal of the phase comparator PC of the transmission-system offset PLL circuit TX_Offset_PLL. One input terminal of the phase comparator PC is supplied with an intermediate frequency transmitting signal f_IF after vector composition of the output of the adder connected to the output of the transmission mixers TX-MIX_I and TX-MIX_Q. This intermediate frequency transmitting signal f_IF is set to f_RFVCO/52 by the dividing ratios N_IF of 26 of the intermediate frequency divider DIV2 (1/N_IF) and the (½)-dividing function in the 90-degree phase shifter 90degShift. The reference frequency of one input terminal of the phase comparator PC and the negative feedback frequency of the other input terminal of the phase comparator PC becomes in agreement by the negative feedback control of the transmission-system offset PLL circuit TX_Offset_PLL. Accordingly, the following relation holds.

$\begin{array}{c}{f}_{\mathrm{RFVCO}}/52=\left(1/4\right)\times f_{\mathrm{RFVCO}}-\left(1/4\right)\times f_{\mathrm{TXVCO}}\left(1/4\right)\times f_{\mathrm{TXVCO}}\\ =\left(1/4\right)\times f_{\mathrm{RFVCO}}-f_{\mathrm{RFVCO}}/52\\ =\left(\left(13-1\right)/52\right)\times f_{\mathrm{RFVCO}}\\ =\left(12/52\right)\times f_{\mathrm{RFVCO}}\end{array}\therefore f_{\mathrm{RFVCO}}=4.33333\times \left(1/4\right)\times f_{\mathrm{TXVCO}}$

Therefore, it is necessary to support the send operation with the frequency band 824 MHz-848 MHz of the RF transmission signal Tx_GSM850 of the band of GSM 850 MHz, and the frequency band 880 MHz-915 MHz of the RF transmission signal Tx_GSM900 of the band of GSM 900 MHz. For this purpose, it is just good to set up the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO as 4.33333 times the (¼)-dividing signal ((¼)×f_TXVCO) of the oscillating frequency f_TXVCO of the RF transmission voltage controlled oscillator TXVCO. Therefore, it is just good to set the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO as 3570.6639 MHz-3678.9971 MHz corresponding to the frequency band 824 MHz-849 MHz of the RF transmission signal Tx_GSM850 of the band of GSM 850 MHz. It is just good to set the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO as 3813.3304 MHz-3974.997 MHz corresponding to the frequency band 880 MHz-915 MHz of the RF transmission signal Tx_GSM900 of the band of GSM 900 MHz.

It is necessary to support the send operation with the frequency band 1710 MHz-1785 MHz of the RF transmission signal Tx_DCS1800 of the band of DCS 1800 MHz and the frequency band 1850 MHz-1910 MHz of the RF transmission signal Tx_PCS1900 of the band of PCS 1900 MHz. Therefore, the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is supplied to one input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM of the transmission-system offset PLL circuit TX_Offset_PLL, via one divider DIV1 (½) with the dividing ratio of 2. Dividing ratio N_IF is set as 26 for the intermediate frequency divider DIV2 (1/N_IF) which is connected to the 90-degree phase shifter 90degShift (½) connected to two mixing circuits TX-MIX_I and TX-MIX_Q constituting the transmission mixer of the transmission-system offset PLL circuit TX_Offset_PLL. Therefore, the oscillation output signal of the oscillating frequency f_TXVCO of the RF-transmission voltage controlled oscillator TXVCO is supplied to one input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM via one divider DIV5 (½) with the dividing ratio of 2. A (½)-dividing signal of the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO is supplied to the other input terminal of the phase-controlled-feedback frequency down mixer DWN_MIX_PM via one divider DIV1 (½). In the phase-controlled-feedback frequency down mixer DWN_MIX_PM, mixing with a (½)-dividing signal of the oscillating frequency f_RFVCO and a (½)-dividing signal of the oscillation output signal of the oscillating frequency f_TXVCO of the RF transmission voltage controlled oscillator TXVCO is performed. Therefore, from the output of the phase-controlled-feedback frequency down mixer DWN_MIX_PM, the feedback signal of the frequency of difference of ((½)×f_RFVCO−(½)×f_TXVCO) is created and supplied to the other input terminal of the phase comparator PC of the transmission-system offset PLL circuit TX_Offset_PLL. One input terminal of the phase comparator PC is supplied with the intermediate frequency transmitting signal f_IF after vector composition of the output of the adder connected to the output of the transmission mixer TX-MIX_I and TX-MIX_Q. This intermediate frequency transmitting signal f_IF is set to f_RFVCO/52 by the dividing ratio N_IF of 26 of the intermediate frequency divider DIV2 (1/N_IF) and the (½)-dividing function in the 90-degree phase shifter 90degShift. The reference frequency of one input terminal of the phase comparator PC and the negative feedback frequency of the other input terminal of the phase comparator PC become in agreement by the negative feedback control of the transmission-system offset PLL circuit TX_Offset_PLL. Accordingly, the following relation holds.

$\begin{array}{c}{f}_{\mathrm{RFVCO}}/52=\left(1/2\right)\times f_{\mathrm{RFVCO}}-\left(1/2\right)\times f_{\mathrm{TXVCO}}\left(1/2\right)\times f_{\mathrm{TXVCO}}\\ =\left(1/2\right)\times f_{\mathrm{RFVCO}}-f_{\mathrm{RFVCO}}/52\\ =\left(\left(26-1\right)/52\right)\times f_{\mathrm{RFVCO}}\\ =\left(25/52\right)\times f_{\mathrm{RFVCO}}\end{array}\therefore f_{\mathrm{RFVCO}}=2.08\times \left(1/2\right)\times f_{\mathrm{TXVCO}}$

Therefore, it is necessary to support the send operation with the frequency band 1710 MHz-1785 MHz of the RF transmission signal Tx_DCS1800 of the band of DCS 1800 MHz, and the frequency band 1850 MHz-1910 MHz of the RF transmission signal Tx_PCS1900 of the band of PCS 1900 MHz. Therefore, it is just good to set up the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO to 2.08 times the (½)-dividing signal ((½)×f_TXVCO) of the oscillating frequency f_TXVCO of the RF transmission voltage controlled oscillator TXVCO. Therefore, it is just good to set the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO as 3556.8 MHz-3712.8 MHz corresponding to the frequency band 1710 MHz-1785 MHz of the RF transmission signal Tx_DCS1800 of the band of DCS 1800 MHz. It is just good to set the oscillating frequency f_RFVCO of the RF voltage controlled oscillator RFVCO as 3848 MHz-3972.8 MHz corresponding to the frequency band 1850 MHz-1910 MHz of the RF transmission signal Tx_PCS1900 of the band of PCS 1900 MHz.

FIG. 16 is a drawing illustrating constitution of the RF analog signal processing integrated circuit RF IC according to still more concrete embodiment of the present invention.

This RF IC adopts the transmission system of the polar loop system for supporting the EDGE system which uses amplitude modulation as well as phase modulation in communication between a base station and a communication terminal apparatus.

One semiconductor chip of RF IC contains three subunits Frct_Synth, RX SPU, and TX SPU. In FIG. 16, besides the RF IC, an antenna ANT for transmission and reception and a front-end module FEM of the mobile phone terminal apparatus are also illustrated. The front-end module FEM includes an antenna switch ANT_SW, a transmission RF power amplifier RF_PA, and a power coupler CPL for detecting the transmission power from the transmission RF power amplifier RF_PA.

In FIG. 16, Frct_Synth is an RF carrier synchronization subunit which is comprised of the fractional PLL circuit or the fractional synthesizer explained using FIG. 14. In the RF carrier synchronization subunit Frct_Synth, the fractional frequency synthesizer is supplied with a system reference clock signal from a system reference clock oscillator VCXO of which the oscillating frequency f_REF is stably maintained with a crystal oscillator Xtal arranged outside the integrated circuit RF IC, thereby, the fractional frequency synthesizer can also maintain the RF oscillating frequency f_RFVCO of the RF oscillator RFVCO stably. By supplying the RF output of the RF oscillator RFVCO to a divider DIV1 (DIV4) (½ or ¼), an RF signal ΦRF is obtained from the output of the divider DIV1 (DIV4) (½ or ¼). This RF signal ΦRF is supplied to the RF-reception-signal analog signal processing subunit RX SPU and the RF-transmission-signal analog signal processing subunit TX SPU in the analog signal processing integrated circuit for RF communication RF IC. That is, the RF-transmission-signal analog signal processing subunit 302 TX SPU is constituted by the polar loop system for supporting the EDGE system.

In the time slot set as the receive state, the antenna switch ANT_SW of the front-end module FEM is connected to the upper side. Therefore, the RF reception signal received with the antenna ANT is supplied to the input of the low-noise amplifier LNA of the RF-reception-signal analog signal processing subunit RX SPU via the receive filter SAW which is comprised of the surface acoustic wave device, for example. The RF-amplified output of the low-noise amplifier LNA is supplied to one input of two mixing circuits RX-MIX_I and RX-MIX_Q which constitute a receive mixer. The other input of the two mixing circuits RX-MIX_I and RX-MIX_Q is supplied with two RF carrier signals which have a phase of 90 degrees and is created by a 90-degree phase shifter 90degShift (½) based on the RF signal ΦRF from the divider DIV1 (DIV4) (½ or ¼). Consequently, in the mixing circuits RX-MIX_I and RX-MIX_Q of the receive mixer, the direct frequency down-conversion from the RF reception signal frequency to a baseband signal frequency is performed, and received analog baseband signals RxABI and RxABQ are obtained from the output thereof. After the received analog baseband signals RxABI and RxABQ are amplified by variable gain amplifiers PGAI1, PGAI2, PGAI3, PGAQ1, PGAQ2, and PGAQ3, of which gain is adjusted by the receive time-slot setup, then the received analog baseband signals RxABI and RxABQ are converted into a digital signal by an A/D converter in the chip of RF IC. The digital reception signal is supplied to a baseband signal processing LSI (not shown).

In the time slot set as the send state, a digital transmission baseband signal is supplied to RF IC from a baseband signal processing LSI (not shown). Consequently, from the output of a D/A converter (not shown) in RF IC, an analog baseband transmitting signals TxABI and TxABQ are supplied to one input of two mixing circuits TX-MIX_I and TX-MIX_Q of the transmission mixer of the RF-transmission-signal analog signal processing subunit TX SPU. A signal ΦIF of the intermediate frequency f_IF is obtained by dividing an RF oscillating frequency f_RFVCO Of the RF oscillator RFVCO by an intermediate frequency divider DIV2 (1/N_IF). Two IF transmission carrier signals which have the 90-degree phase and are created by a 90-degree phase shifter 90degShift based on the IF signal ΦIF are supplied to the other input of the two mixing circuits TX-MIX_I and TX-MIX_Q. Consequently, in the mixing circuits TX-MIX_I and TX-MIX_Q of the transmission mixer, the frequency up-conversion from the frequency of the analog baseband transmitting signal to the IF transmitting signal is performed, and one IF transmitting modulation signal after vector composition is obtained from an adder. The IF transmitting modulation signal from the adder is supplied to one input of a phase comparator PC which constitutes a PM loop circuit PM LP for transmission of the phase modulation component in the RF-transmission-signal analog signal processing subunit TX SPU. In the PM loop circuit PM LP, the output of the phase comparator PC is transmitted to the control input of a transmission voltage controlled oscillator TXVCO via a charge pump CP and a low pass filter LF1.

The operating voltage from a voltage regulator Vreg is supplied to a buffer amplifier BF of which the input is connected to the output of the transmission voltage controlled oscillator TXVCO. The output of the transmission voltage controlled oscillator TXVCO is supplied to the input of a frequency down mixer DWN_MIX_PM for the PM loop to which the RF signal ΦRF is also supplied from the divider DIV1 (DIV4) (½ or ¼). Thereby, a first IF transmission feedback signal is obtained from the output of DWN_MIX_PM. As the phase modulation information in the case where the transmission time slot is the GSM system, this first IF transmission feedback signal is supplied via a switch SW_1 to another input of the phase comparator PC which constitutes the PM loop circuit PM LP. As a result, the sending signal which is an output of the transmission RF power amplifier RF_PA comes to contain the accurate phase modulation information of the GSM system. Transmission power information (amplification gain of the transmission RF power amplifier RF_PA) in the case where the transmission time slot is the GSM system is specified by a ramp output voltage Vramp of a ramp-signal D/A converter Ramp DAC in RF IC. This ramp output voltage Vramp is supplied to a 10 MHz filter (10 MHz Filter) via a switch SW2. The ramp output voltage Vramp from the filter and a transmission power detecting signal Vdet from the power coupler CPL and a power detection circuit PDET, which detect the transmission power of the transmission RF power amplifier RF_PA, are supplied to an error amplifier Err_Amp. The amplification gain of the transmission RF power amplifier RF_PA is set up, in proportion to the distance between the base station and the portable communication terminal, by the power supply voltage control or bias voltage control based on an automatic power control voltage Vapc from the output of the error amplifier Err_Amp. In addition, the digital ramp input signal supplied to the ramp-signal D/A converter Ramp DAC from a baseband signal processing unit like the baseband LSI is a transmission power level indication signal which indicates the level of transmission power, and controls the transmission power level high in proportion to the distance between the base station and the communication terminal apparatus. The ramp output voltage Vramp in an analog form is generated from the output of the ramp-signal D/A converter Ramp DAC.

When the transmission time slot is the EDGE system, on the other hand, the IF transmitting modulation signal from the adder will include not only the phase modulation information but also the amplitude modulation information. Therefore, the IF transmitting modulation signal from the adder is not only supplied to one input of the phase comparator PC which constitutes the PM loop circuit PM LP, but is supplied to one input of an amplitude comparator AC which constitutes an AM loop circuit AM LP. At this time, the other input of the phase comparator PC is not supplied with the output of the transmission voltage controlled oscillator TXVCO via the PM-loop frequency down mixer DWN_MIX_PM. Rather, the other input of the phase comparator PC will be supplied with the information related to the transmission power of the transmission RF power amplifier RF_PA (RF transmission power level RFPLV), via the power coupler CPL, a variable gain circuit MVGA, and an AM-loop frequency down mixer DWN_MIX_AM. The other input of the amplitude comparator AC which constitutes the AM loop circuit AM LP is supplied with the information related to the transmission power of the transmission RF power amplifier RF_PA (RF transmission power level RFPLV), via the power coupler CPL, the variable gain circuit MVGA, and the AM-loop frequency down mixer DWN_MIX AM. In the AM loop circuit AM LP, the output of the amplitude comparator AC is supplied to the 10 MHz filter (10 MHz Filter) via a low pass filter LF2, a variable gain circuit IVGA, a voltage/current converter V/I, a charge pump CP, and the switch WS2. As a result, the transmission power signal of the output of the transmission RF power amplifier RF_PA which amplifies the RF oscillation output signal of the transmission voltage controlled oscillator TXVCO comes to contain the accurate phase modulation information of the EDGE system first by the PM loop circuit PM LP. Furthermore, the transmission power signal of the output of the transmission RF power amplifier RF_PA comes to contain the accurate amplitude modulation information of the EDGE system by the AM loop circuit AM LP.

As the power coupler CPL which detects the transmission power of the transmission RF power amplifier RF_PA, a coupler which detects the transmission power of the RF power amplifier RF_PA electromagnetically or in terms of capacitance may be employed. As the power coupler CPL, a current-sense-type coupler is also employable in addition to the above examples. In the current-sense-type coupler, a detection DC-AC operating current, of small amplitude but proportional to the DC-AC operating current of the final stage power amplifying element of the RF power amplifier RF_PA, is flowed through a detection amplifying element.

In RF IC of FIG. 16, a control circuit CNTL generates two 8-bit control signals in response to the 10-bit digital ramp signal, so that the gain of two variable gain circuits MVGA and IVGA of the AM loop circuit AM LP, which respond to the ramp voltage Vramp of the ramp-signal D/A converter Ramp DAC, may have an opposite slop each other. That is, in response to the ramp voltage Vramp, when the gain of the variable gain circuit MVGA decreases, the gain of the variable gain circuit IVGA increases, thereby the sum of the gain of two variable gain circuits MVGA and IVGA becomes almost constant. As a result, the phase margin of the open-loop frequency characteristic of the AM loop circuit AM LP is avoided to become too small in response to the ramp voltage Vramp.

FIG. 17 is a drawing illustrating constitution of a different RF IC from the RF IC which adopts the transmission system of the polar loop system as illustrated in FIG. 16, in order to support the EDGE system which uses amplitude modulation as well as phase modulation. That is, RF IC illustrated in FIG. 17 adopts the transmission system of the polar modulator system in order to support the EDGE system in which communication with the base station uses amplitude modulation as well as phase modulation, and the RF-transmission-signal analog signal processing subunit TX SPU is constituted by the polar modulator system to support the EDGE system.

That is, an amplitude-modulation-loop control circuit AM_LP controls the amplitude of an RF transmission output signal from a transmission RF power amplifier RF_PA, based on a transmission intermediate frequency signal created by transmission modulation circuits TX_MIX_I and TX_MIX_Q. The amplitude-modulation-loop control circuit AM_LP is constituted as follows.

In the AM loop circuit AM LP, the output of an amplitude comparator AC is supplied to an amplitude-modulation variable gain amplifier VGA which is inserted between the output of a buffer amplifier BF and the input of a transmission voltage controlled oscillator TXVCO, via a Low pass filter LF2, a variable gain circuit IVGA, a voltage/current converter V/I, and a charge pump CP. One input terminal of the amplitude comparator AC of the AM loop circuit AM LP is supplied with the transmission intermediate frequency signal created by the transmission modulation circuit (TX_MIX_I, TX_MIX_Q). The other input terminal of the amplitude comparator AC is supplied with the information related to the transmission power of a transmission RF power amplifier RF_PA (RF transmission power level RFPLV) via a power coupler CPL, a variable gain circuit MVGA, and an AM-loop frequency down-mixer DWN_MIX_AM. As a result, the gain of the amplitude-modulation variable gain amplifier VGA is controlled by the output of the amplitude comparator AC, via the low pass filter LF2, the variable gain circuit IVGA, the voltage/current converter V/I, and the charge pump CP, so that the IF signal amplitude of one input terminal of the amplitude comparator AC may be in agreement with the IF signal amplitude of the other input terminal thereof, wherein the amplitude-modulation variable gain amplifier VGA is inserted between the output of the buffer amplifier BF and the input of the transmission voltage controlled oscillator TXVCO. Consequently, the transmission power of the transmission RF power amplifier RF_PA will contain the accurate amplitude modulation information of the EDGE system.

In the case of the GSM system and in the case of the EDGE system, the ramp output voltage Vramp of the ramp-signal D/A converter Ramp DAC and the transmission power detecting signal Vdet from the power coupler CPL and the power detection circuit PDET, which detect the transmission power of the transmission RF power amplifier RF_PA, are supplied to the error amplifier Err_Amp. By the power supply voltage control or the bias voltage control in terms of an automatic power control voltage Vapc from the output of the error amplifier Err_Amp, the amplification gain of the transmission RF power amplifier RF_PA is set up in proportion to the distance of a base station and a portable communication terminal. Consequently, the APC control is successfully performed.

FIG. 19 is a circuit diagram illustrating other constitution of the CMOS rail-to-rail amplifier which includes three voltage followers (AMP1, AMP2, AMP3) of AFC-control D/A converter (AFCDAC) 315 of the mobile terminal device of FIG. 1 as one embodiment of the present invention. It is mainly the constitution of a CMOS output circuit OUT_CKT that the CMOS rail-to-rail amplifier illustrated in FIG. 19 is different from the CMOS rail-to-rail amplifier illustrated in FIG. 10. In the CMOS output circuit OUT_CKT illustrated in FIG. 19, MP6 and MP7 of PMOS as the load elements of MN1 and MN2 of NMOS of an NMOS differential input circuit (NMOS_DA) are connected in series with MP10 and MP11 of other PMOS, respectively. MN8 and MN9 of NMOS as the load element of MP1 and MP2 of PMOS of a PMOS differential input circuit (PMOS_DA) are connected in series with MN12 and MN13 of other NMOS, respectively. A first bias voltage Vb1 is supplied to the gates of MN8 and MN9 of NMOS, a second bias voltage Vb2 is supplied to the gates of MN12 and MN13 of NMOS, and a third bias voltage Vb3 is supplied to the gates of MP10 and MP11 of PMOS.

Therefore, the NMOS differential pair MN1 and MN2 of the NMOS differential input circuit (NMOS_DA) pulls up the voltage of an output terminal (Vout) to the direction of the external power voltage (Vdd_ext), in response to the differential input signal of a high level mainly from near the intermediate level between a non-inverted input terminal (Vinp) and an inverted input terminal (Vinn), due to the high degree of conduction of MP7 of PMOS of the CMOS output circuit (OUT_CKT). For example, if the voltages of the inverted input terminal (Vinn) and the output terminal (Vout) are low and a comparatively-high-level analog input voltage is supplied to the non-inverted input terminal (Vinp), MN1 and MN2 will be in an ON state and an OFF state, respectively, and the voltage of the output terminal (Vout) can be pulled up to the direction of the external power voltage (Vdd_ext) due to the high degree of conduction of MP7 of PMOS of the CMOS output circuit (OUT_CKT). On the contrary, the PMOS differential pair MP1 and MP2 of the PMOS differential input circuit (PMOS_DA) pulls down the voltage of the output terminal (Vout) to the direction of the ground potential (GND), in response to the low differential input signal mainly from near the intermediate level between the non-inverted input terminal (Vinp) and the inverted input terminal (Vinn), due to the high degree of conduction of MN9 of NMOS of the CMOS output circuit (OUT_CKT). For example, if the voltages of the inverted input terminal (Vinn) and the output terminal (Vout) are high in level and the comparatively-low analog input voltage is supplied to the non-inverted input terminal (Vinp), MP1 and MP2 will be in an ON state and an OFF state, respectively, and the voltage of the output terminal (Vout) can be pulled down to the direction of the ground potential (GND) due to the high degree of conduction of MN9 of NMOS of the CMOS output circuit (OUT_CKT). When the voltage of both the inputs of the non-inverted input terminal (Vinp) and the inverted input terminal (Vinn) is near the intermediate level, the NMOS differential input circuit (NMOS_DA) and the PMOS differential input circuit (PMOS_DA) illustrated in FIG. 19 perform a voltage follower operation jointly, rendering the voltage level of the inverted input terminal (Vinn) and the output terminal (Vout) to follow the voltage level of the non-inverted input terminal (Vinp).

The invention made by the present inventors has been concretely explained based on the embodiment in the above. However, the present invention is not limited to what has been described, and it cannot be overemphasized that the present invention can change variously in the range which does not deviate from the gist.

In the above-mentioned embodiment, the baseband signal processing LSI 400 and the application processor are constituted by the individually different semiconductor chip. However, in another embodiment, the application processor may be included in the semiconductor chip of the baseband signal processing LSI 400, thereby providing an integrated one-chip.

Claims

1. A semiconductor integrated circuit for RF communications to perform bidirectional signal transfer using a digital interface and an LSI which performs at least a baseband digital signal processing, the semiconductor integrated circuit for RF communications comprising: an RF-reception-signal analog signal processing subunit; andan RF-transmission-signal analog signal processing subunit,the RF-reception-signal analog signal processing subunit being operable to perform frequency down-conversion of an RF reception signal to an analog baseband reception signal, and the RF-transmission-signal analog signal processing subunit being operable to perform frequency up-conversion of an analog baseband transmitting signal to an RF transmission signal,the semiconductor integrated circuit for RF communications further including:a reference frequency oscillator operable to generate a reference frequency signal for generating a high frequency signal to be used in the frequency down-conversion performed in the RF-reception-signal analog signal processing subunit and in the frequency up-conversion performed in the RF-transmission-signal analog signal processing subunit; andan AFC-control D/A converter operable to control frequency of the reference frequency signal generated by the reference frequency oscillator, by converting an AFC control digital input signal supplied from the LSI into an AFC control analog output signal,wherein the AFC-control D/A converter includes:a first variable voltage divider operable to generate an analog rough selection voltage in response to a higher-order bit of the AFC control digital input signal;a first voltage follower operable to be supplied with a voltage of one side of the analog rough selection voltage;a second voltage follower operable to be supplied with a voltage of another side of the analog rough selection voltage;a second variable voltage divider operable to be supplied with an output voltage of the first voltage follower and an output voltage of the second voltage follower and to generate an analog fine selection voltage in response to a lower-order bit of the AFC control digital input signal; anda third voltage follower operable to be supplied with an output voltage of the second variable voltage divider, wherein each voltage follower of the first voltage follower, the second voltage follower, and the third voltage follower included in the AFC-control D/A converter is comprised of a CMOS rail-to-rail amplifier,wherein the CMOS rail-to-rail amplifier is comprised of an NMOS differential input circuit, a PMOS differential input circuit, a CMOS output circuit, and a bias circuit,wherein a non-inverted input terminal of the CMOS rail-to-rail amplifier is coupled to a gate of a first NMOS in the NMOS differential input circuit and to a gate of a first PMOS in the PMOS differential input circuit, an inverted input terminal of the CMOS rail-to-rail amplifier is coupled to an output terminal and to a gate of a second NMOS in the NMOS differential input circuit and to a gate of a second PMOS in the PMOS differential input circuit, a source of the first NMOS and a source of the second NMOS in the NMOS differential input circuit are coupled to a drain of a third NMOS serving as a first current source transistor, a source of the first PMOS and a source of the second PMOS in the PMOS differential input circuit are coupled to a drain of a third PMOS serving as a second current source transistor, a current flowing through the third NMOS serving as the first current source transistor in the NMOS differential input circuit and a current flowing through the third PMOS serving as the second current source transistor in the PMOS differential input circuit are set respectively by the bias circuit,wherein the CMOS output circuit includes: an output PMOS operable to pull up an output voltage of the output terminal in response to a first output signal from at least one of the first NMOS and the second NMOS in the NMOS differential input circuit; and an output NMOS operable to pull down an output voltage of the output terminal in response to a second output signal from at least one of the first PMOS and the second PMOS in the PMOS differential input circuit,wherein the semiconductor integrated circuit for RF communications further includes a reference voltage generator operable to generate a roughly-stabilized internal regulated power supply voltage from a power supply voltage,wherein the power supply voltage is supplied to the NMOS differential input circuit, the bias circuit, and the CMOS output circuit in the CMOS rail-to-rail amplifier constituting the first voltage follower, the second voltage follower, and the third voltage follower included in the AFC-control D/A converter, andwherein the internal regulated power supply voltage generated by the reference voltage generator is supplied to the PMOS differential input circuit in the CMOS rail-to-rail amplifier constituting at least the third voltage follower included in the AFC-control D/A converter.

2. The semiconductor integrated circuit for RF communications according to claim 1, wherein the internal regulated power supply voltage generated by the reference voltage generator is supplied to the first variable voltage divider in the AFC-control D/A converter as a reference voltage.

3. The semiconductor integrated circuit for RF communications according to claim 1, further comprising: a PLL circuit operable to serve as a frequency synthesizer by including: a phase comparator of which one input terminal is supplied with the reference frequency signal generated by the reference frequency oscillator; a charge pump circuit responsive to an output of the phase comparator; a low-pass filter responsive to an output of the charge pump circuit; an RF voltage controlled oscillator responsive to a control output voltage of the low-pass filter; and a divider coupled between an output terminal of the RF voltage controlled oscillator and another input terminal of the phase comparator; andan RF transmission voltage controlled oscillator operable to generate an RF transmission frequency signal for an RF transmission signal in RF communications, by using an RF oscillation output signal present in the output terminal of the RF voltage controlled oscillator in the PLL circuit,wherein the PLL circuit serving as the frequency synthesizer is a fractional PLL circuit of which average division ratio includes an integer and a fraction as a result of alteration of division ratio in the divider.

4. The semiconductor integrated circuit for RF communications according to claim 3, wherein the PLL circuit serving as the frequency synthesizer includes an intermediate frequency divider operable to generate an intermediate frequency signal by dividing the RF oscillation output signal generated by the RF voltage controlled oscillator,wherein the semiconductor integrated circuit for RF communications includes: a transmission mixer operable to create an intermediate frequency transmitting signal from the intermediate frequency signal generated by the intermediate frequency divider and a transmitting baseband signal; a transmission-system offset PLL circuit; and an RF divider operable to generate a dividing RF frequency signal by dividing the RF oscillation output signal generated by the RF voltage controlled oscillator,wherein the transmission-system offset PLL circuit includes: a phase comparator circuit of which one input terminal is supplied with the intermediate frequency transmitting signal generated by the transmission mixer; the RF-transmission voltage controlled oscillator responsive to an output of the phase comparator circuit; and a phase-controlled-feedback frequency down mixer, one input terminal of the phase-controlled-feedback frequency down mixer being supplied with the RF transmission frequency signal generated by the RF-transmission voltage controlled oscillator and another input terminal of the phase-controlled-feedback frequency down mixer being supplied with the dividing RF frequency signal generated by the RF divider, andwherein an output signal of the phase-controlled-feedback frequency down mixer is supplied to another input terminal of the phase comparator circuit.

5. The semiconductor integrated circuit for RF communications according to claim 4, wherein the RF-reception-signal analog signal processing subunit includes: a low-noise amplifier operable to amplify an RF reception signal; and a receive mixer operable to generate a receiving baseband signal by supply of an RF-amplified reception output signal from the low-noise amplifier,wherein the PLL circuit serving as the frequency synthesizer includes: a first divider operable to create an RF carrier signal to be supplied to the receive mixer, by dividing the RF oscillation output signal of the oscillation frequency generated by the RF voltage controlled oscillator; and a second divider operable to divide an output signal of the first divider,wherein, when the semiconductor integrated circuit for RF communications receives the RF reception signal in one of frequency bands of GSM 850 MHz and GSM 900 MHz, a receiving baseband signal is created by the receive mixer after frequency conversion of the RF reception signal in the one of frequency bands of GSM 850 MHz and GSM 900 MHz, by transmission of a dividing output signal generated by the first divider to the receive mixer as the RF carrier signal,wherein, when the semiconductor integrated circuit for RF communications receives the RF reception signal in one of frequency bands of DCS 1800 MHz and PCS 1900 MHz, a receiving baseband signal is created after frequency conversion of the RF reception signal in the one of frequency bands of DCS 1800 MHz and PCS 1900 MHz, by transmission of the RF oscillation output signal of the oscillation frequency generated by the RF voltage controlled oscillator to the receive mixer as the RF carrier signal,wherein, when the semiconductor integrated circuit for RF communications creates the RF transmission frequency signal in one of frequency bands of GSM 850 MHz and GSM 900 MHz, the intermediate frequency transmitting signal is created by the transmission mixer from the intermediate frequency signal and a transmitting baseband signal, a diving output signal of the second divider is transmitted, as the dividing RF frequency signal, to the other input terminal of the phase-controlled-feedback frequency down mixer in the transmission-system offset PLL circuit by the action of the first and second dividers as the RF divider, and the intermediate frequency transmitting signal is converted in frequency to the RF transmission frequency signal in the one of frequency bands of GSM 850 MHz and GSM 900 MHz in the transmission-system offset PLL circuit, andwherein, when the semiconductor integrated circuit for RF communications creates the RF transmission frequency signal in one of frequency bands of DCS 1800 MHz and PCS 1900 MHz, the intermediate frequency transmitting signal is created by the transmission mixer from the intermediate frequency signal and a transmitting baseband signal, a diving output signal of the first divider is transmitted, as the dividing RF frequency signal, to the other input terminal of the phase-controlled-feedback frequency down mixer in the transmission-system offset PLL circuit by the action of the first divider as the RF divider, and the intermediate frequency transmitting signal is converted in frequency to the RF transmission frequency signal in the one of frequency bands of DCS 1800 MHz and PCS 1900 MHz in the transmission-system offset PLL circuit.

6. The semiconductor integrated circuit for RF communications according to claim 4, wherein the semiconductor integrated circuit for RF communications is constituted by a polar loop system in order to support an EDGE system,wherein the transmission-system offset PLL circuit includes a phase loop for phase modulation in the polar loop system and an amplitude loop in the polar loop system, andwherein the phase comparator circuit, the RF-transmission voltage controlled oscillator, and the phase-controlled-feedback frequency down mixer in the transmission-system offset PLL circuit constitute the phase loop.

7. The semiconductor integrated circuit for RF communications according to claim 4, wherein the semiconductor integrated circuit for RF communications includes a polar modulator system in order to support an EDGE system,wherein the transmission-system offset PLL circuit includes a phase loop for phase modulation in the polar modulator system and an amplitude loop in the polar modulator system, andwherein the phase comparator circuit, the RF-transmission voltage controlled oscillator, and the phase-controlled-feedback frequency down mixer in the transmission-system offset PLL circuit constitute the phase loop.

8. The semiconductor integrated circuit for RF communications according to claim 4, wherein the RF-reception-signal analog signal processing subunit includes: a low-noise amplifier operable to amplify an RF reception signal; and a receive mixer operable to generate a receiving baseband signal by supply of an RF-amplified reception output signal generated by the low-noise amplifier and a receiving carrier signal generated by the frequency synthesizer, andwherein the RF-transmission-signal analog signal processing subunit includes a transmission mixer operable to be supplied with the transmitting baseband signal, and the RF-transmission-signal analog signal processing subunit creates an RF transmission signal by supply of a transmitting carrier signal generated by the frequency synthesizer.

9. The semiconductor integrated circuit for RF communications according to claim 4, wherein the fractional PLL circuit includes a ΣΔ modulator for calculation of the fraction of the average division ratio.