Sensorless field oriented controller for two-phase motor

Abstract

A power converter circuit for providing maximum utilization of a DC bus voltage to a two-phase Permanent Magnet Synchronous Motor (PMSM) is disclosed. The circuit includes first, second, and third nodes, each node being the junction between series connected high and low side switches connected across a DC bus; a PMSM having first and second windings and a star point at which the first and second windings are coupled to each other, the first winding having a terminal connected to the first node, the second winding having a terminal connected to the second node, and the star point being connected to the third node; and a controller for performing a three-point Pulse Width Modulation (PWM) coupled to a gate of each switch.

Description

BACKGROUND OF THE INVENTION

The present invention relates to two-phase Permanent Magnet Synchronous Motors (PMSM), and more particularly to a strategy that provides maximum utilization of a DC bus voltage by employing a three-point Pulse Width Modulation (PWM).

Motor drives are used in a vast range of applications such as fans, pumps, compressors, washing machines, and etc. Such applications require motor drives to have high efficiency, low noise, and robustly stable operation. A two-phase Permanent Magnet Synchronous Motor (PMSM) has been recently introduced by motor manufacturers as an alternative to a well established but more expensive three-phase PMSM.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a strategy to control the two-phase PMSM.

In accordance with the present invention, a power converter circuit for providing maximum utilization of a DC bus voltage to a two-phase Permanent Magnet Synchronous Motor (PMSM) is disclosed. The circuit includes first, second, and third nodes, each node being the junction between series connected high and low side switches connected across a DC bus; the circuit being adapted to be coupled to a PMSM having first and second windings and a star point at which the first and second windings are coupled to each other, the first winding having a terminal connected to the first node, the second winding having a terminal connected to the second node, and the star point being connected to the third node; and a controller for performing a three-point Pulse Width Modulation (PWM) coupled to a gate of each switch.

The present invention improves over the prior art in that it provides PWM control of the PMSM neutral point, which yields a better utilization of the DC bus voltage than a simple connection of the neutral point to a capacitor split DC bus voltage.

Other features and advantages of the present invention will become apparent from the following description of the invention that refers to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a two-phase synchronous motor drive controlled by a three-point Pulse Width Modulation according to the invention;

FIG. 2 is a block diagram of a sensorless field oriented controller for a two-phase motor in a three-point modulation;

FIG. 3 is a block diagram of the speed observer of the sensorless field oriented controller;

FIG. 4 is a graph of simulated waveforms of average drive voltages for a DC bus voltage of 100V, winding voltage of 63V peak and control described by a v_star calculation;

FIG. 5 is a diagram defining an angle of an operating point;

FIG. 6 is a graph showing gating pulses for low switches and two-phase Space Vector PWM gating pulses that are active high; and

FIG. 7 shows the prior art circuit.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

FIG. 1 illustrates a circuit 10 that includes a two-phase non-salient PMSM 18 controlled by a three-point PWM. The circuit 10 further includes a U winding switching node 12 having a U winding of the motor 18 coupled between the in series connected high and low side switches 12H and 12L; a V winding switching node 14 having a V winding of the motor 18 coupled between the in series connected high and low side switches 14H and 14L; and an S switching node 16 having a star point 22 interconnecting the U and V windings of the two-phase motor 18 coupled between the in series connected high and low side switches 16H and 16L. The nodes 12, 14, and 16 and their high and low side switches are controlled by a controller 20. The circuit 10 is powered by a DC bus voltage supply.

FIG. 2 illustrates the sensorless field oriented controller 20 that is based on the Field Oriented Control (FOC) principle described in “Power Electronics and Variable Frequency Drives”, IEEE Computer Society Press, October 1996, by Bimal Bose (“Bose”). The controller 20 includes an outer speed loop and two inner current loops. The speed of the motor 18 is regulated to be equal to a command value provided by the action of an outer speed loop 24. One inner current loop 26 regulates a quadrature component of the motor current i_q and the other inner current loop 28 regulates a direct component of the motor current i_d. The motor current i_d is the equivalent motor stator current projected onto the d-axis, which is aligned with the rotor field, and i_q is the equivalent motor stator current projected on the q-axis, which is perpendicular to the rotor field (see Bose).

A model of the PMSM motor in d-q coordinates is given by the following equations as described in “Mathworks: SimPowerSystems Library” for Simulink, which may be found on the Internet at mathworks.com: $\begin{array}{cc}\frac{d}{dt}{i}_d=\frac{1}{L_d}{v}_d-\frac{R}{L_d}{i}_d+\frac{L_q}{L_d}p{\omega }_r{i}_q& \left(1\right)\\ \frac{d}{dt}{i}_d=\frac{1}{L_q}{v}_q-\frac{R}{L_q}{i}_q-\frac{L_d}{L_q}p{\omega }_r{i}_d-\frac{\lambda p{\omega }_r}{L_q}& \left(2\right)\\ T_e=1.5p\left[\lambda i_q+\left(L_d-L_q\right)i_d{i}_q\right]& \left(3\right)\\ \frac{d}{dt}{\omega }_r=\frac{1}{J}\left(T_e-F{\omega }_r-T_m\right)& \left(4\right)\\ \frac{d\theta }{dt}={\omega }_r& \left(5\right)\end{array}$ where

• L_q, L_d q and d axis inductances
• R Resistance of the stator windings
• i_q, i_d q and d axis currents
• v_q, v_d q and d axis voltages
• ω_r Angular velocity of the rotor
• λ Amplitude of the flux induced by the permanent magnets of the rotor in the stator phases
• p Number of pole pairs
• T_e Electromagnetic torque
• J Combined inertia of rotor and load
• F Combined viscous friction of rotor and load
• θ_r Rotor angular position
• T_m Shaft mechanical torque

The inner current loops 26 and 28 of the controller 20 calculate values of motor winding voltages v_v and v_u (39) with respect to the star point 22 (FIG. 1) of the motor 18, averaged on one switching period of the power converter circuit 10 (FIG. 1). The inner current loops determine quadrature component of the motor current i_q and the direct component of the motor current i_d as follows. A phase current reconstruction section 30 receives the six gate pulse signals 41 provided to the node 12, 14, and 16 switches and the DC bus voltage supply current and provides the U and V winding current values i_u and i_v to section 32. Section 32 performs a Clarke transformation on the i_u and i_v current values, as follows: $\left[\begin{array}{c}{i}_{\alpha }\\ i_{\beta }\end{array}\right]=\left[\begin{array}{cc}1& 0\\ 0& 1\end{array}\right]\left[\begin{array}{c}{i}_u\\ i_v\end{array}\right]$ outputting i_α and i_β to section 34, where the Park transformation is performed as follows: $\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right]=\left[\begin{array}{cc}\cos \left({\theta }_r\right)& \sin \left({\theta }_r\right)\\ -\sin \left({\theta }_r\right)& \cos \left({\theta }_r\right)\end{array}\right]\left[\begin{array}{c}{i}_{\alpha }\\ i_{\beta }\end{array}\right]$ producing the i_q and i_d values. These values are then transformed into voltages v_q and v_d and provided to section 36 to be reverse Park transformed as follows: $\left[\begin{array}{c}{v}_{\alpha }\\ v_{\beta }\end{array}\right]=\left[\begin{array}{cc}\cos \left({\theta }_r\right)& -\sin \left({\theta }_r\right)\\ \sin \left({\theta }_r\right)& \cos \left({\theta }_r\right)\end{array}\right]\left[\begin{array}{c}{v}_d\\ v_q\end{array}\right]$ providing voltage values v_α and v_β to section 38, where the reverse Clarke transformation $\left[\begin{array}{c}{v}_u\\ v_v\end{array}\right]=\left[\begin{array}{cc}1& 0\\ 0& 1\end{array}\right]·\left[\begin{array}{c}{v}_{\alpha }\\ v_{\beta }\end{array}\right]$ provides the voltages v_v and v_u to the PWM section 40 for generating the gate signals for driving the switches controlling the U and V windings of the PMSM18.

Additionally, a speed observer section 42 of the outer speed loop receives transformation currents i_α and i_β from section 32 and transformation voltages v_α and v_β from section 36 and outputs an angular velocity of a rotor ω_r as a reference in the setting of the speed and an angular position of the rotor θ_r to the Park and reverse Park transformation sections 34 and 36. Details of the speed observer section 42 are illustrated in FIG. 3. One example of the speed observer section 42 is described in U.S. Pat. No. 6,910,389.

The inventive control strategy also controls an average voltage v_star on the switching period of the motor's star point 22 with respect to the GND. Compared to motor time constants, the node 12, 14, and 16 switches in the power converter circuit 10 are switched at a high frequency and the average of converter's 22 output voltage is regulated to be equal to the commanded values of v_v, v_u, and v_star.

Voltages applied across the windings U and V differ by π/2 in phase angle. In a conventionally utilized control strategy for a two-phase motor, the voltage of the motor's star point 22 is fixed at mid DC bus voltage by a capacitor divider. In this configuration, neglecting voltage drops across the wires and power switches, the maximum instantaneous voltage that can be generated across the motor winding is equal to a half of the DC bus voltage.

If the voltage of the star point is not fixed, but instead modulated at an appropriate phase angle with respect to the winding voltages, the maximum instantaneous voltage which can be generated across the motor winding increases. The optimum modulation function for the star point voltage, in terms of utilization of the DC bus voltage, is found to be the following:v_star=0.5·V_{DC bus}−0.5·(v_v+v_u) This calculation of the v_star voltage is performed in section 44, which receives its input from voltages v_α and v_β from section 36.

As described above, the controller 20 performs pulse width modulation (PWM) control of all three points 12, 14, and 16 driving a two-phase motor 18, i.e., both phases U and V as well as the neutral point 22 of the motor. Proper PWM control of the motor's neutral point 22 yields a better utilization of the DC bus voltage than a simple connection of the neutral point to a capacitor split DC bus voltage. The prior art circuit is shown in FIG. 7.

FIG. 4 illustrates simulated waveforms of average drive voltages for a DC bus voltage of 100V, winding voltage of 63V peak, and control as described above including control of v_star by the above-presented formula. The illustrated voltages are for K=0.3 and κ_k=45. All voltages are averaged on the switching period. In this Figure the available winding voltage exceeds half of the DC bus voltage by 26%. Therefore the utilization of the DC bus voltage is increased by 26% compared to the conventionally utilized control strategy.

FIG. 5 illustrates the implementation of a two-phase Space Vector PWM for the inventive control strategy for a two-phase PMSM. In the two-phase Space Vector PWM only two out of three legs in the power converter are switched during each given switching period. The third leg does not switch but instead is left in the initial position. This is beneficial because switching losses reduce by a third. Depending on the angle of the operating point, as shown in FIG. 5, there are 6 sectors of operation.

The controller 20 generates gating pulses according to a pattern displayed in Table 1.

TABLE 1
Pattern for generation of gating pulses for the proposed control strategy and two-phase SVPWM
Sectors	1	2	3	4	5	6
Angle value	angle < π/4	π/4 < angle < π/2	π/2 < angle < π	π < angle < 5π/4	5π/4 < angle < 3π/2	3π/2 < angle < 2π
$\frac{T_a}{0.5·T_s}$	(Dus − Dvs)	(Dvs − Dus)	Dvs	−Dvs	−Dus	Dus
$\frac{T_b}{0.5·T_s}$	Dvs	Dus	−Dus	(Dvs − Dus)	(Dus − Dvs)	−Dvs
Gating pattern	Phase 1 = u	Phase 1 = v	Phase 1 = v	Phase 1 = s	Phase 1 = s	Phase 1 = u
	Phase 2 = v	Phase 2 = u	Phase 2 = s	Phase 2 = v	Phase 2 = u	Phase 2 = s
	Phase 3 = s	Phase 3 = s	Phase 3 = u	Phase 3 = u	Phase 3 = v	Phase 3 = v

In Table 1, D_us and D_vs are defined as: $D_{\mathrm{us}}=\frac{v_u}{V_{\mathrm{DC\_bus}}}\mathrm{and}{D}_{\mathrm{vs}}=\frac{v_v}{V_{\mathrm{DC\_bus}}}$

Finally, FIG. 6 illustrates gating pulses used for the low switches of the power converter circuit 10 of the present invention. It further shows two-phase Space Vector PWM where the gating pulses are active high.

Although the present invention has been described in relation to particular embodiments thereof, many other variations and modifications and other uses will become apparent to those skilled in the art. It is preferred, therefore, that the present invention not be limited by the specific disclosure herein.

Claims

1. A circuit for driving a two-phase Permanent Magnet Synchronous Motor (PMSM), the circuit comprising: first, second, and third nodes, each node being a junction between series connected high and low side switches connected across a DC bus; the circuit being adapted to be coupled to a PMSM having first and second windings and a star point at which the windings are coupled to each other, the first winding having a terminal connected to the first node, the second winding having a terminal connected to the second node, and the star point being connected to the third node; and a controller for performing a three-point Pulse Width Modulation (PWM) coupled to a gate of each switch, wherein one of the points is the star point.

2. The circuit of claim 1, wherein the controller operates on the Field Oriented Control (FOC) principle to regulate a speed of the PMSM and includes an outer speed loop and first and second inner current loops.

3. The circuit of claim 2, wherein the outer speed loop sets the desired speed of the PMSM and the first inner current loop regulates a quadrature component iq of the motor current and the second inner current loop regulates a direct component id of the motor current, the motor current id is the equivalent motor stator current projected onto a d-axis aligned with a rotor field, and the motor current iq is the equivalent motor stator current projected on a q-axis, which is perpendicular to the rotor field.

4. The circuit of claim 2, wherein the first and second inner current loops calculate values of the first and second motor winding voltages vv and vu with respect to the star point of the motor averaged on one switching period of the circuit.

5. The circuit of claim 3, wherein the inner current loops determine quadrature component iq of the motor current and the direct component id of the motor current by (a) receiving gate pulse signals from each switch and the DC bus voltage supply current, (b) determining iu and iv current values of the first and second winding, (c) performing the Clarke transformation on the current values iu and iv, in accordance with [iαiβ]=[1001]⁡[iuiv] to determine current values iα and iβ, (d) performing the Park transformation on the determined current values iα and iβ, in accordance with [idiq]=[cos⁡(θr)sin⁡(θr)-sin⁡(θr)cos⁡(θr)]⁡[iαiβ], to determine current values iq and id, (e) converting the current values iq and id into corresponding voltage values vq and vd, (f) performing the reverse Park transformation on the voltage values vq and vd, in accordance with [vαvβ]=[cos⁡(θr)-sin⁡(θr)sin⁡(θr)cos⁡(θr)]⁡[vdvq], to determine voltage values vα and vβ, and (g) performing the reverse Clarke transformation on the voltage values vα and vβ, in accordance with [vuvv]=[1001]·[vαvβ], to determine voltages vv and vu required to drive the PMSM at the desired speed sets by the outer control loop.

6. The circuit of claim 5, wherein the determined voltages vv and vu are used to generate the three-point PWM for driving the first and second windings.

7. The circuit of claim 5, wherein an average voltage vstar on the switching period of the star point with respect to the GND is regulated to be equal to the set values of vv, vu, and vstar.

8. The circuit of claim 6, wherein the voltages applied across the first and second windings differ by π/2 in phase angle.

9. The circuit of claim 6, wherein the voltage of the star point is modulated at a phase angle with respect to the voltages of the first and second windings, thereby increasing a maximum instantaneous voltage that can be generated across the first and second windings.

10. The circuit of claim 6, wherein, the modulation of the star point voltage is determined as vstar=0.5·VDC bus−0.5·(vv+vu).

11. The circuit of claim 1, further including a speed observer circuit having an input of voltages vα and vβ and currents iα and iβ and outputting an angular position of a rotor θr and angular velocity of the rotor ωr.

12. A method of driving a two-phase Permanent Magnet Synchronous Motor (PMSM), the method comprising the steps of: providing a circuit having a controller for performing a three-point Pulse Width Modulation (PWM), for driving a PMSM having first and second windings and a star point at which the windings are coupled to each other, and first, second, and third nodes, each node being a junction between series connected high and low side switches connected across a DC bus, the controller having a connection to a gate of each switch; connecting the first winding to the first node, the second winding to the second node, and the star point to the third node; and controlling the voltage applied to the PMSM, wherein the voltage of the star point is controlled along with voltages of the first and second windings.