Patent Application
20250102550
This application is based on Japanese Patent Application No. 2023-166003 filed on Sep. 27, 2023, the disclosure of which is incorporated herein by reference.
The present disclosure relates to a shunt resistance measurement circuit for a shunt-based current sensor.
Electric vehicles (EVs), hybrid electric vehicles (HEVs) and other electrically powered vehicles are equipped with a battery. To estimate the remaining charge of this type of battery, a circuit may be required to measure the current flowing through the battery. The shunt-based current sensor may be provided to measure the current flowing through the battery. In the shunt-based current sensor, a shunt resistor is connected to the battery in series. The shunt-based current sensor may measure the current by measuring a voltage across the shunt resistor.
The present disclosure describes a shunt resistance measurement circuit for a shunt-based current sensor, and further describes that the shunt resistance measurement circuit includes a control circuit, a switching power supply circuit, a voltage measurement circuit, and a signal processing circuit.
A shunt resistor may change its resistance value over time due to degradation. Therefore, by accurately determining the resistance value of the shunt resistor and then correcting the value, the current may be measured with high accuracy. According to a shunt-based current sensor in a comparative example, a highly accurate reference excitation current is supplied to a shunt resistor, the voltage across the shunt resistor is measured. The shunt resistance value is calculated from the supplied excitation current and the measured voltage across the shunt resistor. Therefore, the shunt resistance value is corrected to improve accuracy.
When applying the technique in the comparative example, a reference current source that generates a highly accurate reference current is required. A reference current source with small temperature characteristics and small deterioration may not be desirable because the respective physical sizes of a reference diode, a transistor, and a correction circuit may increase. Thus, an increased cost may be anticipated. Furthermore, the power supply power for passing the excitation current tends to become larger, which may be undesirable.
A shunt resistance measurement circuit according to an aspect of the present disclosure is adapted to a shunt-based current sensor configured to measure a resistance value of a shunt resistor. The shunt resistance measurement circuit includes a control circuit, a switching power supply circuit, a voltage measurement circuit, and a signal processing circuit. The control circuit outputs a control signal including an AC carrier signal. The switching power supply circuit is operated by the control signal to output a power supply to the shunt resistor. The voltage measurement circuit measures a voltage across the shunt resistor. The signal processing circuit measures the resistance value of the shunt resistor by executing signal processing, based on the voltage measured by the voltage measurement circuit.
According to the above configuration, it is possible to measure the resistance value of the shunt resistor based on the AC excitation current generated by the switching power supply circuit. Therefore, there is no need to provide another current supply source for passing the current with higher accuracy.
The following describes several embodiments related to shunt resistance measurement circuits. In each of the embodiments described below, the same or similar reference numerals are attached to the same or similar configuration, and the description is omitted as necessary.
The following describes a first embodiment with reference to
The MCU 57 includes a processor and a memory, and operates based on a program preliminarily stored in the memory. The memory is a non-transitory tangible storage medium. The current excitation circuit 52 includes, for example, a switching power supply circuit 11. The current excitation circuit 52 supplies an alternating current (AC) excitation current IE to the shunt resistor 12. The analog front end 50a is connected between the terminals of the shunt resistor 12. The analog front end 50a measures the voltage across the shunt resistor 12, and performs analog processing. The shunt resistor 12 is connected in series with a battery 58. The shunt resistor 12 is provided to measure a load current IL flowing through the battery 58 with high accuracy.
The battery 58 is a main battery and/or an auxiliary battery in which battery cells are combined in series or in series-parallel. The battery 58 is provided to drive a load 80 such as a main motor of a vehicle such as an electric vehicle or a hybrid vehicle, or a DC-DC converter, and/or to supply power to a vehicle electronic control device provided within the vehicle.
A relay 59 and the load 80 are connected in series to the battery 58 and the shunt resistor 12. When the ignition switch is turned on by a user to start the vehicle, the relay 59 is turned on. When the relay 59 is turned on, the load 80, the battery 58, and the shunt resistor 12 are connected. As a result, the load current IL flows through the shunt resistor 12.
The shunt resistor 12 has a resistance in the order of several tens of microohms. When the load current IL is supplied to the battery 58, a current in the order of several thousand amperes flows through the shunt resistor 12. The resistance value RS of the shunt resistor 12 changes due to deterioration over time. Therefore, the shunt resistance measurement circuit 1 can accurately measure the load current IL flowing through the battery 58 by accurately determining the resistance value RS of the shunt resistor 12 and then correcting for the influence of deterioration over time.
In addition, a temperature sensor 54 is disposed in the vicinity of the shunt resistor 12. The temperature sensor 54 is connected to the analog front end 53 for temperature measurement. The analog front end 53 for temperature measurement processes the temperature measurement signal TS provided from the temperature sensor 54, removes noise by waveform shaping, and outputs the temperature measurement signal TS to the DSP 51.
The DSP 51 corrects the load current IL flowing through the shunt resistor 12 and the resistance value RS of the shunt resistor 12 in accordance with the temperature measurement signal TS, and outputs these values to the MCU 57. The DSP 51 functions as a signal processing circuit 14 shown in
On the other hand, an integrated circuit 55 is connected to the battery 58 for monitoring the battery cells. The integrated circuit 55 monitors the voltage VC of each battery cell of the battery 58 and measures the remaining battery charge (SOC). The temperature sensor 56 is also located near the battery 58 and connected to the integrated circuit 55. When the integrated circuit 55 receives a temperature measurement signal TCELL from the temperature sensor 56, the integrated circuit 55 corrects the battery cell voltage VC in accordance with the temperature measurement signal TCELL and outputs the corrected voltage VC to the MCU 57. This enables the MCU 57 to monitor the state of each battery cell of the battery 58. The integrated circuit 55 may transmit a temperature measurement signal TCELL to the MCU 57, and the MCU 57 may correct the voltage VC of the battery cell in response to the temperature measurement signal TCELL.
The MCU 57 receives information related to the load current IL and the resistance value RS of the shunt resistor 12 from the DSP 51. Therefore, the MCU 57 can accurately measure information related to the load current IL flowing through the battery cells included in the battery 58 according to the resistance value RS.
The following describes the configuration block of the shunt resistance measurement circuit 1 with reference to
The carrier signal SC is a square wave signal having a frequency fC of several hundred kHz, for example, about 100 kHz. Hereinafter, the reciprocal of the frequency fC is represented as a period TC. The modulation signal SM is a signal having a frequency fM lower than that of the carrier signal SC, and is, for example, a square wave signal of about several hundred Hz. Hereinafter, the reciprocal of the frequency fM is represented as the period TM. The mixer 15 multiplies the carrier signal SC and the modulation signal SM to generate a modulated carrier signal SMC, and outputs the modulated carrier signal SMC to the switching power supply circuit 11.
The modulated carrier signal SMC is used as a signal for driving the switching power supply circuit 11. The modulated carrier signal SMC is a signal in which a relatively high frequency carrier signal SC is superimposed on a relatively low frequency modulation signal SM. In the present application, signals having the above relationship will be described as carrier signal SC and modulation signal SM, but the names of carrier signal SC and modulation signal SM are not limited to these names.
The switching power supply circuit 11 shown in
The switching power supply circuit 11 illustrated in
Furthermore, the switching power supply circuit 11 includes a charge sampling switching element M1 connected in series between the DC power supply node Ni and the upstream switching element M2. The switching power supply circuit 11 further includes a capacitor C1 disposed in a charge sampling stage between an intermediate node Nb and a ground node Ng. The intermediate node Nb is located between the upstream switching element M2 and the charge-sampling switching element M1.
The drive signal SM1 drives the gate of the switching element M1, which is the P-channel MOSFET. The drive signal SM3 drives the gate of the switching element M3, which is the N-channel MOSFET. Therefore, the switching elements M1 and M3 are turned on or off simultaneously. Furthermore, the driver circuit 11a outputs the drive signal SM2 that is at a low voltage only for a section of the time during which the drive signals SM1 and SM3 are OFF drive signals. The OFF drive signals may also be referred to as drive signals at an off level. The drive signal SM2 is applied to the gate of the switching element M2, which is the P-channel MOSFET.
When the switching elements M1 and M3 are in the on-state, the switching element M2 is maintained in the off-state. In this case, the capacitor C1 is charged from the node Ni through the switching element M1. Conversely, when the switching elements M1 and M3 are in the off-state, the switching element M2 is in the on-state only for a section of the time.
The switching power supply circuit 11 outputs the AC excitation current IEfrom the capacitor C1 through the inductor L1 only for this section of the time. As a result, the AC excitation current IE has a current waveform in which a small amplitude carrier signal SC is superimposed on a large amplitude waveform of the modulation signal SM with the frequency fm, as shown in
The switching power supply circuit 11 includes a charge sampling stage including a switching element M1 and the capacitor C1. This allows a large current to be supplied from the capacitor C1 of the charge sampling stage, and the switching power supply circuit 11 can be configured even if its power supply capacity is reduced. As a result, the switching power supply circuit 11 can be constructed at lower cost.
Furthermore, since a large current during operation of the switching power supply circuit 11 can be supplied from the capacitor C1 of the charge sampling stage, fluctuation in the power output of the switching power supply circuit 11 can be suppressed. As a result, interference with other circuits can be suppressed, and the resistance value RS of the shunt resistor 12 can be measured with high accuracy.
As shown in
When the timing signal is provided to the first voltage measurement circuit 13, the first voltage measurement circuit 13 samples and holds the measured voltage, and outputs the measured voltage to the signal processing circuit 14. This allows the control circuit 10 to perform control so as to synchronize the power output of the switching power supply circuit 11 with the measurement timing of the voltage measurement circuits 13 and 13a. In addition, the first voltage measurement circuit 13 can measure the voltage applied to the shunt resistor 12 at a timing synchronized with the power output of the switching power supply circuit 11, and can obtain the multiplied value of the resistance value RS of the shunt resistor 12 and the AC excitation current IE.
As illustrated in
The modulated carrier signal SMC is, for example, a signal in which a relatively high frequency carrier signal SC is superimposed on a relatively low frequency modulation signal SM. For this reason, it is sufficient to adjust the component values of the switching power supply circuit 11, such as the inductor L1 and the capacitor C1, and various characteristics of the switching elements M1 to M3, to match the higher frequency carrier signal SC compared to the modulation signal SM. If the switching elements M1 to M3 are selected to accommodate low frequencies, the component values of the inductor L1 and the capacitor C1 will increase, resulting in higher costs. Therefore, by driving the switching power supply circuit 11 with the modulated carrier signal SMC, as in the present embodiment, it is possible to easily generate the AC excitation current IE with effective cost.
According to the present embodiment, the modulation signal SM, which is the multiplication of the carrier signal SC output by the control circuit 10 and the modulation signal SM, is applied to the switching power supply circuit 11 to drive the switching power supply circuit 11. The power supply, including the AC excitation current IE of the switching power supply circuit 11, is applied to the shunt resistor 12. Then, the signal processing circuit 14 calculates the resistance value RS of the shunt resistor 12. As a result, the resistance value RS of the shunt resistor 12 can be measured based on the AC excitation current IE generated by the switching power supply circuit 11.
Furthermore, the power supply V0 is a step-down power supply from the battery 58, and the switching power supply circuit 11 can generate the AC excitation current IE using power supply V0. Therefore, it can be designed with lower power consumption and lower cost. Without the need for additional power circuits such as unnecessary regulators, it can be designed with lower power consumption and lower cost by generating the AC excitation current IE using the switching power supply circuit 11.
According to the present embodiment, it is possible to perform AC excitation at a relative low frequency of the modulation signal SM which is acquired by the carrier signal SC, and minimize the influence of the skin effect of the shunt resistor 12. Thus, it is possible to reduce measurement errors based on the influence of the skin effect of the shunt resistor 12 in order to enable accurate measurement of the resistance value RS of the shunt resistor 12. According to the present embodiment, by generating the AC excitation current IE using a low-frequency modulation signal SM, the respective values of inductor L1 and capacitor C1 can be reduced. As a result, the structure related to the present embodiment can be designed with lower cost.
The following describes a second embodiment with reference to
The reference resistor 12a and the shunt resistor 12 are connected in series between a power supply output node No and a ground node Ng of the switching power supply circuit 11. The reference resistor 12a is a highly accurate resistor used as a measurement reference, and its resistance value is typically 2 to 3 to 6 digits multiplied by the resistance value of the shunt resistor 12, for example, around 10 ohms. As mentioned later, the resistance value RS of the shunt resistor 12 is measured by taking the ratio of the shunt resistor 12 to the reference resistor 12a. For this reason, it would be easier to make measurements if similar resistance values were selected. However, generally available high-precision, low-degradation reference resistor 12a has a resistance value of 10 ohms or higher. Therefore, the reference resistor 12a and the shunt resistor 12 are constructed using resistors with significantly different resistance values, RREF1 and RS, respectively.
The second voltage measurement circuit 13a measures the voltage across the reference resistor 12a. The control circuit 10 controls the power output of the switching power supply circuit 11 and the measurement timing of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a to be synchronized by the timing generation circuit 10a. Furthermore, the timing at which the second voltage measurement circuit 13a measures the voltage across the reference resistor 12a is synchronized with the timing at which the first voltage measurement circuit 13 measures the voltage across the shunt resistor 12. Since the first voltage measurement circuit 13 and the second voltage measurement circuit 13a are synchronized based on the sample base, there is no need for synchronization between the signal processing circuit 14 and the control circuit 10. As a result, it is possible to reduce wiring and other synchronization circuits.
The signal processing circuit 14 processes the signal based on the respective voltages measured by the first voltage measurement circuit 13 and the second voltage measurement circuit 13a, and measures the resistance value RS of the shunt resistor 12. The other configurations are the same as those of the first embodiment.
According to such a configuration, since the reference resistor 12a is used as the measurement reference, a highly accurate reference current generation circuit is not necessary. Therefore, the configuration can be made at lower cost. The measurement timing of the first voltage measurement circuit 13 and the measurement timing of the second voltage measurement circuit 13a are synchronized with the power output of the switching power supply circuit 11. Therefore, the signal processing circuit 14 can perform highly accurate synchronous demodulation and effectively remove noise other than the main component of the AC excitation current IE (for example, the fundamental frequency of the modulation signal SM).
The first voltage measurement circuit 13 includes an AD converter 18 that measures the voltage across the shunt resistor 12 by the AC excitation current IE and converts the voltage measurement result of the shunt resistor 12 into a digital signal. The second voltage measurement circuit 13a includes an AD converter 18a that measures the voltage across the reference resistor 12a by the AC excitation current IE and converts the voltage measurement result of the reference resistor 12a into a digital signal.
The signal processing circuit 14 executes the signal processing at the same frequency as the respective frequencies of the digital output of the AD converter 18 of the first voltage measurement circuit 13 and the digital output of the AD converter 18aof the second voltage measurement circuit 13a for synchronous demodulation at the same frequency as the frequency of the AC excitation current IE induced by the modulation signal SM.
As illustrated in
When the mixers 21i and 21q receive the measurement result of the first voltage measurement circuit 13, they mix (multiply) the sample synchronization signals of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a as signal data that are out of phase with each other by 90°, and output the result to the filters 31i and 31q, respectively. The filters 31i and 31q are low-pass filters that restrict the bandwidth of the respective input signal data, and output the signal data that has passed through the low-frequency range to the absolute value calculator 41.
The outputs of filters 31i and 31q represent the real component Re and imaginary component Im, respectively. The absolute value calculator 41 calculates the square root of the sum of squares of the real component Re and imaginary component Im, and outputs the calculated result as the output of the synchronous demodulation circuit 14a. This result is output to the divider 43. The synchronous demodulation circuit 14a outputs the measurement result of the shunt resistor 12, calculated as the product of the AC excitation current IE and the resistance value RS of the shunt resistor 12, to the divider 43. The product of the AC excitation current IE and the resistance value RS of the shunt resistor 12 can also be expressed as “IE×RS”.
On the other hand, mixers 22i and 22q receive the measurement results from the second voltage measurement circuit 13a, and mix (multiply) the measurement results with respective sample synchronous signals of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a, which have a phase difference of 90 degrees. Thus, signal data are produced with different phases. The signal data are then output to filters 32i and 32q, respectively. The filters 32i and 32q are low-pass filters that restrict the bandwidth of the respective input signal data, and output the signal data that have passed through the low-frequency range to the absolute value calculator 42.
The respective output voltages of filters 32i and 32q represent the real component Re and imaginary component Im. The absolute value calculator 42 calculates the square root of the sum of squares of the real component Re and imaginary component Im, and outputs the calculated result as the output of the synchronous demodulation circuit 14a. This result is output to the divider 43. The synchronous demodulation circuit 14a outputs the measurement result of the reference resistor 12a, calculated as the product of the AC excitation current IE and the resistance value RREF1 of the reference resistor 12a, to the divider 43. The product of the AC excitation current IE and the resistance value RREF1 of the shunt resistor 12 can also be expressed as “IE×RREF1”.
The signal processing circuit 14 performs calculation using the divider 43 and the multiplier 44 based on the two output results of the synchronous demodulation circuit 14a. The signal processing circuit 14 measures the resistance value RS of the shunt resistor 12 by dividing the measurement result IE×RS of the shunt resistor 12 by the measurement result IE×RREF1 of the reference resistor 12a using the divider 43, and then multiplying the obtained quotient with the reference resistance value RREF1 of the reference resistor 12a, which is pre-stored value, using the multiplier 44.
The reference resistance value RREF1 of the reference resistor 12a is measured preliminarily during manufacture or inspection of the reference resistor 12a, and is stored preliminarily in a non-volatile memory (not shown) of the signal processing circuit 14. Therefore, the signal processing circuit 14 can eliminate the influence of the AC excitation current IE by executing the above-mentioned predetermined calculation using the divider 43 and the multiplier 44. Therefore, the influence of noise other than the main component contained in the AC excitation current IE, such as disturbances and low-frequency noise (1/f noise, offset, etc.) of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a, can be eliminated. As a result, the resistance value RS of the shunt resistor 12 can be measured with higher accuracy.
The following describes a third embodiment with reference to
According to the configuration of the shunt resistance measurement circuit 301, the impedance (the resistance value RREF) of the reference resistor 12a, to which the AC excitation current IE is supplied, is small at several tens of ohms. In addition, the impedance (the resistance value RS) of the shunt resistor 12 is small at several tens of microohms. Therefore, as compared with a so-called conventional diode rectification method by using the switching element M3 that meets the condition of lower on-resistance, the switching power supply circuit 11 can output power with relatively low power consumption.
Additionally, by providing a charge sampling stage using the switching element M1 and the capacitor C1, it is possible to supply a large current from the capacitor C1 of the charge sampling stage, in order to lower the power supply capacity of the switching power supply circuit 11. As a result, the switching power supply circuit 11 can be constructed at lower cost.
Furthermore, since a large current during operation of the switching power supply circuit 11 can be supplied from the capacitor C1 of the charge sampling stage, fluctuation in the power supply output of the switching power supply circuit 11 can be suppressed. As a result, interference with other circuits can be suppressed, and the resistance value of the shunt resistor 12 can be measured with higher accuracy.
The following describes a fourth embodiment with reference to
Further, the shunt resistance measurement circuit 401 includes a DC excitation circuit 16 as a power supply different from the switching power supply circuit 11. The shunt resistance measurement circuit 401 also includes a multiplexer 19. The DC excitation circuit 16 is provided with a regulator capable of turning on and off the output under the control of the control circuit 10, and outputs a DC current IE1. The first reference resistor 12a, the second reference resistor 12b, and the shunt resistor 12 are connected in series between the DC excitation circuit 16 and the ground node Ng.
The multiplexer 19 is arranged among the first reference resistor 12a, the second reference resistor 12b, the shunt resistor 12, the first voltage measurement circuit 13 and the second voltage measurement circuit 13a. The multiplexer 19 is capable of switching between input and output under the control of the control circuit 10. In the present embodiment, the series circuit of the second reference resistor 12b, the first reference resistor 12a, and the shunt resistor 12 is constructed so that power can be supplied from the DC excitation circuit 16.
The material of the first reference resistor 12a and the second reference resistor 12b may include tantalum nitride, titanium nitride, or nitride. By using material with minimal characteristic changes and minimal durability degradation due to temperature variation, it is possible to construct the first reference resistor 12a and the second reference resistor 12b, in order to enable high-precision measurement of the resistance value RS of the shunt resistor 12.
The signal processing circuit 14 measures the resistance value RS of the shunt resistor 12 in a first measurement process and a second measurement process.
In the first measurement process, the control circuit 10 stops the operation of the switching power supply circuit 11 by turning off all of the switching elements M1 to M3 included in the switching power supply circuit 11. Then, the control circuit 10 controls the current IE1 of the DC excitation circuit 16 to be in an on-state. Furthermore, the control circuit 10 switches the input and output of the multiplexer 19, allowing the second voltage measurement circuit 13a to measure the voltage across the first reference resistor 12a, and the first voltage measurement circuit 13 to measure the voltage across the second reference resistor 12b. The first measurement process may also be referred to as “step 1” in, for example,
In the first measurement process, the signal processing circuit 14 processes the measurement results obtained by the second voltage measurement circuit 13a and the first voltage measurement circuit 13 for the first reference resistor 12a and the second reference resistor 12b, respectively. Based on these measurement results, the signal processing circuit measures the resistance value RRREF1 of the first reference resistor 12a and the resistance value RRREF2 of the second reference resistor 12b.
In the first measurement process, the signal processing circuit 14 measures the resistance ratio between the resistance value RREF1 of the first reference resistor 12a and the resistance value RREF2 of the second reference resistor 12b using a DC measurement path 14c shown in
The filter 45 passes the low frequency components of the voltage measured by the first voltage measurement circuit 13 and cuts out high frequency noise exceeding the DC component. The filter 46 passes the low frequency components of the voltage measured by the second voltage measurement circuit 13a and cuts out high frequency noise exceeding the DC component.
The adder/subtractor 47 subtracts a pre-stored offset from the DC component of the voltage measured by the first voltage measurement circuit 13. The adder/subtractor 48 subtracts a pre-stored offset from the DC component of the voltage measured by the second voltage measurement circuit 13a. This offset is added or subtracted in order to minimize the influence of low-frequency noise (1/f noise, offset) of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a.
The divider 49 divides the output of the adder/subtractor 48 by the output of the adder/subtractor 47. As a result, the divider 49 can measure the result RREF2/RREF1 acquired by dividing the resistance value RREF2 of the second reference resistor 12b by the resistance value RREF1 of the first reference resistor 12a, while eliminating the influence of the current IE1 of the DC excitation circuit 16. The output of the divider 49 is output to a multiplier 44a.
On the other hand, in the second measurement process, the signal processing circuit 14 processes the measurement voltage across the second reference resistor 12b and the measurement voltage across the shunt resistor 12, and measures the resistance ratio between the resistance value RREF2 of the second reference resistor 12b and the resistance value RS of the shunt resistor 12 based on these measurement voltages. In the second measurement process, the signal processing circuit 14 may measure the resistance ratio using an AC measurement path 14b.
The AC measurement path 14b includes the synchronous demodulation circuit 14a and the divider 43. In the second measurement process, the control circuit 10 controls the switching power supply circuit 11 to output the AC excitation current IE, and controls the current IE1 of the DC excitation circuit 16 to turn off the DC excitation circuit 16. The control circuit 10 switches the input and output of the multiplexer 19 to cause the second voltage measurement circuit 13a to measure the voltage across the second reference resistor 12b and the first voltage measurement circuit 13 to measure the voltage across the shunt resistor 12.
When the synchronous demodulation is performed based on the operation described in the preceding embodiment, the synchronous demodulation circuit 14a outputs a value, which is the product of the AC excitation current IE and the resistance value RREF2 of the second reference resistor 12b (in other words, IE×RREF2), to the divider 43, and also outputs a value, which is the product of the AC excitation current IE and the resistance value RS of the shunt resistor 12 (in other words, IE×RS), to the divider 43. The divider 43 can calculate the resistance ratio RS/RREF2 between the resistance value RS of the shunt resistor 12 and the resistance value RREF2 of the second reference resistor 12b. In addition, the divider 43 can adjust the resistance ratio RS/RREF2 to a practical and manageable value by multiplying the resistance ratio RS/RREF2 with a predetermined gain adjustment (GainCal).
The signal processing circuit 14 includes the multiplier 44a provided after the DC measurement path 14c and the AC measurement path 14b. The multiplier 44a calculates the resistance ratio RS/RREF1 by multiplying the resistance ratio RREF2/RREF1 between the resistance values RREF1 and RREF2 measured in the first measurement process, and the resistance ratio RS/RREF2 between the resistance values RREF2 and RS measured in the second measurement process. Moreover, the multiplier 44a can measure the resistance value RS of the shunt resistor 12 by multiplying the resistance ratio RS/RREF1 by the reference resistance value RREF1 that is preliminarily stored in the memory. In this manner, the resistance value RS of the shunt resistor 12 can be measured.
As described above, the relationship between the resistance value RREF1 and the resistance value RREF2, and the relationship between the resistance value RREF2 and the resistance value RS may be set to a difference of one to two digits, or two to three digits, for example, about 10:1 to 100:1, or about 100:1 to 1000:1. The first voltage measurement circuit 13 and the second voltage measurement circuit 13a selectively measure the respective resistance values RREF1, RREF2, and RS of the first reference resistor 12a, the second reference resistor 12b, and the shunt resistor 12. The signal processing circuit 14 measures the resistance value RS of the shunt resistor 12 using the voltage ratio and resistance ratio of the terminal voltages of two of the resistance values RREF1, RREF2, and RS.
With this configuration, the first voltage measurement circuit 13 and the second voltage measurement circuit 13a can suppress the difference in terminal voltage measured by each circuit to, for example, around −60 dB. Even when the resistance value RS of the shunt resistor 12 is measured based on the resistance ratio as described above, it is possible to perform measurement while minimizing the influence of noise.
As mentioned earlier, the resistance value RREF1 of the first reference resistor 12a is set to a high-precision resistance of 10 ohms, while the resistance value RS of the shunt resistor 12, which is the measurement target, is significantly different, with a magnitude of several tens of microohms and approximately six orders of magnitude apart from the resistance value RREF1. However, by dividing the resistance value RS of the shunt resistor 12 into two measurement processes as described in the above-mentioned first measurement process and second measurement process, it becomes possible to compensate for and reduce the influence of various noises that occur in the first voltage measurement circuit 13 and the second voltage measurement circuit 13a, in order to enable high-precision measurement of the resistance value RS of the shunt resistor 12.
According to the present embodiment, the three resistors can be switched by the multiplexer 19, and the voltages can be measured by the two voltage measurement circuits 13, 13a, so that the configuration can be made at lower cost. When the signal processing circuit 14 executes the first measurement process, the operation of the switching power supply circuit 11 used in the second measurement process is stopped. This is because by eliminating the influence of the output impedance of the switching power supply circuit 11, it is possible to reduce the measurement error of the resistance value RS of the shunt resistor 12.
If the configuration is adopted, as in the present embodiment, where the output of the switching power supply circuit 11 is directly connected to the first reference resistor 12a, by turning off all the switching elements M1 to M3 of the switching power supply circuit 11, it is possible to maintain a high output impedance of the switching power supply circuit 11. As a result, the resistance value RS of the shunt resistor 12 can be measured while minimizing the influence of the switching power supply circuit 11. An additional switch may be provided between the output node No of the switching power supply circuit 11 and the first reference resistor 12a.
In the above example, the resistance value RS is measured in two separate measurement processes. However, the present disclosure is not limited to this example, and the resistance value RS may be measured in three or more separate measurement processes. In other words, not limited to just the first reference resistor 12a and the second reference resistor 12b, the configuration includes connecting other reference resistors, such as the third reference resistor, in series with the shunt resistor 12. By measuring the voltage across these reference resistors in pairs and calculating their resistance ratios, the resistance value RS of the shunt resistor 12 can be measured by multiplying the calculated ratios. This detailed description is omitted.
The following describes a fifth embodiment with reference to
The AC excitation circuit 516 may generate the AC excitation current IE1 by connecting the configuration of the control circuit 10 described in the preceding embodiment. The AC excitation circuit 516 may include a linear regulator (LDO) 516a to which the modulation signal SM is applied. At this time, the linear regulator 516a provides the input DC voltage VIN to a switching element M4, and toggles the switching element M4 with the modulation signal SM provided from the control circuit 10, thereby generating the AC excitation current IE1.
The AC excitation circuit 516 drives the switching element M4 by the modulation signal SM output from the control circuit 10. The use of such a linear regulator 516a allows for lower degree of implementation of the power supply capability of the AC excitation circuit 516, in order to be constructed with low-cost components. The present embodiment has the same advantages as those of the preceding embodiments.
Moreover, the shunt resistance measurement circuit 501 includes the signal processing circuit 514 instead of the signal processing circuit 14. As shown in
If the AC excitation circuit 516 supplies the AC excitation current IE1, the synchronous demodulation process of the signal processing circuit 514 can be shared between the first measurement process and the second measurement process. In this case, the excitation circuits for the first measurement process and the second measurement process, such as the control circuit 10 that generates the modulation signal SM, can be shared and implemented at lower cost.
The signal processing circuit 514 executes measurements through the synchronous demodulation circuit 14a of the AC measurement path 14b in both the first measurement process and the second measurement process. In addition to the second measurement process, by also performing synchronous demodulation in the first measurement process, it is possible to accurately measure and remove noise, such as disturbances, low-frequency noise (1/f noise, offset, etc.) in the first voltage measurement circuit 13 and the second voltage measurement circuit 13a, other than the main component of the AC excitation current IE1.
The signal processing circuit 514 includes dividers 43a,43b, and a multiplier 44a. In the first measurement process, the divider 43a divides the output value IE1×RREF2 of the absolute value calculator 41 by the output value IE1×RREF1 of the absolute value calculator 42, and multiplies the calculated result by the gain. Thus, it is possible divide the resistance value RREF2 of the second reference resistor 12b by the resistance value RREF1 of the first reference resistor 12a. The divider 43a outputs the result of this division to the multiplier 44a. In the second measurement process, the signal processing circuit 514 can divide the resistance value RS of the shunt resistor 12 by the resistance value RREF2 of the second reference resistor 12b by the divider 43b dividing the output value IE×RS of the absolute value calculator 41 by the output value IE×RREF2 of the absolute value calculator 42 and multiplying the calculated result by the gain. The divider 43b outputs the result of this division to the multiplier 44a.
The signal processing circuit 514 multiplies the measurement result acquired by the first measurement process with the measurement result acquired by the second measurement process using the multiplier 44a, and measures the resistance value RS of the shunt resistor 12 by multiplying the calculated result with the pre-stored reference resistance value RREF1 of the first reference resistor 12a. Since one of the first voltage measurement circuit 13 and the second voltage measurement circuit 13a is divided by the output of the other, noise contained in the AC excitation currents IE and IE1 can be removed, and the resistance value RS of the shunt resistor 12 can be measured with higher accuracy. The elements of the signal processing circuit 514 used in the first measurement process and the second measurement process (for example, the multiplier 44a) can be shared, and can be implemented at lower cost.
The following describes a sixth embodiment with reference to
As illustrated in
The first voltage measurement circuit 13 includes a differential switched capacitor amplifier 17 and an AD converter 18 including a delta-sigma (AZ) modulator in the downstream stage. The first voltage measurement circuit 13 is included in the analog front end 50a. The second voltage measurement circuit 13a includes a differential switched capacitor amplifier 17a and the AD converter 18a including the A2 modulator in the downstream stage. The second voltage measurement circuit 13a is included in the analog front end 50a.
As illustrated in
The ASIC 50 illustrated in
Outside the ASIC 50, anti-aliasing filters 20b and 20 are provided. The anti-aliasing filter 20b is arranged between the second reference resistor 12b and the multiplexer 19, and cuts frequency components higher than the Nyquist frequency before the measurement voltage of the second reference resistor 12b is provided to the AD converters 18, 18a. The anti-aliasing filter 20 is arranged between the shunt resistor 12 and the multiplexer 19, and cuts frequency components higher than the Nyquist frequency before the measurement voltage of the shunt resistor 12 is provided to the AD converters 18, 18a.
The current excitation circuit 52 includes the switching power supply circuit 11 as the AC excitation circuit, and also includes the DC excitation circuit 16. In addition, in
The following describes a seventh embodiment with reference to
The shunt resistance measurement circuit 701 according to the present embodiment includes a control circuit 710, the switching power supply circuit 11, a first voltage measurement circuit 13, and a signal processing circuit. The control circuit 710 outputs an AC carrier signal SC. The switching power supply circuit 11 includes the switching elements M1 to M3, drives the switching elements M1 to M3 based on a signal including a carrier signal SC, and outputs a power supply including an AC excitation current IE. The output of the switching power supply circuit 11 is supplied to the shunt resistor 12.
The first voltage measurement circuit 13 measures the voltage across the shunt resistor 12. The signal processing circuit 14 processes the signal based on the voltage measured by the first voltage measurement circuit 13 and measures the resistance value RS of the shunt resistor 12. The other configuration is the same as that of the preceding embodiment, and hence the description will be omitted.
When setting the carrier signal SC of the shunt resistance measurement circuit 701 to lower the frequency to a level where the skin effect of shunt resistor 12 has minimal impact, the respective component values of the inductor L1 and the capacitor C1 are more likely to be larger than the typical values used in a regular operation.
In this case, while adjusting the size of the inductor L1 and capacitor C1 included in the switching power supply circuit 11, the switching elements M1 to M3 can be driven by the carrier signal SC to allow the switching power supply circuit 11 to output an AC excitation current IE at the frequency fC of the carrier signal SC.
As shown in
measurement circuit 701, the switching power supply circuit 11 is driven by the carrier signal SC. The other configurations are similar to those of the preceding embodiments (in particular, the fourth embodiment), and therefore will not be described.
According to this embodiment, since the signal is excited by the carrier signal SC and processed, it is not necessary to generate an excitation current having the frequency of the modulation signal SM, and therefore lower power consumption is possible. Since the modulation circuit (for example, the mixer 15) is not necessary, the configuration area can be reduced.
The present disclosure should not be limited to the embodiments described above, and various modifications may further be implemented without departing from the spirit of the present disclosure. For example, the following modifications or extensions can be performed. Each component is conceptual and is not limited to the above-described embodiments.
The control device such as the control circuit 10 and method described in the present disclosure may be implemented by a special purpose computer which is configured with a memory and a processor programmed to execute one or more particular functions embodied in computer programs of the memory. Alternatively, the control device described in the present disclosure and the method thereof may be realized by a dedicated computer configured as a processor with one or more dedicated hardware logic circuits. Alternatively, the control device and method described in the present disclosure may be realized by one or more dedicated computer, which is configured as a combination of a processor and a memory, which are programmed to perform one or more functions, and a processor which is configured with one or more hardware logic circuits. In addition, the computer program may be stored in a computer-readable non-transition tangible recording medium as an instruction to be executed by a computer.
Although the present disclosure has been described in accordance with the embodiments, it is understood that the present disclosure is not limited such embodiments or structures described in the embodiments. To the contrary, the present disclosure is intended to cover various modification and equivalent arrangements. Furthermore, various combination and formation, and other combination and formation including one, more than one or less than one element may be made in the present disclosure.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2023-166003 | Sep 2023 | JP | national |
