The present invention relates to optical communications networks, and in particular to signal acquisition in a coherent optical receiver.
In the optical communications space, receivers based on coherent detection techniques have suffered disadvantages that have, to date, prevented successful deployment in “real-world” installed communications networks.
One such limitation is that both the transmitted carrier signal and the receiver's local oscillator (LO) signal are generated by respective transmitter and LO lasers, which, in the case of “real world” network systems, will be compact fiber or semi-conductor lasers which are subject to manufacturing and environmental variations. Such lasers are typically designed such that the average output frequency (over a period of 100 s of milliseconds or more) is stable at a value which is nominally fixed by the frequency setting. However, short period frequency excursions due to laser line width and phase noise are permitted. As a result, frequency variations of as much as ±400 MHz, at rates on the order of up to 50KHz are commonly encountered. The resulting frequency mismatch Δf between the LO signal and the received carrier signal appears as a phase error in recovered symbols, which can lead to erroneous data detection.
In prior art coherent receiver systems, this problem is typically addressed by implementing an optical frequency locked loop (FLL) or Phase locked loop (PLL) to actively control the receiver's LO to match the received carrier signal. FLL and PLL circuits for this purpose are described in: “High Capacity Coherent Lightwave Systems”, Linke et al, Journal of Lightwave Technology, Vol. 6, No. 11, November 1988; “Heterodyne Phase Locked Loop by Confocal Fabry-Perot Cavity Coupled AlGaAs lasers”, Shin et al, IEEE Photonics Technology Letters, Vol. 2, No. 4, April 1990; and “Carrier Synchronization for 3 and 4-bit-per-Symbol Optical Transmission”, Ip et al, Journal of Lightwave Technology, Vol. 23, No. 12, December 2005. All of these systems operate to drive the receiver's LO to precisely track excursions of the received optical carrier. A limitation of this approach is that for optical communications systems with multi-gigabit line rates, a PLL/FLL loop bandwidth on the order of hundreds of MHz is needed to effectively compensate the laser phase noise. This is difficult to achieve at acceptable cost.
An alternative approach is to use an electrical carrier recovery circuit for detecting and compensating the frequency mismatch between the LO and received carrier. A carrier recovery circuit designed for this purpose is described in “Phase Noise-Tolerant Synchronous QPSK/BPSK Baseband-Type Intradyne Receiver Concept With Feedforward Carrier Recovery”, R Noé, Journal of Lightwave Technology, Vol. 23, No. 2, February 2005. A limitation of electrical carrier compensation in this manner is that it can only feasibly compensate some aspects of moderate frequency errors. As a result, a large frequency transient can cause severe performance degradations, for example due to limited analog amplifier bandwidth, and clock recovery issues.
A further limitation of coherent detection systems is that they are highly vulnerable to optical impairments of the received carrier signal. In particular, optical signals received through conventional optical links are distorted by significant amounts of chromatic dispersion (CD) and polarization dependent impairments such as Polarization Mode Dispersion (PMD), polarization angle changes and polarization dependent loss (PDL). Chromatic dispersion (CD) on the order of 30,000 ps/nm, and polarization rotation transients at rates of 105Hz are commonly encountered.
Various methods of compensating Polarization angle are known in the art. See, for example, “Phase Noise-Tolerant Synchronous QPSK/BPSK Baseband-Type Intradyne Receiver Concept With Feedforward Carrier Recovery”, R Noé, Journal of Lightwave Technology, Vol. 23, No. 2, February 2005, and “PLL-Free Synchronous QPSK Polarization Multipex/Diversity Receiver Concept with Digital I&Q Baseband Processing”, R Noé, IEEE Photonics Technology Letters, Vol. 17, No. 4, April 2005. In this respect, it will be noted that Noé also alludes (in the introduction) to the possibility of also compensating chromatic dispersion. However, Noé does not provide any teaching as to how this would be done. The applicability of RF channel estimation techniques to the detection of polarization-division multiplexed optical signals in a quadrature coherent receiver is described by Y. Han et al. in “Coherent optical Communication Using Polarization Multiple-Input-Multiple-Output”, OPTICS EXPRESS Vol. 13, No. 19, pp 7527-7534, 19 Sep. 2005.
A limitation that is common throughout the prior art is a lack of satisfactory bandwidth of the various compensation functions. For example, the FLL/PLL and carrier recovery techniques described above are intended to track (and thus compensate) laser phase noise. However, in order to provide sufficient accuracy of compensation, they lack sufficient bandwidth to acquire a signal across the entire possible range of impairment magnitude, such as a frequency error of several GigaHertz. As a result, these systems cannot reliably acquire a signal and stabilize to steady-state operation, even if they could track laser phase transients after a steady state had been achieved. Similarly, the system of Noé [supra] is designed to compensate polarization rotations, but it cannot track high speed transients of the type encountered in real-world communications networks. For example, Noé, claims that with a 10 GBaud signal, the inverse Jones matrix coefficients can be updated with a period of 16 μs. This is far too slow to successfully compensate 20 kHz polarization rotations, which have a period of 50 μs. In addition, the system of Noé tends to fail in the presence of severe Chromatic Dispersion (CD), at least in part due to failure of the clock recovery circuit as inter-symbol interference (ISI) increases, and consequent uncertainty of the sample timing of the A/D converters. While it is mathematically possible to design a filter function that compensates both polarization and chromatic dispersion (as alluded to by Noé), the prior art does not offer any methods by which satisfactory compensation accuracy can be obtained with an adaptation speed high enough to track real-world polarization transients. It follows that the system of Noé will not be able to reliably capture the instantaneous polarization state of the received signal during start-up, especially in the presence of high speed transients, and thus cannot guarantee that it will achieve a stable steady-state operation.
Prior art clock recovery systems suffer the same limitation, in that the PLL bandwidth required to obtain a satisfactory sample phase accuracy is significantly less than the possible range of clock and channel errors. As a result, conventional clock recovery circuits cannot reliably acquire a lock condition, even if they are able to maintain lock once it has been achieved. A further limitation of clock recovery circuits is that they are very vulnerable to distortions in the received optical signal. While this can be overcome by compensating at least some of the distortions prior to the clock recovery circuit, such compensation normally requires the recovered clock signal in order to operate. As a result, the receiver cannot reliably acquire signal and achieve a steady state operation, even if such a state can be maintained once it has been achieved.
Accordingly, methods and techniques that enable a coherent optical receiver to reliably acquire signal and achieve steady-state operation remain highly desirable.
An object of the present invention is to provide methods and techniques that enable a coherent optical receiver to reliably acquire signal and achieve steady-state operation.
Thus, an aspect of the present invention provides a method of initializing a coherent optical receiver. An optical signal is detected. A multi-bit digital sample stream of the optical signal is then processed to initialize each one of a plurality of adaptive control blocks of the coherent optical receiver. The plurality of adaptive control blocks include at least a dispersion compensation block and a clock recovery block, of which the dispersion compensation block is initialized before initializing the clock recovery block.
Further features and advantages of the present invention will become apparent from the following detailed description, taken in combination with the appended drawings, in which:
a-3b are block diagrams schematically illustrating principal elements and operations of the LO control clock of the coherent optical receiver of
a-7c are block diagrams schematically illustrating respective dispersion compensation loops usable in an embodiment of the present invention;
It will be noted that throughout the appended drawings, like features are identified by like reference numerals.
The present invention provides methods and techniques that enable reliable signal acquisition and stabilization to steady-state operation of a coherent receiver unit of an optical communications network. Embodiments of the present invention are described below, by way of example only, with reference to
In general, the present invention provides methods in which a multi-bit sample stream of a received optical signal is digitally processed to find receiver parameters which compensate link impairments with sufficient accuracy that any residual distortions are within a pull-in range of “steady-state” adaptation loops of the receiver. As a result, signal acquisition and steady-state operation of the receiver can be achieved with a high degree of reliability, even in the presence of moderate to severely distorted optical signals.
As may be appreciated, the resolution of the A/D converters 14 is a balance between performance and cost. Increasing the resolution improves sampling accuracy, and thereby improves the extent to which signal distortions can be corrected by downstream dispersion and polarization compensators. However, this increased accuracy is obtained at a cost of increased complexity, silicon area and heat generation. It has been found that a resolution of 5 or 6 bits provides satisfactory performance, at an acceptable cost. Preferably, the sample rate of the A/D converters 14 is selected to satisfy the Nyquist criterion for the highest anticipated symbol rate of the received optical signal. As will be appreciated, Nyquist sampling ensures that the sample streams generated at the A/D converter output contains all of the information content of each signal, even if the sample timing (with reference to each received symbol) is ambiguous and/or unknown. From the A/D converter 14 block, the I and Q sample streams of each received polarization are supplied to a respective dispersion compensator 16, which operates on the sample stream(s) to compensate chromatic dispersion of the optical link.
In the embodiment of
As will be appreciated, the amount of dispersion that can be compensated will be a function of the width of the FFT/IFFT filters 18 and 20, which will be a balance between performance and cost. In some embodiments, each filter has a width of 256 samples, which enables compensation of well over 10000 ps/nm of dispersion. A dispersion training loop 22 for calculating the dispersion compensator coefficients (and thereby training the dispersion compensator) can be implemented using a variety of methods, as will be described in greater detail below.
A dispersion compensation block can be linear or nonlinear, or some combination of both. The FFT method described above is a particularly efficient linear implementation when used in a coherent optical receiver.
A clock recovery block 24 taps the dispersion compensators 16 to obtain an at least partially dispersion compensated sample stream 26, which is then used for recovering a clock signal, as will be described in greater detail below.
The dispersion-compensated sample streams 28 appearing at the output of the dispersion compensators 16 are then supplied to a polarization compensator 30 which operates to de-convolve the transmitted I and Q signal components of each transmitted polarization from the dispersion-compensated sample streams 28. Various methods may be used to implement the polarization compensators 30, such as, for example, a Finite Impulse Response (FIR) filter. A polarization training loop including a SYNC detector 32 and a coefficient calculator 34 compute updated filter coefficients which are then downloaded to the polarization compensator 30 to track and compensate polarization impairments, as will be described in greater detail below.
The distortion-compensated sample streams appearing at the output of each polarization compensator 30 are then supplied to a carrier recovery block 36 for compensating residual frequency mismatch Δf between the LO and the carrier of the received optical signal, as well as symbol detection (for data recovery).
As will be appreciated from the foregoing description, signal acquisition, and achievement of steady-state operation of the coherent optical receiver of
As may be seen in
Determination of the Presence of a Signal
The acquisition process described above with reference to
As will be appreciated, more sophisticated techniques may be employed to determine the presence of an optical signal by analysing optical characteristics of the received light.
LO Control
As mentioned above, the LO control loop 10 is used to control the frequency of the LO 6 to minimize frequency mismatch Δf between the LO and the carrier of the inbound optical signal. As is well known in the art, conventional optical PLL/FLL loops are capable of driving the frequency mismatch Δf to near zero, but lack sufficient loop bandwidth to be able to reliably acquire a signal across the entire possible frequency mismatch range. For example, conventional semiconductor lasers of the type commonly used in optical communications may exhibit frequency transients of as much as ±400 MHz from the nominal frequency, which implies a possible mismatch range of ±800 MHz. Laser line width and manufacturing variations can be expected to increase this. Modern tuneable lasers may exhibit a tuning range of as much as ±3600 MHz. Accordingly, the present invention implements an LO-scan operation, in which the frequency of the LO is varied (e.g. using a predetermined update rate and step size) until the frequency mismatch Δf falls within the pull-in range of the optical PLL/FLL loop.
a illustrates a laser control loop 10 in accordance with Applicant's co-pending U.S. patent application Ser. No. 11/279,042 filed Apr. 6, 2006. In the embodiment of
Clearly, computation of the frequency mismatch parameter ψ(n), and thus the frequency adjustment ΔF, requires that the dispersion compensators and polarization compensators have already acquired signal and achieved steady-state operation. Prior to this, the frequency adjustment ΔF can be held to a value of zero (or otherwise discarded), so that the default LO frequency setting fO governs the LO frequency setting fLO. As a result, the initial frequency mismatch will normally be limited to the known performance tolerances of the Tx and LO lasers.
Using the above LO control loop 10, an LO-scan operation can be readily implemented by iteratively adjusting the value of the “default” LO frequency setting fO. An alternative arrangement is to hold the default LO frequency setting fO constant, and iteratively adjust the frequency adjustment value ΔF, as may be seen in the embodiment of
If desired, the optimum LO frequency (as determined by any of the above methods) may be sent to a transmitter end of the optical link, for example as part of a hand-shake protocol used during System Layout and Test (SLAT) of the optical link.
As mentioned above, the LO control loop 10 of
In the embodiment of
Various methods may be used to implement the hand-shaking protocol.
When a valid data signal is detected, the signal is examined (at S18) to determine if it contains the first pilot signal P1. If the first pilot signal P1 is detected, then the line card enters a “master” mode of operation (at 520), and transmits (at S22) a second pilot signal P2 that is indicative of this state. Preferably, the second pilot signal P2 is selected to be polarization independent and highly robust to dispersion, because it is assumed that the remote node may still be attempting to detect a valid signal. The line card then resumes monitoring the incoming signal from the remote node.
If the received signal does not contain the first pilot signal P1, then the signal is examined (at S24) to determine if it contains the second pilot signal P2. If the second pilot signal P2 is detected, then it is known that the remote node has already entered the “master” mode of operation. Accordingly, the line card enters a “slave” mode (at 526), and transmits an acknowledgement signal P3 (at S28) that is indicative of this state. The line card then enables (at S30) carrier frequency tracking by the LO controller 10, so that the LO frequency can be continuously adjusted to minimize the frequency mismatch Δf between its LO and the carrier of the received optical signal (from the master node). The line card can then continue its start-up and signal acquisition sequence as may be required in order to enter steady state operation.
If the received signal does not contain either the first or second pilot signals P1 and P2, then the signal is examined (at S32) to determine if it contains the acknowledgement signal P3. If the acknowledgement signal P3 is detected, then it is known that the remote node has entered the “slave” mode of operation, and is therefore adjusting its LO frequency setting fLO to track changes in the local (master) line card's LO frequency. Accordingly, the line card disables carrier frequency tracking (at S34) to hold its LO frequency setting fLO constant, and then continues its start-up and signal acquisition sequence as may be required in order to enter steady state operation.
If none of above signals P1-P3 are detected in the received optical signal within a predetermined time-out period, then it may be assumed that the remote node cannot participate in the hand-shaking protocol. This may occur, for example, in cases where the remote node uses a conventional line card in which independent lasers are used for transmission and coherent reception. Accordingly, upon a time-out (at S36), the line card enables carrier frequency tracking (at S38), so that the LO frequency can be continuously adjusted to minimize the frequency mismatch Δf between its LO and the carrier of the received optical signal. The line card can then continue its start-up and signal acquisition sequence as may be required in order to enter steady state operation.
An advantage of the foregoing hand-shaking protocol is that it enables the line card to establish appropriate control of its LO frequency, and ensure stable operation of the optical link, independently of whether or not the remote node contains a similar line card. In fact, legacy line cards with conventional direct detection receivers can be used at the remote node, if desired, without adversely interrupting operation of the above hand-shaking protocol.
In the foregoing hand-shaking protocol, a set of pilot and acknowledgement signals are used to establish continuity and assert (and acknowledge) master/slave modes of operation. As noted above, these signals are transmitted “in-line” through the optical link. However, it will be appreciated that this is not necessary. For example, information concerning the successful detection of a signal, assertion of master/slave status etc. could be conveyed through a control channel, or via a system management network, as shown in
Dispersion Compensation
As mentioned above, the dispersion compensators 16 implement a fist order dispersive function which at least partially compensates chromatic dispersion of the optical link 2. A dispersion control loop 22 for calculating the dispersion compensator coefficients (and thereby training the dispersion compensators) can be implemented using a variety of methods.
In some embodiments, the total chromatic dispersion may be known from measurements taken during installation of the optical link 2, or estimated from known physical characteristics of the optical fibre and equipment provisioned within the optical link 2.
A still further alternative approach is illustrated in
Various known methods may be used to determine the residual dispersion within the “compensated” sample streams 28. For example, a correlation can be computed (at 62) between the “compensated” sample streams 28 and a known pilot signal within the received optical signal. If the known pilot signal is substantially independent of polarization, then the resulting correlation value will be indicative of residual dispersion. In some cases, the pilot signal may conveniently be provided as SYNC bursts comprising a predetermined sequence of SYNC symbols described in greater detail below. With this arrangement, computing a correlation between the “compensated” sample streams 28 appearing at the output of the dispersion compensators 16 and the known SYNC symbol sequence provides a direct indication of the residual dispersion. Other types of signals may be used as the pilot signal, as will be readily apparent to those skilled in the art.
As will be appreciated, if the total dispersion is sufficiently high, it will exceed the “pull-in range” of the dispersion control loop, in the sense that changing the candidate dispersion value will yield no change in the computed correlation, and thus no useful information to guide changes in the adjustment step size and direction. This can be overcome by implementing a “dispersion scan” function, in which the candidate dispersion is forced to scan through a desired range of possible values, until it falls within the pull-in range of the dispersion control loop.
If desired, the dispersion scan process can be designed to scan dispersion values spanning a range of ±50,000 ps/nm, or more, with a desired scan rate and step size. Other scan ranges, and any of a variety of search strategies may be implemented to optimize the dispersion scan operation, and thereby minimize the mean time to signal acquisition.
If desired, the dispersion scan operation may be combined with the LO frequency scan described above. Thus, for example, both of the scan operations can run simultaneously, and may use the same criteria to terminate both scan operations. This combined operation is based on the recognition that, for a polarization independent pilot signal, residual distortion at the dispersion compensator output will be a function of both total link dispersion and the frequency mismatch Δf. Since these parameters are orthogonal, a 2-Dimensional control surface can be defined which relates the total dispersion compensated by the dispersion compensators 16 and the LO frequency setting fLO to the pilot signal correlation value, for example. Signal acquisition then becomes a process of mapping the control surface to find optimum values of the dispersion compensation coefficients and LO frequency setting fLO. One way of doing this is by scanning both parameters at the same time until the computed correlation value reaches a local maximum. The respective scan rates and/or step sizes of each parameter are preferably different, which allow the effects of changes in each parameter on the residual dispersion to be distinguished, so that appropriate decisions regarding step size and direction for each parameter can be made.
If desired, the total dispersion (as determined by any of the above methods) may be sent to the remote node at opposite end of the optical link, for example as part of a hand-shake protocol used during System Layout and Test (SLAT) of the optical link.
Clock Recovery
As noted above, the clock recovery loop 24 taps the dispersion compensators 16 to obtain a (partially) dispersion compensated sample stream 26, which is then used for recovering a clock signal. During an initial phase of operation, a nominal clock can be used to drive the A/D converters 14, so as to generate sample streams having an appropriate sample rate, but with an indeterminate sample timing. As mentioned above, the sample rate is preferably selected to satisfy the Nyquist criterion for the expected symbol rate of the received optical signal. This provides valid multi-bit sample streams which can be input to the dispersion compensators 16 to initialize the dispersion compensation loop. Thus, for example, in an embodiment implementing the dispersion scan operation described above with reference to
Preferably, the clock recovery circuit 24 implements the methods described in Applicant's co-pending U.S. patent application Ser. Nos. 11/315,342 and 11/315,345 filed Dec. 23, 2005, the entire contents of both of which are hereby incorporated herein by reference.
In the embodiment of
The operational status of the clock recovery circuit 24 can be determined by means of a lock detection block 78 implementing respective broadband and narrow band filter functions for of the coarse and fine tuning loops. Thus, for example, the coarse tuning path can be used to tune the VCO 72 until a coarse lock indication is generated by an infinite impulse response (IIR) filter 80 of the lock detection block 78. When coarse lock is achieved, operation continues using the fine tuning loop to tune the VCO 72. A leaky-bucket filter 82 of the lock detection block 78 can conveniently be used to output a fine lock indication when a frequency/phase lock condition has been achieved.
In the embodiment of
In the embodiment of
The arrangement of
Polarization Compensation
In the embodiment of
In some embodiments, the polarization compensation training loop implements the methods described in Applicant's co-pending U.S. patent application Ser. No. 11/294,613 filed Dec. 6, 2005, the contents of which are hereby incorporated herein by reference. Thus, for example, the SYNC detector 32 may operate to compute a correlation between the dispersion compensated sample stream 28 emerging from the polarization compensators 16, and a known SYNC symbol sequence embedded within the received optical signal. Based on the resulting correlation, the coefficient calculator 34 can derive a set of filter coefficients which, when downloaded to the polarization compensator 30, operate to de-convolve the transmitted polarizations from the dispersion compensated sample streams 28.
Representative SYNC symbol sequences of the type which may be used for this purpose are described in Applicant's co-pending U.S. patent application Ser. No. 11/328,199 filed Jan. 10, 2006, the contents of which are hereby incorporated herein by reference.
As will be appreciated, the accuracy with which this can be accomplished will be largely dependent on the loop delay (or equivalently, the adaptation delay) of the training loop. In effect, the filter coefficients must be recalculated with sufficient speed and frequency to track changes in the polarization state of the received optical signal. For optical links in which maximum polarization transients of 2KHz or less are expected, a recalculation frequency as low as 10KHz may be sufficient. As the anticipated polarization transient rates increase, so too must the recalculation frequency of the filter coefficients. Thus, in some embodiments, recalculation rates in excess of 100KHz will be desired.
In the embodiment of
In some embodiments, the SYNC sequence may be repeated at a frequency of about 1000 times lower than the symbol rate of the optical communications signal. Thus, for an optical communications system in which the symbol rate is 10 GHz, the SYNC repetition frequency will be about 10 MHz. The small size of the polarization compensator 30 enabled by cascading the polarization compensator 30 downstream of the dispersion compensators 16 means that the SYNC detector 32 and coefficient calculator 34 can form a “high-speed” training loop capable of re-computing and downloading the filter coefficients during the interval between successive SYNC sequences. Such frequent coefficient updating facilitates near real-time tracking and compensation of polarization transients having rates well in excess of 50KHz.
In the embodiment of
In the illustrated embodiment, the distribution unit 90 is implemented as a “burst switch” controlled by a framer 92, to generate overlapping blocks of samples. One implementation of a burst switch may, for example, include a multi-port Random Access Memory (RAM), which allows samples to be simultaneously supplied to two or more data paths. This arrangement offers the advantage that each sample block contains sufficient information, in its respective SYNC burst(s), to enable polarization compensation and data decoding substantially independently of any other data path.
The framer 92 may, for example, use various correlation techniques to detect the timing of each SYNC burst within the dispersion compensated sample stream(s) 28. In one embodiment, techniques similar to those described above for clock recovery can be used to enable detection of SYNC bursts even in the presence of severe polarization impairments. As may be appreciated, during signal acquisition, it is not necessary for the framing operation to be perfect. In fact, it is sufficient that the framer 92 control the distribution unit 90 with sufficient accuracy that each sample block supplied to each data path contains an intact SYNC burst. When this condition is satisfied, the SYNC detector 32 and coefficient calculator 34 can successfully operate to provide compensation of the polarization impairments. Once the polarization compensators 30 have stabilized, the framer 92 can use the polarization compensated sample streams to accurately determine the SYNC burst timing.
Link phenomena such as dispersion, nonlinearities and polarization effects can be directly estimated or parameterized, or proxy parameters can be used. For example, dispersion can be directly parameterized as D picoseconds per nanometer, or other convenient units. Alternatively, general phase shift effects due to phenomena like dispersion can be parametrized in a manner that is convenient to the particular hardware or firmware implementation.
A nonlinear equalization block can compensate for hardware impairments and for some optical nonlinearities. This block benefits from initialization after major impairments, such as large amounts of dispersion, have been at least partially equalized.
Hard and or soft decision Forward Error Correction (FEC) blocks can be used to reduce the effects of noise.
These can be single blocks, multiple, or iterated along with multiple equalization blocks. A Soft FEC outputs multibit samples that generally have reduced noise, compared to the input samples. FEC blocks benefit from initialization, such as FEC frame acquisition, after impairments have been at least partially equalized by upstream stages within the coherent receiver.
The embodiments of the invention described above are intended to be illustrative only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.
This application claims benefit under 35 U.S.C. 119(e) from U.S. Provisional Patent Application Ser. No. 60/728,751, entitled Automatic Gain Control, which was filed on Oct. 21, 2005.
Number | Name | Date | Kind |
---|---|---|---|
4506388 | Monerie et al. | Mar 1985 | A |
4720827 | Kanaji | Jan 1988 | A |
4723316 | Glance | Feb 1988 | A |
4965858 | Naito et al. | Oct 1990 | A |
5457563 | Van Deventer | Oct 1995 | A |
5473463 | Van Deventer | Dec 1995 | A |
5546190 | Hill et al. | Aug 1996 | A |
5995512 | Pogue et al. | Nov 1999 | A |
6473222 | Hait et al. | Oct 2002 | B2 |
6607311 | Fishman et al. | Aug 2003 | B1 |
6782211 | Core | Aug 2004 | B1 |
20020012152 | Agazzi et al. | Jan 2002 | A1 |
20020181571 | Yamano et al. | Dec 2002 | A1 |
20020186435 | Shpantzer et al. | Dec 2002 | A1 |
20030063285 | Pering et al. | Apr 2003 | A1 |
20030123884 | Willner et al. | Jul 2003 | A1 |
20030175034 | Noe | Sep 2003 | A1 |
20040033004 | Welch et al. | Feb 2004 | A1 |
20040114939 | Taylor | Jun 2004 | A1 |
20050047802 | Jaynes et al. | Mar 2005 | A1 |
20050058456 | Yoo | Mar 2005 | A1 |
20050196176 | Sun et al. | Sep 2005 | A1 |
Number | Date | Country |
---|---|---|
1453239 | Sep 2004 | EP |
2214381 | Aug 1989 | GB |
WO 0060776 | Oct 2000 | WO |
WO 0227994 | Apr 2002 | WO |
Number | Date | Country | |
---|---|---|---|
60728751 | Oct 2005 | US |