Signal localization arrangement

Description

BACKGROUND OF THE INVENTION

The present invention relates to a signal source localization arrangement comprising a plurality of receivers having different positions, the signal source localization arrangement comprising delay estimation means for estimating a delay difference between the signals received by at least two receivers, and position determining means for determining from the delay difference a signal source location.

The present invention relates also to a delay estimation arrangement, a video communication system and a signal source localization method.

An arrangement according to the preamble is known from the article “Voice source localization for automatic camera pointing system in videoconferencing” by Hong Wang and Peter Chu in IEEE, ASSP workshop on applications of signal processing to audio and acoustics, 1997.

Signal localization arrangements are used in several applications. A first example of such applications is automatic camera pointing in video conferencing systems or in security systems. Another application is the determination of the position of a user of an audio system, in order to be able to optimize the reproduction of the audio at said position.

Signal localization arrangements using a plurality of receivers are often based on the determination of a delay difference between the signals at the outputs of the receivers. If the position of the receivers and a delay difference between the propagation paths between the source and the different receivers are known, the position of the source can be determined. If two receivers are used, it is possible to determine the direction with respect to the baseline between the receivers. If three receivers are used, it becomes possible to determine a position of the source in a 2-D plane. If more than 3 receivers, being not placed in a single plane, are used, it becomes possible to determine the position of a source in three dimensions.

In the prior art signal localization arrangements, the delay difference is determined by calculating a cross-correlation function between the signals received by the different receivers. The delay difference is then equal to the delay value in the cross-correlation function at which the highest correlation value occurs.

A problem with the prior art signal localization arrangement is that its operation depends heavily on the properties of the signal generated by the source. Especially voiced speech signals in a reverberant environment can disturb the operation. To reduce this large influence of the signal properties, a long averaging time has to be used in determining the cross-correlation function of the received signals.

SUMMARY OF THE INVENTION

The object of the present invention is to provide a signal localization arrangement in which the adverse influence of the signal properties has been reduced.

To achieve said purpose, the signal localization arrangement is characterized in that the signal source localization arrangement comprises impulse response determining means for determining a plurality of functions representing the impulse responses of the paths between the signal source and the receivers, and in that the delay estimation means are arranged for determining the delay difference from said functions.

A function representing the impulse response is a function that represents an important aspect of the impulse response, but it may differ substantially in other aspects from the real impulse response of the paths between signal sources and receivers.

By determining the delay difference from functions representing the impulse responses of the paths between the signal source and the receivers instead of from the received signals themselves, the influence of the properties of the signals on the determination of the delay difference is strongly reduced. Experiments have shown that the averaging time to be used in the determination of the delay difference can be strongly reduced.

Preferably, the delay difference is determined by calculating a cross correlation function of the functions representing the impulse responses.

An embodiment of the invention is characterized in that the impulse response determining means comprise adjustable filters for deriving filtered signals from the signals provided by the receivers, the signal source localization arrangement comprising combining means for deriving a combined signal from the filtered signals, in that the impulse response determining means comprises control means for controlling the adjustable filters in order to maximize a power measure of the combined signal, and in that the control means are arranged for limiting a combined power gain measure of the filtered audio signals to a predetermined value.

By combining a plurality of filtered signals and adjusting the filters for maximizing a power of the combined signal under the constraint of a limited combined power gain measure, it is obtained that the filters converge to a transfer function leading to filtered signals having a maximum degree of coherence before they are added. This means that the delay differences between the impulse responses of the adjustable filters correspond to the delay difference between the signals at the outputs of the receivers.

A further embodiment of the invention is characterized in that the control means comprise a plurality of further adjustable filters having a transfer function being the conjugate of the transfer function of the adjustable filters, said further adjustable filters being arranged for deriving from the combined audio signal filtered combined audio signals, and in that the control means are arranged for maximizing the power measure of the combined audio signal, and for restricting a combined power gain measure of the processed audio signals to a predetermined value by controlling the transfer functions of the adjustable filters and the further adjustable filters in order to minimize a difference measure between the input audio signals and the filtered combined audio signal corresponding to said input audio signals.

Experiments have shown that by using two sets of adjustable filters, the quality of the speech signal can be further enhanced. By minimizing a difference measure between the input audio signal and the corresponding filtered combined audio signal, it is obtained that a power measure of the combined audio signal is maximized under the constraint that per frequency component the sum of the power gains of the adjustable filters is equal to a predetermined constant. The correspondence between the two criteria mentioned above will be shown in the detailed description of the drawings by using a simplified example.

BRIEF DESCRIPTION OF THE DRAWING

The present invention will now be explained with reference to the drawings.

FIG. 1

shows a block diagram of a video communication system using a signal source localization means according to the invention.

FIG. 2

shows a general block diagram of impulse response determining means according to the present invention.

FIG. 3

shows a more detailed impulse response determining means according to the invention in which frequency domain adaptive and frequency domain programmable filters are used;

FIG. 4

, shows an embodiment of the normalization means

73

used in the arrangement according to FIG.

2

.

FIG. 5

shows an implementation of the frequency domain adaptive filters

62

,

66

and

68

used in

FIG. 3

;

FIG. 6

shows an implementation of the frequency domain programmable filters

44

,

46

and

50

used in

FIG. 3

;

FIG. 7

shows an implementation of the impulse response determining means according to the invention in which time domain adaptive filters and time domain programmable filters are used.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the video communication system according to

FIG. 1

a microphone

1

is connected to impulse response determining means

5

, and a microphone

2

is connected to impulse response determining means

6

. Additional microphones

3

and

4

can be added to corresponding impulse response determining means in order to be able to determine, besides the direction, the exact position of the signal source in two or three dimensions. The impulse response determining means

5

and

6

determine a function representing the impulse response from the signal source to the respective microphones.

The outputs of the impulse response determining means

5

and

6

, carrying the functions representing the impulse responses, are connected to inputs of correlation calculation means

7

.

If the impulse response determining means

5

and

6

operate in the frequency domain, they will provide Fourier transforms W

1

* and W

2

* of the estimates of the impulse responses h

1

and h

2

. The crosscorrelation function can now be easily determined by first determining the cross power spectrum according to

Φ

w

1

,w

2

(

f

)=

W

1

·W

2

* (1)

In (1) W

2

is the complex conjugate of the function W

2

* provided by the impulse response determining means. From the function Φw

1

,w

2

(f) the cross-correlation function ρw

1

,w

2

(k) can be determined by performing an IFFT on the function

In the impulse response determining means

5

and

6

operate in the time domain, the functions h

1

and h

2

will be provided by these impulse response determining means. In that case, the cross-correlation function ρw

1

,w

2

(k) can be calculated according to:

\begin{matrix} ρ_{w_{1} w_{2}} (k) = \sum_{n = 0}^{n = N} h_{1} (k) \cdot h_{2} (n - k); k = 0, 1, \dots, N - 1 ρ_{w_{1} w_{2}} (k) = \sum_{n = - k}^{N - 1} h_{1} (k) \cdot h_{2} (n - k); k = - (N - 1), \dots, - 1 & (2) \end{matrix}

In (2) N is the length of h

1

and h

2

.

After the function ρw

1

,w

2

(k) has been determined, the resolution of this function can be improved by upsampling the function ρw

1

,w

2

(k) by introducing samples with value zero between the non-zero samples, followed by an interpolation. Useful interpolation factors are in the range from 4 to 8.

The (upsampled and interpolated) cross correlation function is passed by the correlation function determining means

7

to delay difference calculating means

8

. These delay difference calculation means

8

determine the value of k having the largest correlation value. The corresponding delay difference Δ is then equal to k·T

s

in which T

s

is the sampling period.

The output of the delay calculating means

8

is connected to the input of position calculating means. If two receivers are used, the direction θ defined as the angle between the connection line between the receivers and the direction of the source can be calculated according to:

\begin{matrix} θ = \arccos (\frac{c \cdot k \cdot T_{s}}{d}) & (3) \end{matrix}

In (3) c is the propagation speed of the signal to be received and d is the distance between the receivers. A signal representing the angle θ is passed to camera positioning means

10

which rotates the position of the camera over an angle θ with respect to the baseline between the receivers

1

and

2

.

In the impulse response determining means according to

FIG. 2

, an output of a first receiver, being here a microphone

1

, is connected to a first input of the impulse response determining means

5

,

6

and an output of a second receiver, being here a microphone

2

, is connected to a second input of the impulse response determining means

5

,

6

.

If it is assumed that the microphones

4

and

6

receive a signal V

IN

via propagation paths with impulse responses H

1

and H

2

respectively, the output signal of microphone

1

is equal H

1

·V

IN

and the output signal of microphone

2

is equal to H

2

·V

IN

. The output of the microphone

1

is connected to an input of a filter

10

with transfer function W

1

, and the output of the microphone

2

is connected to an input of a filter

12

with transfer function W

2

. At the output of the filters

10

and

12

the processed signals V

P

and V

Q

are available. For these processed signals can be written:

V

P

=H

1

·W

1

·V

IN

(4)

and

V

Q

=H

2

·W

2

·V

IN

(5)

At the output of the combination means

18

the sum V

SUM

of the processed signals V

P

and V

Q

is available. This signal V

SUM

is equal to:

V

SUM

=(

H

1

·W

1

+H

2

·W

2

)

V

IN

(6)

The output of the adder

18

is connected to the input of two further adjustable filters

14

and

16

. The further adjustable filters

14

and

16

derive filtered combined signals from the combined signal using transfer functions W

1

* and W

2

*. The first filtered combined signal is equal to

V

FC1

=(

H

1

·W

1

+H

2

·W

2

)·

W

1

*·V

IN

(7)

and the second filtered combined signal is equal to:

V

FC2

=(

H

1

·W

1

+H

2

·W

2

)·

W

2

*·V

IN

(8)

A first difference measure between the first input audio signal and the first filtered combined audio signal is determined by a subtractor

24

. For the output signal of the subtractor

24

can be written:

V

DIFF1

={H

1

−(

H

1

·W

1

+H

2

·W

2

)·

W

1

*}·V

IN

(9)

A second difference measure between the second input audio signal and the second scaled combined audio signal is determined by a subtractor

26

. For the output signal of the subtractor

26

can be written:

V

DIFF2

={H

2

−(

H

1

·W

1

+H

2

·W

2

)·

W

2

*}·V

IN

(10)

The arrangement according to

FIG. 2

comprises a control element

20

for adjusting the coefficients of filter

10

and

14

to make the power of the output signal of V

DIFF1

of the subtractor

24

equal to 0. The arrangement further comprises a control element

22

for adjusting the coefficients of filter

12

and

16

to make the power of the output signal V

DIFF2

of the subtractor

26

equal to 0. In order to find the values for x and y to make both difference signals equal to 0, the following set of equations has to be solved:

(

H

1

·W

1

+H

2

·W

2

)·

W

1

*=H

1

(11)

(

H

1

·W

1

+H

2

·W

2

)·

W

2

*=H

2

(12)

Eliminating the term (H

1

·W

1

+H

2

·W

2

) from (11) and (12) by dividing (11) by (12) results in:

\begin{matrix} \frac{W_{1}^{*}}{W_{2}^{*}} = \frac{H_{1}}{H_{2}} \Rightarrow W_{1}^{*} = \frac{H_{1} \cdot W_{2}^{*}}{H_{2}} & (13) \end{matrix}

By conjugating the left-hand side and the right-hand side of (13) for W

1

can be written:

\begin{matrix} \frac{W_{1}^{}}{W_{2}^{}} = \frac{H_{1}^{*}}{H_{2}^{*}} \Rightarrow W_{1} = \frac{H_{1}^{*} \cdot W_{2}}{H_{2}^{*}} & (14) \end{matrix}

Substituting (14) into (12) gives the following expression:

\begin{matrix} (\frac{{&LeftBracketingBar; H_{1} &RightBracketingBar;}^{2} \cdot W_{2}}{H_{2}^{*}} + H_{2} \cdot W_{2}) \cdot W_{2}^{*} = H_{2} & (15) \end{matrix}

Rearranging (15) gives for |W

2

|

2

:

\begin{matrix} {&LeftBracketingBar; W_{2} &RightBracketingBar;}^{2} = \frac{{&LeftBracketingBar; H_{2} &RightBracketingBar;}^{2}}{{&LeftBracketingBar; H_{1} &RightBracketingBar;}^{2} + {&LeftBracketingBar; H_{2} &RightBracketingBar;}^{2}} & (16) \end{matrix}

For |W

1

|

2

can be found in the same way:

\begin{matrix} {&LeftBracketingBar; W_{1} &RightBracketingBar;}^{2} = \frac{{&LeftBracketingBar; H_{1} &RightBracketingBar;}^{2}}{{&LeftBracketingBar; H_{1} &RightBracketingBar;}^{2} + {&LeftBracketingBar; H_{2} &RightBracketingBar;}^{2}} & (17) \end{matrix}

From (16) and (17) it is clear that the value of |W

1

|

2

increases when |H

1

|

2

increases (or |H

2

|

2

decreases) and that the value of |W

2

|

2

increases when |H

2

|

2

increases (or |H

1

|

2

decreases). In such a way the strongest input signal is pronounced. This is of use to enhance a speech signal of a speaker over background noise and reverberant components of the speech signal without needing to know the frequency dependence of the paths from the speaker to the microphones as was needed in the prior art arrangement.

Below will be demonstrated that maximizing the power of the combined audio signal under the constraint that the sum of the power gains of the processing means is limited, results in the same values for |H

1

|

2

and |H

2

|

2

as making the output signals of the subtractors

24

and

26

equal to 0.

For the power measure P

SUM

of the combined audio signal V

SUM

can be written:

P

SUM

=V

SUM

2

=|H

1

·W

1

+H

2

·W

2

|

2

·V

IN

2

(18)

For the boundary condition that the sum of the power gains of the scaling means is limited to a constant value can be stated:

G

P

=|W

1

|

2

+|W

2

|

2

=1 (19)

Consequently, the term |H

1

·W

1

+H

2

·W

2

|

2

has to be maximized under the boundary condition |W

1

|

2

+|W

2

|

2

−1=0. This can be done by using the well-known Lagrange multiplier method. According to said method, the following expression has to be maximized:

(|

H

1

·W

1

+H

2

·W

2

|

2

+λ·(|

H

1

|

2

+|H

2

|

2

−1) (20)

Differentiating (20) with respect to Re{W

1

}, Jm{W

1

}, Re{W

2

} and Jm{W

1

} and setting the derivatives to zero gives four equations with four variables. By solving these equations, and calculating the values of |W

1

|

2

and |W

2

|

2

(16) and (17) are found. Consequently it is clear that controlling W

1

and W

2

to make the difference signals equal to 0 is equivalent to maximizing the power of the combined signal under the boundary condition that the sum of the power gains of the different branches of the processing means is limited to a maximum value. The above can easily be generalized for N input signals each having a transfer factor H

i

with 1≦i≦N. If it assumed that the processing means have N branches each corresponding to a signal i and having a power transfer factor |W

i

|

2

, for these values of |W

i

|

2

can be written:

\begin{matrix} {&LeftBracketingBar; W_{i} &RightBracketingBar;}^{2} = \frac{{&LeftBracketingBar; H_{i} &RightBracketingBar;}^{2}}{\sum_{i = 1}^{N} {&LeftBracketingBar; H_{i} &RightBracketingBar;}^{2}} & (21) \end{matrix}

It is observed that in general it is not possible to make the output signals of the subtractors exactly equal to zero, because the impulse response of the transmission paths cannot exactly be modeled by commonly used digital filters with reasonable complexity. In practice, the power of the output signals of the subtractors averaged over a given period in minimized. This way of operating the adaptive filters has turned out to be effective. In the present implementation, the functions representing the impulse response have the property that the differences of their phases is equal to the differences of the phases in the impulse responses of the transmission paths from the signal source to the receivers. This is one possibility for functions representing the impulse reponse of the transmission paths, but it is obvious that different functions could be used.

In the impulse response determining means

5

,

6

to

FIG. 3

, input signals from audio sources being here microphones

30

,

32

and

34

are converted into digital signals which are converted into block of L samples by respective series to parallel converters

36

,

38

and

40

. The output of the series to parallel converters

36

,

38

and

40

are connected to corresponding inputs of the processing means

41

, and to input of respective block delay elements

54

,

56

and

58

.

In the processing means

41

the output signal of the series to parallel converter

36

is applied to a block concatenation unit

42

. The block concatenating unit

42

constructs blocks of N+L samples from the present block of L samples and N samples from previous blocks of samples available at the output of the series to parallel converter

36

. The output of the block concatenation unit

42

is connected to an input of a frequency domain programmable filter

44

. The output of the frequency domain programmable filter

44

, carrying a processed audio signal, is connected to a first input of the combining means being here an adder

76

. The frequency domain programmable filter

44

presents blocks of N+L samples at its output.

In the same way the output signal of the series to parallel converter

38

is processed by a block concatenating unit

48

and a frequency domain programmable filter

46

and the output signal of the series to parallel converter

40

is processed by a block concatenating unit

52

and a frequency domain programmable filter

50

. Outputs of the frequency domain programmable filters

46

and

50

, carrying processed audio signals, are connected to corresponding inputs of the adder

76

.

The output of the adder

76

is connected to an input of an IFFT unit

77

which determines an Inverse Fast Fourier Transformed signal from the output signal of the adder

76

. The output of the IFFT unit

77

is connected to an input of a unit

79

which discards the last N samples of the N+L samples at the output of the IFFT unit

77

.

The output signal of the unit

79

is converted into a serial stream of samples by the parallel to series converter

78

. At the output of the parallel to series converter

78

the output signal of the audio processing arrangement is available. The output signal of the unit

79

is also applied to a block concatenating unit

74

which derives blocks of N+L samples from the present block of L samples at the output of the unit

79

and a block of N previous samples at the output of the unit

79

. The output of the block concatenating unit

74

is connected to an input of an Fast Fourier Transformer

72

which calculates a N+L points FFT from the N+L samples at its input. The output signal of the Fast Fourier Transformer

72

represents the frequency spectrum of the combined signal. This frequency spectrum is applied to inputs of frequency domain adaptive filters

62

,

66

and

68

, and to an input of a normalizer

73

. An output of the normalizer

73

is connected to inputs of the frequency domain adaptive filters

62

,

66

and

68

.

The output of the block delay element

54

is connected to a first input of a subtractor

60

. The output of the block delay element

56

is connected to a first input of a subtractor

64

and the output of the block delay element

58

is connected to a first input of a subtractor

70

. The block delay elements

54

,

56

and

58

are present to compensate the delay to which the audio signals are subjected in the frequency domain programmable filters

44

,

46

.

An output of the frequency domain adaptive filter

62

is connected to a second input of the subtractor

60

and the output of the subtractor

60

is connected to a control input of the frequency domain adaptive filter. An output of the frequency domain adaptive filter

66

is connected to a second input of the subtractor

64

and the output of the subtractor

64

is connected to a control input of the frequency domain adaptive filter. An output of the frequency domain adaptive filter

68

is connected to a second input of the subtractor

70

and the output of the subtractor

70

is connected to a control input of the frequency domain adaptive filter.

The frequency domain adaptive filters

62

,

66

and

68

are arranged to adjust their transfer function in order to minimize the power of the input signal at their control inputs. The frequency domain adaptive filters

62

,

66

and

68

provide their N+L filter coefficients to the frequency domain programmable filters

44

,

46

and

48

. These frequency domain adaptive filters determine the conjugate value of the N+L filter coefficients before using them to filter the signals received from the block concatenating units

42

,

48

and

52

.

The functions representing the impulse response are here constituted by the sets of coefficients U

1

, U

2

. . . U

M

for the frequency domain programmable filters

44

,

46

and

50

.

In the frequency domain adaptive filters

62

,

66

and

68

according to

FIG. 4

, a padding element

80

combines the L samples available at the control input of the respective frequency domain adaptive filter with N samples having a value of 0 to a block of data having N+L samples. This block of N+L samples is subjected to a N+L points Fast Fourier Transform executed by a FFT element

82

. The extension of blocks of L samples to blocks of N+L samples before executing the FFT is done to prevent distortion of the signal due to cyclic convolution effects. This measure is well known to those skilled in the art of (adaptive) digital filters.

At the output of the FFT element

82

the frequency spectrum of the signal at the control input of the frequency domain adaptive filter(=the output of the subtractor

60

,

64

and

70

respectively) is available. The output signal of the FFT element

82

is multiplied with the output signal of the normalizer

73

. The N+L components of the output signal of the normalizer

73

represents adaptation speed values determining the speed of adaptation of the coefficients of the frequency domain adaptive filter.

The output signal of the multiplier

84

is added to the output signal of a block delay element

112

by an adder

86

. The output signal of the block delay element

112

represents the previous values of the filter coefficients of the frequency domain adaptive filter. The output signal of the adder

86

is subjected to an Inverse Fast Fourier Transform executed by an IFFT element

94

. From the N+L output samples of the IFFT element

94

, the value of the final L block is set to zero by the element

96

. Subsequently the N+L samples (of which L samples are zero) are subjected to an FFT operation executed by an FFT element

110

. The combination of the IFFT element

94

, the element

96

and the FFT element

110

constitutes a “constrained” FDAF where a time domain constraint is put on the FDAF coefficients to prevent cyclic convolution effects.

The output of the FFT element

110

is connected to an input of the block delay element

112

. At the output of the block delay element

112

N+L coefficients are available for use in the filter operation. These coefficients are also passed to the corresponding programmable filter. The combination of the adder

86

, the IFFT element

94

, the element

96

, the FFT element

110

and the block delay element

112

determine the filter coefficient according to the following expression.

ν

i,k+1

=ν

i,k

+λ

i,k

·E

i,k

(22)

In (22) v

i,k+1

represents the N+L filter coefficients at instant k+1, v

i,k

represents the N+L filter coefficients at instant λ

i,k

represents the adaptation coefficients provided by the normalizer

73

to the second input of the multiplier

84

and E

k,i

represents the frequency spectrum of the error signal at the output of the subtractor

60

,

64

or

70

in FIG.

2

.

In the normalizer

73

according to

FIG. 4

, the input signal provided by the FFT unit

72

in

FIG. 2

a conjugating element

106

determines the conjugate value of said input signal. This conjugate value is multiplied with said input signal by a multiplier

104

. At the output of the multiplier

104

the power spectrum of the input signal is available. The output of the multiplier

104

is connected to an input of a multiplier

102

.

A low pass filter constituted by the multiplier

102

, an adder

100

, a multiplier

98

and a block delay element

92

determines a time average of the power spectrum of the input signal of the frequency domain adaptive filter as available at the output of the multiplier

104

. A suitable value for b is:

\begin{matrix} b = 1 - \frac{20 \cdot L}{f_{sample}} & (23) \end{matrix}

In (23) f

sample

is the sample frequency with which the audio signals are sampled and processed. A value of 32 or 64 for L has proven to be a useful value when the sample rate is equal to 8 KHz. The output of the adder

100

carrying the time averaged power spectrum is connected to a first input of a divider

88

. The output signal of the conjugating element

106

is scaled with a scaling factor 2a by a scaling element

90

. A suitable value for a is 0.01. The output signal of the scaling element

90

is connected to a second input of the divider

88

.

The divider

88

determines the values of λ

i,k

by calculating the ratio of the conjugated FFT transform (scaled with scaling factor 2a) of the input signal of the digital filter and the time averaged power spectrum of the input signal of the normalizer

73

. The value of λ

i,k

increases proportional to the ratio between the k

th

component of the spectrum of the input signal and the k

th

component of the time averaged power spectrum. This results an adaptation speed that is the same for all frequency components irrespective of their strength.

In the frequency domain programmable filter

44

,

46

and

50

according to

FIG. 6

, the input signal is applied to the input of an FFT element

120

which calculates a N+L points FFT from said input signal. A conjugating element

122

determines the conjugate value of the parameters received from the frequency domain adaptive filters

62

,

66

,

68

. A multiplier

124

calculates a filtered signal by multiplying the FFT of the input signal with the conjugated filter coefficients received from the frequency domain adaptive filters.

It is observed that a suitable choice for N is making it equal to L, but it is also possible to choose N smaller or larger than L. It is desirable to make N+L equal to a power of two in order to enable an easy implementation of the FFT and IFFT operations.

In the time domain implementation of the impulse response determining means according to

FIG. 7

the outputs of microphones

30

,

32

and

34

are connected to inputs of processing means

131

and to delay elements

186

,

188

and

190

. The processing means

131

comprise time domain programmable filters

133

,

135

and

137

.

The time domain programmable filter

133

comprises a plurality of cascaded delay elements

130

,

132

and

134

, and an adder

146

which adds the output signals of the delay elements weighted with a weighting factor W

1,1

. . . W

1,N

. The weighting is performed by the weighting elements

136

,

138

,

140

,

142

and

144

. The time domain programmable filter

135

comprises a plurality of cascaded delay elements

148

,

150

and

152

, and an adder

164

which adds the output signals of the delay elements weighted with a weighting factor W

2,1

. . . W

2,N

. The weighting is performed by the weighting elements

154

,

156

,

158

,

160

and

162

. The time domain programmable filter

137

comprises a plurality of cascaded delay elements

166

,

168

and

170

, and an adder

182

which adds the output signals of the delay elements weighted with a weighting factor W

M,1

. . . W

M,N

.

The outputs of the time domain programmable filters

133

,

135

and

137

, carrying the processed audio signals, are connected to the combination means being here an adder

184

. At the output of the adder

184

an enhanced audio signal is available. The output of the adder

184

is connected to inputs of time domain adaptive filters

191

,

193

and

195

.

The time domain adaptive filter

191

comprises a plurality of delay elements

194

,

196

and

198

. The output signals of the delay elements

194

,

196

and

198

are weighted with weighting factors W

1,1

. . . W

1,N

by weighting elements

200

,

202

,

204

,

206

and

208

. The output signals of the weighting elements

200

. . .

208

are added by an adder

192

which provides the output signal of the adaptive filter

191

.

The time domain adaptive filter

193

comprises a plurality of delay elements

226

,

228

and

230

. The output signals of the delay elements

226

,

228

and

230

are weighted with weighting factors W

2,1

. . . W

2,N

by weighting elements

216

,

218

,

220

,

222

and

224

. The output signals of the weighting elements

216

. . .

224

are added by an adder

210

which provides the output signal of the adaptive filter

193

.

The time domain adaptive filter

195

comprises a plurality of delay elements

236

,

240

and

246

. The output signals of the delay elements

236

,

240

and

246

are weighted with weighting factors W

M,1

. . . W

M,N

by weighting elements

234

,

238

,

242

,

244

and

248

. The output signals of the weighting elements

234

. . .

248

are added by an adder

232

which provides the output signal of the time domain adaptive filter

195

.

The outputs of the delay elements

186

,

188

and

190

are connected to first inputs of subtractors

212

,

214

and

230

. The delay elements

186

,

188

and

190

are present to make the impulse response of the programmable filters relatively anti-causal (earlier in time. Second inputs of the subtractors

212

,

214

and

230

are coupled to outputs of the time domain adaptive filters

191

,

193

and

195

. The outputs of the subtractors

212

,

214

and

230

are connected to control means

231

,

233

and

235

respectively. The control means are arranged to adjust the transfer function of the corresponding adaptive filter

191

,

193

and

195

in order to minimize the power of the output signal of the corresponding subtractor.

The control means

231

,

233

and

235

are arranged for adjusting the coefficients of the adaptive filters

191

,

193

and

195

according to the following expression:

W

j,k

(

n

+1)=

W

j,k

(

n

)=μ·

y[n−k

+1

]·e

j

[n

] (24)

In (24) W

j,k

(n) is the weight factor of the k

th

(k=1, 2, . . . N) weighting element in the j

th

adaptive filter, μ is a adaptation constant and e

j

[n] is the difference between the output signal of the j

th

block delay element delaying the input signal and the output signal of the j

th

adaptive filter. y

j

[n−k+1] is the over k−1 sample periods delayed output signal of the audio processing arrangement. These signals y[n−k+1] are available at the output of the delay elements of the adaptive filters. Because the adaptive filters all have the same input signals, the delay elements can be shared leading to a reduction of the required number of delay elements.

After the coefficients W

j,k

(n) have been determined, these coefficients are reversely passed to the time domain programmable filters

133

,

135

and

137

. This means that the coefficients corresponding to the first taps in the adaptive filters are passed to coefficients of the last taps in the corresponding programmable filter.

The functions representing the impulse responses are here the sets of coefficients W

1,1

, . . . W

1,N

; . . . W

M,1

, . . . W

M,N

. As explained before, these functions representing the impulse responses are passed to the correlation function correlation means

7

.

Claims

1. A signal source localization arrangement comprising:at least three receivers having different positions; delay difference estimation means for estimating a delay difference between signals received by at least two receivers; and position determining means for determining from the delay difference a signal source location; impulse response determining means for determining a plurality of functions representing impulse responses of paths between the signal source and the at least two receivers, wherein the delay estimation means is arranged for determining the delay difference from the plurality of functions and wherein the impulse response determining means comprises adjustable filters for deriving filtered signals from the signals provided by the receivers, the signal source localization arrangement comprising combining means for deriving a combined signal from the filtered signals, in that the impulse response determining means comprises control means for controlling the adjustable filters in order to maximize a power measure of the combined signal, and in that the control means are arranged for limiting a combined power gain measure of the filtered signals to a predetermined value.
2. The signal source localization arrangement according to claim 1, wherein the delay estimation means is arranged for determining the delay difference by calculation of a correlation function from the plurality of functions.
3. The signal source localization arrangement according to claim 1, wherein the control means comprise a plurality of further adjustable filters having a transfer function being the conjugate of the transfer function of the adjustable filters, said further adjustable filters being arranged for deriving from the combined audio signal filtered combined audio signals, and in that the control means are arranged for maximizing the power measure of the combined signal, and for restricting a combined power gain measure of the processed signals to a predetermined value by controlling the transfer functions of the adjustable filters and the further adjustable filters in order to minimize a difference measure between the input signals and the filtered combined signal corresponding to said input signals.
4. A device comprising:delay estimation means for estimating a delay difference between signals received by at least two receivers,: and impulse response determining means for determining a plurality of functions representing impulse responses of paths between a signal source and the receivers, wherein the delay estimation means determines a delay value from the plurality of functions, and wherein the impulse response determining means comprises adjustable filters for deriving filtered signals from the signals provided by the receivers, the signal source localization arrangement comprising combining means for deriving a combined signal from the filtered signals, in that the impulse response determining means comprises control means for controlling the adjustable filters in order to maximize a power measure of the combined signal, and in that the control means are arranged for limiting a combined power gain measure of the filtered audio signals to a predetermined value.
5. The delay estimation means according to claim 4, wherein the delay estimation means is arranged for determining the delay difference by correlating the plurality of functions.
6. The delay estimation means according to claim 4, wherein the control means comprises a plurality of further adjustable filters having a transfer function being the conjugate of the transfer function of the adjustable filters, said further adjustable filters being arranged for deriving from the combined audio signal filtered combined audio signals, and in that the control means are arranged for maximizing the power measure of the combined audio signal, and for restricting a combined power gain measure of the processed audio signals to a predetermined value by controlling the transfer functions of the adjustable filters and the further adjustable filters in order to minimize a difference measure between the input audio signals and the filtered combined audio signal corresponding to said input audio signals.
7. A video communication arrangement comprising:camera pointing means for pointing a camera in a direction towards a signal source; a plurality of receivers having different positions; a signal source localization arrangement for determining a position of the signal source including delay estimation means for estimating a delay difference between the signals received by the receivers, and position determining means for determining from the delay difference a signal source location, impulse response determining means for determining a plurality of functions representing the impulse responses of the paths between the signal source and the receivers, wherein the delay estimation means is arranged for determining the delay value from the plurality of functions, wherein the impulse response determining means comprise adjustable filters for deriving filtered signals from the signals provided by the receivers, the signal source localization arrangement comprising combining means for deriving a combined signal from the filtered signal and the impulse response determining means comprises control means for controlling the adjustable filters in order to maximize a power measure of the combined signal, and in that the control means are arranged for limiting a combined power gain measure of the filtered audio signals to a predetermined value.
8. A signal source localization method using at least three receivers having different positions, comprising the steps of:estimating a delay difference between the signals received by at least two receivers; determining from the delay difference a signal source location; determining a plurality of functions representing the impulse responses of the paths between the signal source and the receivers, by filtering the signals provided by the receivers and combining filtered signals, controlling the filtering in order to maximize a power measure of the combined signal, and limiting a combined power gain measure of the filtered signals to a predetermined value.; and determining the delay value from the plurality of functions.

Priority Claims (1)

Number	Date	Country	Kind
98203802	Nov 1998	EP

US Referenced Citations (8)

Number	Name	Date	Kind
3601530	Edson et al.	Aug 1971	A
4536887	Kaneda et al.	Aug 1985	A
5465302	Lazzari et al.	Nov 1995	A
5686957	Baker	Nov 1997	A
5737431	Brandstein et al.	Apr 1998	A
6469732	Chang et al.	Oct 2002	B1
6483923	Marash	Nov 2002	B1
6618485	Matsuo	Sep 2003	B1

Foreign Referenced Citations (1)

Number	Date	Country
0615387	Sep 1994	EP

Non-Patent Literature Citations (1)

Entry
“Voice Source Localization for Automatic Camera Pointing System in Videoconferencing” by Hong Wang and Peter Chu, IEEE, ASSP Workshop on Applications of Signal Processing to Audio and Acoustics, 1997.

Signal localization arrangement

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