Space-time doppler coding schemes for time-selective wireless communication channels

Abstract

Space-time Doppler (STDO) coding techniques are describe for time-selective channels. In particular, a STDO coded system is capable of achieving a maximum Doppler diversity for time-selective frequency-flat channels. As demonstrated herein, this maximum diversity may be represented as Q+1, where Q is the number of bases in a generalized Basis Expansion Model (BEM) for time-frequency selective channels. In addition, linearly precoded transmissions are described to collect this maximum diversity gain.

Description

TECHNICAL FIELD

[0003] The invention relates to communication systems and, more particularly, transmitters and receivers for use in wireless communication systems.

BACKGROUND

[0004] Broadband wireless communications call for high data-rate and high performance. Temporal channel variations and time-selective fading, however, may result in lower data-rate and an overall decrease in system performance. Time-selectivity may arise for a variety of reasons, such as oscillator drifts, phase noise, multipath propagation, and relative motion between the transmitter and the receiver. Therefore, it is important for broadband wireless applications to design single- or multi-antenna systems that account for time-varying channels.

SUMMARY

[0005] In general, space-time doppler (STDO) coding techniques are described for time-selective channels. In particular, the described STDO coded system is capable of achieving a maximum Doppler diversity for time-selective frequency-flat channels.

[0006] As demonstrated herein, this maximum diversity may be represented as Q+1, where Q is the number of bases in a generalized Basis Expansion Model (BEM) for time-frequency selective channels. In addition, linearly precoded transmissions are described to collect this maximum diversity gain.

[0007] The STDO coded design utilizes knowledge of a maximum Doppler spread at the transmitter. For exemplary purposes, three STDO codes are described. Each is capable of collecting the maximum STDO diversity. In addition, each offers a uniquely desirable tradeoff among: high spectral efficiency, low decoding complexity, and high coding gain.

[0008] The first STDO codec comprises a digital phase sweeping (DPS) scheme that renders the set of time-selective fading multi-antenna channels mathematically equivalent to a single combined channel offering the maximum joint STDO diversity. The other two STDO codecs are block orthogonal designs, and emerge from a simple but neat duality property that we establish between the time-selectively captured by the BEM, and the frequency-selectivity that can be approximated well by a finite impulse response (FIR) tapped delay line model. This duality property underpins the transformation of ST designs for single- and multi-carrier transmissions over frequency-selective channels to our time-selective channels parameterized by the BEM.

[0009] In one embodiment, a wireless communication device comprises a first coder, a second coder, and an antenna. The first coder linearly precodes a data stream to produce a precoded data stream. The second coder processes the precoded data stream in accordance with a doppler spread of time-selective communication channels to produce a transformed data stream. The antenna outputs a waveform in accordance with the transformed data stream.

[0010] In another embodiment, a wireless communication device comprises a demodulator and a decoder. The demodulator receives a waveform from a plurality of time-selective multipath communication channels and produces a demodulated data stream. The decoder decodes the demodulated data in accordance with a Basis Expansion Model (BEM) that parameterizes time-selective communication multi-path channels as a function of frequency-selective multi-path communication channels.

[0011] In another embodiment, a method comprises estimating a doppler spread for a plurality of channels from a transmitter to a receiver, and applying a linear precoder to a data stream to form a precoded data stream. The method further comprises processing the precoded data stream in accordance with the doppler spread to produce a transformed data stream; and transmitting a waveform with an antenna in accordance with the transformed data stream.

[0012] In another embodiment, a computer-readable medium comprises instructions to cause a programmable processor to estimate a doppler spread for a plurality of channels from a transmitter to a receiver, and apply a linear precoder to a data stream to form a precoded data stream. The instructions further cause the processor to process the precoded data stream in accordance with the doppler spread to produce a transformed data stream, and transmit a waveform with an antenna in accordance with the transformed data stream.

[0013] The details of one or more embodiments of the invention are set forth in the accompanying drawings and the description below. Other features, objects, and advantages of the invention will be apparent from the description and drawings, and from the claims.

BRIEF DESCRIPTION OF DRAWINGS

[0014] FIG. 1 is a block diagram illustrating an exemplary telecommunication system in which a transmitter and receiver implement the space-time Doppler (STDO) coding techniques described herein.

[0015] FIG. 2 illustrates digital phase sweeping (DPS) techniques.

[0016] FIG. 2A is a flowchart that illustrates exemplary operation of the transmitter and receiver of FIG. 1.

[0017] FIG. 3 illustrates channel matrix partitioning.

[0018] FIG. 4 is a block diagram illustrating an example CP-based STDO transceiver.

[0019] FIG. 5 is a block diagram illustrating an example ZP-based STDO transceiver.

[0020] FIGS. 6-10 are graphs that illustrate exemplary results of simulations of the described techniques.

DETAILED DESCRIPTION

[0021] FIG. 1 is a block diagram illustrating a telecommunication system 2 in which transmitter 4 communicates data to receiver 6 through wireless channels 8. In general, transmitter 4 employs space-time doppler (STDO) coding techniques to achieve a maximum doppler diversity for time-selective channels 8.

[0022] Transmitter 4 includes a plurality of antennas 201-20Nt for transmitting data to receiver 6. In particular, each antenna 20 outputs a waveform that propagates to receiver 6 through one or more multi-path communication channels. Transmitter 4 may output the waveforms using one of a number of conventional multi-user transmission formats, including Code Division Multiple Access (CDMA) and Orthogonal Frequency Division Multiplexing (OFDM). The former is an example of single-carrier multiple access scheme, while the latter is a multi-carrier scheme. OFDM has been adopted by many standards including digital audio and video broadcasting (DAB, DVB) in Europe and high-speed digital subscriber lines (DSL) in the United States. OFDM has also been proposed for local area mobile wireless broadband standards including IEEE802.11a, IEEE802.11g, MMAC and HIPERLAN/2.

[0023] The techniques described herein may be applied to uplink and/or downlink transmissions, i.e., transmissions from a base station to a mobile device and vice versa. Consequently, transmitters 4 and receivers 6 may be any device configured to communicate using a multi-user wireless transmission including a cellular distribution station, a hub for a wireless local area network, a cellular phone, a laptop or handheld computing device, a personal digital assistant (PDA), and the like.

[0024] As illustrated, transmitter 4 includes an outer space-time linear coder (OSTDO coder) 16, a middle space-time doppler coder (MSTDO coder) 17, a set of inner space-time doppler coders (ISTDO coder) 22, a plurality of parallel-to-serial (P/S) converters 181-18Nt., and a plurality of transmit antennas 201-20Nt. Receiver 6 includes a plurality of receive antennas 281-28Nr, a plurality of inner space-time decoders (ISTDO coder) 321-32Nr, a middle space-time doppler coder (MSTDO coder) 34, and an outer space-time doppler coder (STDO coder) decoder 36.

[0025] Channels 8 experience multipath propagation, i.e., a number of reflected or scattered rays arrive at the receiving end. Time-selectivity arises when all the rays arrive at the receiver almost simultaneously with a common propagation delay that can be set to zero without loss of generality. Let h(t) denote the time-varying impulse response of the resulting channel that includes transmit-receive filters as well as time-selective propagation effects; and let H(f) denote the Fourier transform of h(t). Although the bandwidth of h(t) over a finite time horizon is theoretically infinite, we practically have that H(f)=0 for f ∉ [−fmaxTfmax], where fmax is the maximum frequency offset (Doppler shift) of all the rays. Considering that a block of N symbols with symbol period Ta is time-limited, we sample H(f) along f with period l/(NTa), and collect Q+1 samples 1${\left\{H\left(q/\left({\mathrm{NT}}_s\right)\right)\right\}}_{q=-Q/2}^{Q/2},$

[0026] where Q:=2┌fmaxTaN┐. Transforming these frequency-domain samples back to the time-domain and sampling along the time t, we obtain samples h(n). Using the serial index n, we can describe the block index as └n/N┘, and write our discrete-time baseband equivalent channel model as: 2$\begin{array}{cc}h\left(n\right)=\sum _{q=0}^Q{h}_q\left(\lfloor n/N\rfloor \right)e^{j{\omega }_0\left(n\mathrm{mod}N\right)},& \left(1\right)\end{array}$

[0027] where wq:=2π(q-Q/2)/N. Eq. (1) constitutes our Basis Expansion Model (BEM). Over each interval, say the kth (k:=└n/N┘), the BEM represents h(t) for t ∈ [kNTa(k+1)NTa) using:

[0028] a) Q+1 coefficients {hq(k)}q=0Q that remain invariant per block, but are allowed to change with k; and,

[0029] b) Q+1 Fourier bases that capture even rapid time variations, but are common ∀k.

[0030] The complex exponential bases allow h(n) to vary not only across blocks, but also within every block. Notice also that the only physical parameter dictating the BEM order Q is the Doppler spread fmax, since N and Ta in Q:=2┌fmaxNTa┐ are up to the designer's choice.

[0031] The information bearing symbols {s(n)} are drawn from a finite alphabet A, and parsed into blocks of size P×1: s(k):=[s(kP), . . . , s((k+1)P−1)]T. Each block s(k) is linearly preceded by the K×P matrix Θ, resulting into u(k):=Θs(k). This operation will be termed the outer Space-Time-Doppler (STDO) coding. Each block u(k) is further transformed into N2 blocks {cμ(k)}μm1Nt of size N×1 by a mapper M(.): {cμ(k)}M(u(k)). This operation will be termed the middle STDO coding. Each block cμ(k) is finally linearly processed by the N×N matrix Φμ resulting into vμ(k) :=Φμcμ(k). This operation will be termed the inner STDO coding. Not all specific STDO designs will rely on all three (outer, middle, inner) stages of our unifying structure. If one, e.g., the inner stage is inactive, we will simply set Φμ=Iμ.

[0032] The sequence vμ(n) obtained by serial-to-parallel converting the blocks {vμ(k)} is then pulse-shaped, carrier modulated, and transmitted from the μth transmit-antenna. The nth sample at the νth antenna's receive-filter output is: 3$x_{\mu }\left(n\right)=\sum _{\mu =1}^{N_t}{h}^{\left[v,\mu \right]}\left(n\right)v_{\mu }\left(n\right)+{\zeta }_v\left(n\right),\forall v\in \left[1,N_r\right],$

[0033] where h(p,μ)(n) is the time-selective channel response from the μth transmit-antenna to the νth receive-antenna (notice the channel dependence on n), and ζν(n) is complex additive white Gaussian noise (AWGN) at the νth receive-antenna with mean zero, and variance σζ2=No. According to (1), we have ∀v∈[1,N],μ∈[1,N] 4$h^{\left[v,\mu \right]}\left(n\right):=\sum _{q=0}^Q{h}_q^{\left(v,\mu \right)}\left(\lfloor n/N\rfloor \right)e^{j{\omega }_0\left(n\mathrm{mod}N\right)},$

[0034] where wq:=2π(q−Q/2)/N, and Q:=2┌fmaxNTa┐ as in (1).

[0035] At each receive-antenna, the symbol rate sampled sequence xν(i) at the receive-filter output is serial-to-parallel converted to form the N×1 blocks xν(k) :=[xp(kN),xv(kN+1), . . . , xv(kN+N−1)]T. The matrix-vector counterpart of (2) can then be expressed as: 5$x_v\left(k\right)=\sum _{\mu =1}^{N_t}{D}_H^{\left[v,\mu \right]}\left(k\right)v_{\mu }\left(k\right)+{\zeta }_v\left(k\right),\forall v\in \left[1,N_r\right],$

[0036] where DH(υ,μ)(k) is an N×N diagonal channel matrix that obeys the BEM 6$D_H^{\left[v,\mu \right]}\left(k\right):=\sum _{q=0}^Q{h}_q^{\left(v,\mu \right)}\left(k\right)D_q,$

[0037] with Dq:=diag[1,exp(jwq), . . . , exp(jwq(N−1))], and ζν(k)'s independent identically distributed (i.i.d.) AWGN noise vectors, which are defined similar to xν(k). Each block xp(k) is linearly processed by the N×N matrix Tνto yield yν(k) :=Tνxν(k). This operation is termed the inner STDO decoding. The blocks yν(k) are further “de-mapped” to a block z(k), by {overscore (M)}(·): z(k) :={overscore (M)}({yp(k)}μm1Nt). This operation is termed the middle STDO decoding. The block z(k) is finally decoded by (·) to obtain an estimate of s(k) as: s(k) :=(z(k)). This operation is termed the outer STDO decoding.

[0038] The following operating conditions are assumed for system 2:

[0039] A2) BEM coefficients hz(ν,μ) are zero-mean, complex Gaussian random variables;

[0040] A3) Channel state information (CSI) is available at the receiver, but unknown at the transmitter;

[0041] A4) High SNR is considered for deriving the STDO diversity and coding gains.

[0042] When transmissions experience rich scattering, and no line-of-sight is present, the central limit theorem validates A2). Notice that we allow not only for independent random channel coefficients, but also for correlated ones.

[0043] Pairwise error probability (PEP) to design our optimality criteria. Define the PEP P(s→s′|DH(ν,μ), ∀ν, μ) as the probability that maximum likelihood (ML) decoding of s erroneously decides s′ instead of the actually transmitted s.

[0044] As xνin (3) depends on s, we will use ML decoding to detect s from {xν}ν=1Nr. Conditioned on the DH(ν,μ)'s, the Chernoff bound yields: 7$\begin{array}{cc}P\left(s\to s^{\prime }❘D_H^{\left[v,\mu \right]},\forall v,\mu \right)\le \exp \left(-\frac{d^2\left(\tilde{x},{\tilde{x}}^{\prime }\right)}{4N_0}\right),& \left(5\right)\end{array}$

[0045] where {tilde over (x)} :=[(Σμ=1NtDH(1,μ)vμ)T, . . . , (Σμ=1NtDH(Nr,μ)vμ)T]T, the distance in the exponent is 8$d^2\left(\tilde{x},{\tilde{x}}^{\prime }\right)=\sum _{v=1}^{N_r}{\&LeftDoubleBracketingBar;\sum _{\mu =1}^{N_t}{D}_H^{\left[v,\mu \right]}{e}_{\mu }\&RightDoubleBracketingBar;}^2,$

[0046] and eμ:=vμ−v′μ. Based on (4), and using the commutativity between a diagonal matrix and a vector we can write DH(ν,μ)eμ=De(μ)dH(ν,μ), with De(μ) :=diag[eμ], dH(ν,μ) :=Σq=0Qhq(ν,μ)wq, and wq :=[1, exp(jwq), . . . , exp(jwq(N−1))]T. With these definitions, we can rewrite d2({tilde over (x)},{tilde over (x)}′) as:

d2({tilde over (x)},{tilde over (x)}′)=hHAch,

[0047] where

h:=[h0(1,1), . . . , hQ(1,1), . . . , hQ(1,Nt), . . . , hQ(Nr,Nt)]T, NrNt(Q+1)×1

Ae :=INr{circle over (x)} (ΦeHΦe), NrNt(Q+1)×NrNt(Q+1)

Φe :=[Dc(1)Ω, . . . , De(Nt)Ω], N×Nt(Q+1)

Ω :=[w0, . . . , wQ]. N×(Q+1)  (6)

[0048] Since our analysis will allow for correlated channels, we will denote the NtNr(Q+1)×NtNr(Q+1) channel correlation matrix and its rank, respectively, by:

Rh :=E[hhH], and Th :=rank(Rh)≦NtNr(Q+1).  (7)

[0049] Eigenvalue decomposition of Rh yields Rh=UhΛhUhH, where Λh :=diag[σ02, . . . , σrh2−1], and Uh is an NtNr(Q+1)×rh para-unitary matrix satisfying UhHUh=Irh. Define an rh×1 normalized channel vector {overscore (h)} with entries {overscore (h)}q(ν,μ) are that i.i.d. Gaussian distributed, with zero mean and unit variance. It is easy to show that h and UhΛh1/2{overscore (h)} have identical distributions. Therefore, the PEP remains statistically invariant when one replaces h by UhΛh1/2{overscore (h)}. To proceed, let us define the rh×rh matrix:

Ψe :=(UhΛh1/2)HAcUhΛh1/2.  (8)

[0050] Since Ψe is Hermitian, there exists a unitary matrix Ue, and a real non-negative definite diagonal matrix Λe so that UeHΨeUe=Λe. The rh×rh diagonal matrix Λe :=diag[λ0, . . . , λrh−1], holds on its diagonal the eigenvalues of Ψe, that satisfy λm≧0, ∀m ∈ [0, rh−1]. The vector {overscore (h)}=Ue{overscore (h)} has the same distribution as {overscore (h)}, because Ue is unitary. Thus, d2({overscore (x)},{overscore (x)}′) can be rewritten in terms of the eigenvalues of the matrix Ψe as 9$\begin{array}{cc}{d}^2\left(\tilde{x},{\tilde{x}}^{\prime }\right)=\sum _{m=0}^{{\tau }_h-1}{\lambda }_m{\&LeftBracketingBar;{\tilde{h}}_m\&RightBracketingBar;}^2.& \left(9\right)\end{array}$

[0051] Since we wish our STDO coders to be independent of the particular channel realization, it is appropriate to average the PEP over the independent Rayleigh distributed |{overscore (h)}m|2's. If re :=rank(Ψe), then re eigenvalues of Ψe are non-zero; without loss of generality, we denote these eigenvalues as λ0≧ . . . ≧λre−1. At high SNR, the resulting average PEP is bounded as follows [c.f. (5)]: 10$\begin{array}{cc}P\left(s\to s^{\prime }\right)\le \prod _{m=0}^{r_e-1}{\left(\frac{{\lambda }_m}{4N_0}\right)}^{-1}={\left(G_{e,c}\frac{1}{4N_0}\right)}^{-G_{e,d}},& \left(10\right)\end{array}$

[0052] where Ge,d :=re is the diversity order, and Ge1c:=(rc−1m=0λmis the coding gain for the error pattern e :=s−s′. Accounting for all possible pairwise errors, the diversity and coding gains for out STDO multi-antenna systems are defined, respectively, as: 11$\begin{array}{cc}{G}_d:=\underset{\forall e\ne 0}{\min }{G}_{e,d},\mathrm{and}{G}_c:=\underset{\forall e\ne 0}{\min }{G}_{e,c}.& \left(11\right)\end{array}$

[0053] Because the performance of STDO depends on both Gd and Ge, it is important to maximize both of them. But before specializing to particular STDO designs that accomplish this, we wish to quantify the maximum possible Gd and Gc supplied by our BEM.

[0054] Eq. (10) discloses that Gd depends on the rank re of Ψe. As the rank can not exceed the dimensionality, checking the dimensionality of Ψe, we recognize that by appropriately designing our transmission, it is possible to achieve a maximum (and thus optimum) diversity gain 12$\begin{array}{cc}{G}_d^{\max }=r_h\le N_t{N}_r\left(Q+1\right),& \left(12\right)\end{array}$

[0055] if and only if the matrix Ψe in (8) has full rank rh, ∀e≠0.

[0056] Eq. (10) also indicates that Ge,c is the product of the non-zero eigenvalues of Ψe. It is not easy however, to express Gc in closed form. But we can benchmark it, when Rh has full rank NrNt(Q+1).

[0057] Since Rh is not known at the transmitter, we will allocate the transmit-power equally to the Nt sub-streams corresponding to the Nt transmit-antennas. For this reason, we set 13$\begin{array}{cc}E\left[v_{\mu }^{\mathscr{H}}{v}_{\mu }\right]=\frac{1}{N_t}E\left[s^{\mathscr{H}}s\right]=\frac{P}{N_t}{\sigma }_s^2,& \left(13\right)\end{array}$

[0058] where σs2 is the power per information symbol. If the mapping from s to vμsatisfies 14$\begin{array}{cc}{v}_{\mu }=\sum _{p=0}^{P-1}{a_p^{\left(\mu \right)}\left[s\right]}_p+{b_p^{\left(\mu \right)}\left[s\right]}_p^{*},\forall \mu \in \left[1,N_t\right],& \left(14\right)\end{array}$

[0059] where ap(μ) and bp(μ) are N×1 vectors, then we call this ST transmitter a linearly coded one. 15$\begin{array}{cc}{G}_c^{\max }={\left(\det \left(R_h\right)\right)}^{\frac{1}{r_h}}\frac{d_{\min }^2}{N_t},& \left(15\right)\end{array}$

[0060] where dmin is the minimum Euclidean distance of the constellation points in the finite alphabet As.

[0061] In deriving performance bounds, we have assumed ML decoding which comes with high computational complexity. Therefore, when we design the STDO encoders to guarantee the maximum diversity order, we will keep in mind the need to reduce decoding complexity while preserving the optimality in decoding. Before we proceed to design STDO coders, we summarize our results so far in the following proposition:

[0062] Proposition 1 Consider (Nt,Nr) multi-antenna transmissions through time-selective channels adhering to a BEM as in (1) with Q+1 bases. If the correlation matrix of the channel coefficients in (6) has rank rh, then the maximum diversity order of transmissions in (3) is 16$G_d^{\max }=r_h\le N_t{N}_r\left(Q+1\right).$

[0063] For linearly coded systems, if Rh has full rank rh=NrNt(Q+1), then the maximum coding gain is 17$G_c^{\max }={\left(\det \left(R_h\right)\right)}^{\stackrel{\perp }{r_h}}{d}_{\min }^2/N_t.$

[0064] For the DPS method, the middle STDO encoder M(·) is just a power splitter (see FIG. 1). By equally allocating the signal power, we obtain cμ=u/{square root}{square root over (N)}t, ∀μ∈ [1, Nt]. This means that for DPS we have K=N. Using (3), xνand s can then be related via: 18$\begin{array}{cc}{x}_v=\frac{1}{\sqrt{N_t}}\sum _{\mu =1}^{N_t}{D}_H^{\left(v,\mu \right)}{\Phi }_{\mu }\Theta s+{\zeta }_v,v\in \left[1,N_r\right].& \left(16\right)\end{array}$

[0065] Observing (4), we notice that different channels share the same exponential bases, but they have different channel coefficients. Suppose that we shift the Q+1 bases of each chamnel corresponding to one of the Nt transmit antennas so that all the bases are consecutive on the FFT grid of complex exponentials, as shown in FIG. 2 for Q+1=Nt=3. Then we can view the Nt channels to each receive-antenna as one equivalent time-selective channel with Nt(Q+1) bases. To realize this intuition, we select the matrices {Φμ}μ=1Nt, which determine the inner STDO encoder, as

Φμ=diag[1,ejφμ, . . . , ejφμ(N−1)], ∀μ∈ [1, Nt],  (17)

[0066] where φμ=2π(μ−1)(Q+1)/N. As Φ1=I, the exponentials of the channel corresponding to the first (μ=1) transmit antenna remain unchanged. But those corresponding to the second channel (μ=2) are shifted from their original location in 19${\left\{D_q\right\}}_{q=0}^Q\mathrm{to}{\left\{D_q\right\}}_{q=Q+1}^{2Q+1},$

[0067] after multiplication with the DPS matrix Φ2 that takes place at the second transmit-antenna; i.e., 20${\left\{D_q{\Phi }_2\right\}}_{q=0}^Q={\left\{D_q\right\}}_{q=Q+1}^{2Q+1}.$

[0068] Proceeding likewise with all Nt DPS matrices, it follows that Eq. (16) can be rewritten as 21$\begin{array}{cc}{x}_v=\frac{1}{\sqrt{N_t}}\sum _{q=0}^{N_t\left(Q+1\right)-1}{h}_q^{\left(v\right)}{D}_q\Theta s+{\zeta }_v,\forall v\in \left[1,N_r\right],& \left(18\right)\end{array}$

[0069] where 22$h_q^{\left(v\right)}:=h_{q\mathrm{mod}\left(Q+1\right)}^{\left(v_t\lfloor q/\left(Q+1\right)\rfloor +1\right)}.$

[0070] Comparing (16) with (18), we arrive at:

[0071] Property 1: DPS converts the Nt transmit-antenna system, where each channel can be expressed via Q+1 exponential bases, to a single transmit-antenna system, where the equivalent channel is expressed by Nt(Q+1) exponential bases.

[0072] Notice that since Φμ operates in the digital domain, the sweeping wraps the phases around (−π,π) which explains why DPS does not incur bandwidth expansion.

[0073] Remark 1 To avoid overlapping the shifted bases, we should make sure that N>Nt(Q+1). From the definition of Q :=2┌fmaxNTs┐, we have that for fixed fmax and N, we can adjust sampling period Ts to satisfy this condition. As for each receive-antenna, we have Nt(Q+1) unknown BEM coefficients corresponding to Nt channels every N symbols, this condition guarantees that the number of unknowns is less than the number equations. Therefore, even from a channel estimation point of view, this condition is reasonable.

[0074] With the equivalence established by Property 1, our outer STDO codec, which is determined by Θ and (·), can be any single-input codec that achieves the maximum diversity gain for the single transmit-antenna time-selective channels corresponding to each receive-antenna. From [16, Proposition 2], we know that ML decoding by means of (·) achieves the maximum diversity gain rh if the linear precoder Θ is designed in such a way that Θe has at least Nt(Q+1) non-zero entries, for all possible error vectors e=s−s′≠0. However, as Nt increases, the block length N increases as well (see Remark 1). ML decoding for the entire N×1 block entails high computational complexity. To reduce the decoding complexity, we will split the design of the outer STDO encoder Θ in groups of smaller size.

[0075] In particular, we design Θ using a Grouped Linear Constellation Precoding (GLCP) scheme described in U.S. Provisional Application Serial No. 60/374,935, entiled “LINEAR CONSTELLATION PRECODING FOR FADING COMMUNICATION CHANNELS,” filed Apr. 22, 2002, and U.S. patent application Ser. No. ______, filed Apr. 21, 2003, entitled “WIRELESS COMMUNICATION SYSTEM HAVING LINEAR ENCODER,” the entire contents of which are incorporated herein by reference.GLCP provides a means of reducing decoding complexity without sacrificing diversity. Here, we design the outer STDO coder 16 based on the BEM for time-selective channels described herein. In particular, we select the transmitted block size N=NgNsub, and demultiplex the information vector s into Ng groups: {sg}g=0Ng−1. Each group has length Nsub, and contains the symbols collected in a vector sg as follows:

sg=[[s]Nsubg, . . . , [s]Nsub(g+1)−1]T, ∀g ∈[0, Ng−1].  (19)

[0076] Correspondingly, we define the linearly precoded block of the gth group as:

ug=Θsubsg, ∀g ∈[0, Ng−1],  (20)

[0077] where Θsub is an Nsub×Nsub matrix. The overall transmitted block u consists of multiplexed sub-blocks {ug}g=0Ng−1 as follows:

u=[[u0]0, . . . , [uNg−1]0; . . . ; [u0]Nsub−1, . . . , [uNg−1]Nsub−1]T.  (21)

[0078] Notice that u can be obtained from {ug}g=0N−1's via a block interleaver with depth Nsub, Equivalently, we can relate u with s as 23$\begin{array}{cc}u=\Theta s,\mathrm{with}\Theta :=\left[\begin{array}{c}\begin{array}{c}{I}_{N_g}\otimes {\theta }_1^T\\ ⋮\end{array}\\ I_{N_g}\otimes {\theta }_{N_{\mathrm{sub}}}^T\end{array}\right],& \left(22\right)\end{array}$

[0079] where θmT is the mth row of Θsub. Equations (19)-(21), or equivalently (22), summarize our STDO transmitter design based on DPS.

[0080] Following the “reverse order” of DPS encoding, we start from the inner decoder. The inner decoder for the νth receive antenna is designed as Tν=IN, ∀ν∈[1, Nr]. Hence, in the unifying block-diagram of FIG. 1, we have yν=xν, ∀ν∈[1, Nr]. Let us denote the equivalent faster single transmit-antenna channel matrix as: 24$\begin{array}{cc}{D}_H^{\left(v\right)}=\sum _{q=0}^{N_t\left(Q+1\right)-1}{h}_q^{\left(v\right)}{D}_q,\forall v\in \left[1,N_r\right].& \left(23\right)\end{array}$

[0081] Since the received blocks yνfrom all Nr receive-antennas contain the information block s, we need to combine the information from all received blocks to decode s. To retain decoding optimality, we choose the maximum ratio combining (MRC) method. The MRC for xν=yνin (18), amounts to selecting the middle decoder {overscore (M)}(·) as [c.f. (23)] 25$\begin{array}{cc}G={\left(\sum _{v=1}^{N_r}{D_H^{\left(v\right)}\left(D_H^{\left(v\right)}\right)}^{*}\right)}^{-\frac{1}{2}}\left[{\left(D_H^{\left(1\right)}\right)}^{*}\dots {\left(D_H^{\left(N_r\right)}\right)}^{*}\right].& \left(24\right)\end{array}$

[0082] Existence of the inverse in (24), requires (only for the DPS design) the channels to satisfy: 26$\begin{array}{cc}\mathrm{Channels}{D}_H^{\left(v\right)}\mathrm{are}\mathrm{coprime};i.e.,\det \left(\sum _{v=1}^{N_r}{D_H^{\left(v\right)}\left(D_H^{\left(v\right)}\right)}^{*}\right)\ne 0.& \mathrm{A5})\end{array}$

[0083] Assumption A5) is more technical rather than restrictive, since it requires that not all equivalent channels are identically zero at the same time slot. For random channels, A5) excludes an event with probability measure zero.

[0084] With the MRC of (24), the output z of {overscore (M)}(·), is given by: 27$\begin{array}{cc}z=G\left[\begin{array}{c}{y}_1\\ ⋮\\ y_{N_r}\end{array}\right]=\frac{1}{\sqrt{N_t}}{\left(\sum _{v=1}^{N_r}{D_H^{\left(v\right)}\left(D_H^{\left(v\right)}\right)}^{*}\right)}^{\frac{1}{2}}\Theta s+\eta ,& \left(25\right)\end{array}$

[0085] where η :=G|ζ1T, . . . , ζNrT]T. Thanks to A5), it can be verified that G satisfies GGH=I. Since the ζν's are i.i.d. AWGN noise vectors, the noise vector η retains their whiteness.

[0086] Following MRC, we split z into Ng groups: 28$\begin{array}{cc}{z}_g=\frac{1}{\sqrt{N_t}}{D}_{H,g}{\Theta }_{\mathrm{sub}}{s}_g+{\eta }_g,\forall g\in \left[0,N_g-1\right],& \left(26\right)\end{array}$

[0087] where zg :=[[z]g,[z]Nsub+g, . . . , [z]Nsub(Ng−1)+g]T, DH,g is the corresponding diagonal sub-matrix from 29${\left(\sum _{v=1}^{N_r}{D_H^{\left(v\right)}\left(D_H^{\left(v\right)}\right)}^{*}\right)}^{\frac{1}{2}}$

[0088] for the gth group; and similarly defined, ηg is the corresponding AWGN noise vector. Maximum likelihood (ML) decoding can, for example, be implemented by applying a Sphere Decoding (SD) algorithm of sub-blocks zg of small size Nsub. Compared to the exponentially complex ML decoder, the SD offers near-ML performance at complexity that is polynomial in Nsub, (Nsubα). The SD complexity depends on the block size Nsub, but unlike ML, it is independent of the constellation size.

[0089] FIG. 2A is a flowchart that illustrates operation of the DPS-based space-time multipath techniques describe herein. For exemplary purposes, the operation is described in reference to FIG. 1.

[0090] Given Nt, Nr and Q, transmitter 4 selects the number of groups Ng, and the corresponding group size Nsub depending on affordable complexity; and selects N=P=K=NgNsub>Nt(Q+1) (step 40).

[0091] Outer (OSTDO) coder 16 applies the Nsub×Nsub linear constellation precoder Θsub to form a precoded data stream, i.e., the precoded vectors u, according to equations (19)-(21) (step 42). Middle STDO (MSTDO) coder 17 splits the power of u to form mirrored precoded data streams cμ=u{square root}{square root over (N)}t (step 44).

[0092] Inner STDO coders 22 apply DPS via Φμ, and transmit vμ=Φμcμ, ∀μ∈ [1, Nt] (step 46). In particular, transmitter 4 has the knowledge of the upper-bound of the number of bases for the time-varying channels from transmitter 4 to receiver 6. ISTDO coders 22 process the precoded data streams so that the channels from transmitter 4 to receiver 6 can be equivalently viewed as one channel with more bases. In particular, ISTDO coders 22 compute a single channel vector from doppler offsets for the channels, and processes the mirrored precoded data streams with the single channel vector to shift the delay lag of each of the channels so that channel taps become consecutive within a Fast Fourier Transform (FFT) grid. Finally, transmitter 4 modulates each block vμ to produce a transmission waveform via transmission antennas 20 (step 48).

[0093] Receiver 6 receives a waveform via receive antennas 28, and demodulates the received waveform (step 50). Next, receiver 6 performs MRC of blocks from all of the receive antennas 28 as in equation (25) (step 52). Finally, receiver 6 splits the MRC output block into Ng groups (step 64), and implements a scheme, e.g., ML or Sphere, to decode each reduced size group as in equation (26) to provide the estimated data (step 56). The performance of system 2 is a function of the sub-block size Nsub, and when Nsub≧rh, the maximum diversity order in (12) is achieved. However, when Nsub<rh, the achieved diversity order is Nsub.

[0094] The diversity gain for the STDO techniques described herein can be summarized in the following proposition:

[0095] Proposition 2 The maximum achievable STDO diversity order Gd=rh is guaranteed by our DPS design when the group size is selected as Nsub≧Nt(Q+1). When the channel correlation matrix Rh has full rank rh=NrNt(Q+1), our DPS design enables also the maximum possible coding gain among all linearly coded transmissions that is given in closed form by: 30$G_c^{\max }={\left(\det \left(R_h\right)\right)}^{\stackrel{\perp }{r_h}}{d}_{\min }^2/N_t.$

[0096] Transmission rate 1 symbol/sec/Hz is achieved by DPS design.

[0097] In fact, the group size Nsub controls the tradeoff between performance and decoding complexity. When Nsub≦Nt(Q+1), as Nsub decreases, the decoding complexity decreases, while at the same time, the diversity order decreases. By adjusting Nsub, we can balance the affordable complexity with the required performance.

[0098] The described DPS scheme utilizes channel knowledge at the receiver. As described DPS converts the multi-input single output (MISO) channel corresponding to each receive-antenna to an equivalent single-input single-output (SISO) system. As a result, existing SISO training algorithms can be utilized with the described techniques. The following describes a PSAM training method that may be used with the described GLCP-DPS transmission.

[0099] Initially, the BEM coefficients are estimated per block of N symbols. Starting from (19), we select one group sg to serve as the pilot group. Without loss of generality, we select this group to be the first one, and design it as:

s0=[b, b, . . . , b]Nsub×1T.  (27)

[0100] After precoding the remaining groups as before, and interleaving among all the groups, the transmitted block u becomes:

u=[b,[u1]0, . . . , [uNg−1]0;b, . . . ; b,[u1]Nsub−1, . . . , [uNg−1]Nsub−1]T.  (28)

[0101] At the receive filter output, samples are collected that depend only on the pilot symbols. The resulting input-output relationship can be expressed as: 31$\begin{array}{cc}{x}_{v,b}=\frac{b}{\sqrt{N_t}}{D}_{H,b}^{\left(v\right)}1+{\zeta }_{v,b},\forall v\in \left[1,N_r\right],& \left(29\right)\end{array}$

[0102] where DH,b(ν) contains the channel entries from DH(ν) that correspond to the pilot slots, 1 is a vector with all-ones, and ζν,b is the corresponding noise vector. Recalling (18), it can be verified that 32$\begin{array}{cc}{D}_{H,b}^{\left(v\right)}1=\sum _{q=0}^{N_t\left(Q+1\right)-1}{h}_q^{\left(v\right)}{\omega }_{q,b}={\Omega }_b{h}^{\left(v\right)},& \left(30\right)\end{array}$

[0103] where Ωb :=[w0,b . . . wNt(Q+1)−1,b], and wq,b :=[1, exp(jwqNg), . . . , exp(jwqNg(Nsub−1))]T. Substituting (30) into (29), we obtain 33$\begin{array}{cc}{x}_{v,b}=\frac{b}{\sqrt{N_t}}{\Omega }_b{h}^{\left(v\right)}+{\zeta }_{v,b},\forall v\in \left[1,N_r\right].& \left(31\right)\end{array}$

[0104] The linear minimum mean square error (LMMSE) channel estimator for the νth receive-antenna is: 34$\begin{array}{cc}{\hat{h}}^{\left(v\right)}=\frac{b^{*}}{{\sigma }_{Ϛ}^2\sqrt{N_t}}{\left(R_{h^{\left(v\right)}}^{-1}+\frac{{\&LeftBracketingBar;b\&RightBracketingBar;}^2}{{\sigma }_{Ϛ}^2{N}_t}{\Omega }_b^{\mathscr{H}}{\Omega }_b\right)}^{-1}{\Omega }_b^{\mathscr{H}}{x}_{v,b},\forall v\in \left[1,N_r\right],& \left(32\right)\end{array}$

[0105] and the mean square error of ĥ(ν) is 35$\begin{array}{cc}{\sigma }_{{\hat{h}}^{\left(v\right)}}^2=\mathrm{tr}\left({\left(R_{h^{\left(v\right)}}^{-1}+\frac{{\&LeftBracketingBar;b\&RightBracketingBar;}^2}{{\sigma }_{\zeta }^2{N}_t}{\Omega }_b^{\mathscr{H}}{\Omega }_b\right)}^{-\mathrm{1`}}\right).& \left(33\right)\end{array}$

[0106] When Nsub≧Nt(Q+1), thanks to the equi-spaced placement of pilot symbols, we have ΩbHΩb=NsubINt(Q+1). Suppose we fix the total power allocated to pilot symbols; i.e., |b|2=Pb/Nsub, and Pb is fixed. Then (33) can be simplified to 36$\begin{array}{cc}{\sigma }_{{\hat{h}}^{\left(v\right)}}^2=\mathrm{tr}\left({\left(R_{h^{\left(v\right)}}^{-1}+\frac{{𝓅}_b}{{\sigma }_{\zeta }^2{N}_t}{I}_{N_t\left(Q+1\right)}\right)}^{-\mathrm{1`}}\right).& \left(34\right)\end{array}$

[0107] Eq. (34) confirms that the more power we allocate to pilot symbols, the more reliable our channel estimator is. For a fixed power Pb, as the number of transmit-antennas increases, σh(ν)2 increases. Eqs. (32) and (34) provide a reliable training-based channel estimator and its variance, that is needed for realizing our DPS-based STDO scheme. Optimizing the training design also from a capacity perspective will be one of our future topics.

[0108] In the described STDO coding scheme, inner STDO coders 22 and inner STDO decoders 32 transform the time-selective channels into frequency-selective channels, e.g., by means of FFT and IFFT operations. The middle and outer coders 17, 16, respectively, can be used to implement orthogonal space-time multipath (STDO) designs to achieve the maximum diversity, and large coding gains. The following describes the duality between our finite basis expansion model for time-selective channels, and the finite impulse response (FIR) tapped delay model for frequency-selective channels, as further described in U.S. patent application Ser. No. ______, entitled “SPACE-TIME MULTIPATH CODING SCHEMES FOR WIRELESS COMMUNICATION SYSTEMS,” filed Apr. 21, 2003, having Attorney Docket No. 1008-007US01, the entire content of which is incorporated herein by reference.

[0109] Because circulant matrices can be diagonalized by (I)FFT matrices, and that the BEM described in (4) is based on the FFT grid, we can rewrite DH(ν,μ) as 37$\begin{array}{cc}{D}_H^{\left(v,\mu \right)}=\sum _{q=0}^Q{h}_q^{\left(v,\mu \right)}{D}_q=F_N{H}^{\left(v,\mu \right)}{F}_N^{\mathscr{H}},& \left(35\right)\end{array}$

[0110] where H(ν,μ) is a circulant N×N matrix with first column 38${\left[h_{Q/2}^{\left(v,\mu \right)}\cdots h_0^{\left(v,\mu \right)}0\cdots 0h_Q^{\left(v,\mu \right)}\cdots h_{Q/2+1}^{\left(v,\mu \right)}\right]}^T,$

[0111] and FN denotes the N-point Fast Fourier Transformation (FFT) matrix with (m+1, n+1)st entry [FN]m,n=1/{square root}{square root over (N)})e−j2πmn/N. If we now design the inner STDO codec in FIG. 1 as

Φμ=FN, ∀μ∈[1, Nt], and Γν=FNH, ∀ν∈[1, Nr],  (36)

[0112] then based on (3), (35) and (36), we obtain 39$\begin{array}{cc}\begin{array}{c}{y}_v=F_N^{\mathscr{H}}{x}_v=\sum _{\mu =1}^{N_t}{F}_N^{\mathscr{H}}{D}_H^{\left(v,\mu \right)}{F}_N{c}_{\mu }+F_N^{\mathscr{H}}{\zeta }_v\\ =\sum _{\mu =1}^{N_t}{H}^{\left(v,\mu \right)}{c}_{\mu }+{\eta }_v,\forall v\in \left[1,N_r\right].\end{array}& \left(37\right)\end{array}$

[0113] Equations (35)-(37) suggest that due to the described BEM, it is possible to convert the diagonal time-selective channel DH(ν,μ) to a circulant matrix after IFFT and FFT operations. The Q+1 BEM coefficients are dual to the channel taps of a frequency-selective channel. Hence, the inner STDO codec of system 2 is capable of transforming multi-antenna ST time-selective channels into ST frequency-selective channels. This time-frequency duality on channel models can be summarized in the following:

[0114] Property 2: Based on the BEM in (1) for time-selective channels, a block transmission over time-selective channels can be equivalently viewed as a transmission over frequency-selective channels after FFT processing at the transmitter, and IFFT processing at the receiver. An equivalent diagonal (or circulant) channel matrix can be generated either from a time-selective, or, from a frequency-selective channel with appropriate (I)FFT operations.

[0115] In order to achieve the maximum diversity gain rh, we can adopt some of the existing Space-Time-Multipath (STM) codecs as our middle and outer STDO codecs. In the following, a low-complexity CP-based approach, and a high-performance Zero Padding (ZP) based approach.

[0116] In the CP-based approach, the middle STDO codec consists of two stages. The first stage implements ST block coding, that is used to collect the spatial diversity. The second stage implements a GLCP-OFDM-based module to collect the Doppler (now viewed as multipath) diversity.

[0117] The ST block coding stage comprises an extension of the generalized complex orthogonal design (GCOD) for flat channels to time-selective channels. Consider splitting u into Ns equally long sub-blocks of size Nsub as the input of GCOD; i.e., K=NsNsub. Define the size of the output of GCOD as: NdNsub. Therefore, the rate of ST block code is Ns/Nd. Out ST block code matrix is: 40$\begin{array}{cc}{C}_{\left(N_d{N}_{\mathrm{sub}}\right)\times N_t}=\sum _{i=0}^{N_s-1}\left(A_i\otimes u_i+B_i\otimes u_i^{*}\right),& \left(38\right)\end{array}$

[0118] where ui :=[[u]Nsubi, . . . , [u]Nsub(i+1)−1]T, and the real matrices {Ai ∈ Nd×Nt, Bi ∈ Nd×Nt, ∀i ∈ [0, Ns−1]}, satisfy the following properties: 41$\begin{array}{cc}\begin{array}{c}{A}_i^T{A}_{i^{\prime }}+B_i^T{B}_{i^{\prime }}=\frac{1}{N_t}{I}_{N_t}\delta \left(i-i^{\prime }\right)\\ A_i^T{B}_{i^{\prime }}=0,\forall i,i^{\prime }\in \left[0,N_s-1\right].\end{array}& \left(39\right)\end{array}$

[0119] The symbols of the μth column of C are directed to the μth transmit-antenna.

[0120] Thanks to the FFT inner codec, the time-selective channel is converted to a frequency-selective channel. Dealing now with a frequency-selective channel, the second stage of our middle STDO codec forms an OFDM-based block transmission. The advantage of invoking an OFDM-structure here is two-fold: i) to cope with interblock interference (IBI) so that ST block codes can be decoded in block-form; and ii) to diagonalize the channel so that out low-complexity GLCP-based codec can be used in the outer STDO encoder.

[0121] After the first stage of ST block coding, an IFFT is performed and the CP is added, to each sub-block with length Nsub. In matrix form, these operations can be described as:

[c1 . . . cNt]=ρ(INd{circle over (x)}Tcp)(INd{circle over (x)}FNsubH)C,  (40)

[0122] where 42$\rho =\sqrt{N_{\mathrm{sub}}/\left(N_{\mathrm{sub}}+Q\right)}$

[0123] is a power-normalizing constant, and Tcp :=[T1, INsub, T2]T, is a matrix implementing the CP insertion, with

T1 :=[0(Q/2)×(Nsub−Q/2), IQ/2]T, and T2 :=[IQ/2,0(Q/2)×(Nsub−Q/2)]T.

[0124] Correspondingly, at the receiver, we design the middle STDO decoder {overscore (M)}(·) following the reverse order of the two encoding stages. Specifically, we remove the CP and perform FFt by pre-multiplying with (INd{circle over (x)}FNsub)(INd{circle over (x)}Rcp) the received block on each antenna, where Rcp :=[0Nsub×(Q/2),INsub,0Nsub×(Q/2)] is a matrix description of the CP removal operation. Recalling (37), and our inner codec, we infer that the equivalent channel matrix facing the middle STDO codec is a circulant matrix H(ν,μ). With the OFDM module, the equivalent channel becomes: 43$\begin{array}{cc}\begin{array}{c}\left(I_{N_d}\otimes F_{N_{\mathrm{sub}}}\right)\left(I_{N_d}\otimes R_{\mathrm{cp}}\right)H^{\left(v,\mu \right)}\left(I_{N_d}\otimes T_{\mathrm{cp}}\right)\left(I_{N_d}\otimes F_{N_{\mathrm{sub}}}^{\mathscr{H}}\right)=I_{N_d}\otimes {\stackrel{\_}{D}}_H^{\left(v,\mu \right)},\\ \mathrm{where}{\stackrel{\_}{D}}_H^{\left(v,\mu \right)}:=\mathrm{diag}\left[H^{\left(v,\mu \right)}\left(0\right),\dots ,H^{\left(v,\mu \right)}\left(N_{\mathrm{sub}}-1\right)\right],\mathrm{and}{H}^{\left(v,\mu \right)}\left(m\right):=\sum _{q=0}^Q{h}_q^{\left(v,\mu \right)}{e}^{-j\frac{2\pi \eta }{N_{\mathrm{sub}}}m}.\end{array}& \left(41\right)\end{array}$

[0125] To decode the ST block code, we need to simplify our input-output relationship using (41). Based on (37) and (41), after CP removal and FFT processing, we obtain 44$\begin{array}{cc}\begin{array}{c}{\stackrel{\_}{y}}_v:=\left(I_{N_d}\otimes F_{N_{\mathrm{sub}}}\right)\left(I_{N_d}\otimes R_{\mathrm{cp}}\right)y_v\\ =\rho \sum _{\mu =1}^{N_t}{\left(I_{N_d}\otimes {\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right)\left[C\right]}_{\mu }+\left(I_{N_d}\otimes F_{N_{\mathrm{sub}}}\right)\left(I_{N_d}\otimes R_{\mathrm{cp}}\right){\eta }_v,\end{array}& \left(42\right)\end{array}$

[0126] where [C]μ stands for the μth column of C. Plugging [C]μ into (38), we rewrite {overscore (y)}ν as: 45$\begin{array}{cc}{\stackrel{\_}{y}}_v=\rho \sum _{i=1}^{N_s}\left(\sum _{\mu =1}^{N_t}{\left[A_i\right]}_{\mu }\otimes \left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}{u}_i\right)+\sum _{\mu =1}^{N_t}{\left[B_i\right]}_{\mu }\otimes \left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}{u}_i^{*}\right)\right)+{\stackrel{\_}{\eta }}_v,& \left(43\right)\end{array}$

[0127] where {overscore (η)}ν:=(INd{circle over (x)}FNsub)(INd{circle over (x)}Rep)ην, and [Ai]μ and [Bi]μ are the μth columns of Ai and Bi, respectively.

[0128] Similar to the DPS decoding scheme, MRC is utilized to combine the received blocks from different antennas. Based on the orthogonality of Ai's and Bi's in (39), for each receive antenna, our decoding matrix is: 46$\begin{array}{cc}{G}_i^{\left(v\right)}=\left[\sum _{\mu =1}^{N_t}{\left[A_i\right]}_{\mu }^T\otimes {\left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right)}^{*}\sum _{\mu =1}^{N_t}{\left[B_i\right]}_{\mu }^T\otimes {\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right],\forall i\in \left[0,N_s-1\right],v\in \left[1,N_r\right].& \left(44\right)\end{array}$

[0129] To implement the MRC, we use the combiner: 47$\begin{array}{cc}{G}_i=\sqrt{N_t}{\left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right)}^{*}\right)}^{-\frac{1}{2}}\left[G_i^{\left(1\right)},\dots ,G_i^{\left(N_r\right)}\right],\forall i\in \left[0,N_s-1\right].& \left(45\right)\end{array}$

[0130] Here, for the CP-based scheme only, we need to modify A5) as follows: 48${\mathrm{A5}}^{\prime })\mathrm{Channels}{\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\mathrm{are}\mathrm{coprime};i.e.,\det \left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right)}^{*}\right)\ne 0.$

[0131] Under A5′), we have GiGiH=I. At the receiver, the ith sub-block corresponding to ui is 49$\begin{array}{cc}\begin{array}{c}{z}_i={G_i\left[{\stackrel{\_}{y}}_1^T,{\stackrel{\_}{y}}_1^{\mathscr{H}},\cdots ,{\stackrel{\_}{y}}_{N_r}^T,{\stackrel{\_}{y}}_{N_r}^{\mathscr{H}}\right]}^T\\ ={\rho \left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\left({\stackrel{\_}{D}}_H^{\left(v,\mu \right)}\right)}^{*}\right)}^{\frac{1}{2}}{u}_i+{\xi }_i,\forall i\in \left[0,N_s-1\right],\end{array}& \left(46\right)\end{array}$

[0132] where ξi :=Gi[{overscore (η)}1T,{overscore (η)}1H, . . . , {overscore (η)}NrT,{overscore (η)}NrH]T is a circular AWGN vector.

[0133] The outer STDO encoder Θ is designed as

Θ=INa{circle over (x)}{overscore (Θ)}.  (47)

[0134] The ith sub-block is precoded by {overscore (Θ)}; i.e., ui={overscore (Θ)}si, ∀i ∈[0, Ns−1]. As in the DPS design, in order to reduce the decoding complexity, we again pursue the design of {overscore (Θ)} in a grouped form. Since the equivalent channel matrix between zi and si is diagonal, we can write the gth group of zi, defined as zi,g, as: 50$\begin{array}{cc}{z}_{i,g}=\frac{\rho }{\sqrt{N_t}}{\left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{D}}_{H,g}^{\left(v,\mu \right)}\left({\stackrel{\_}{D}}_{H,g}^{\left(v,\mu \right)}\right)}^{*}\right)}^{\frac{1}{2}}{\Theta }_{\mathrm{sub}}{s}_{i,g}+{\xi }_{i,g},\forall i\in \left[0,N_s-1\right],g\in \left[0,N_g-1\right],& \left(48\right)\end{array}$

[0135] where Θsub is an {overscore (N)}sub×{overscore (N)}sub matrix designed according to [24] with {overscore (N)}sub=Nsub/Ng. Again, ML decoding for si,g can be performed by using sphere-decoding with block size {overscore (N)}sub.

[0136] Based on (14), we can verify that out CP-based approach constitutes also a linearly coded transmission. When Nsub≧(Q+1), the maximum diversity order rh is guaranteed. Furthermore, when rh=NrNt(Q+1), and we select {overscore (N)}sub=Q+1, the coding gain for this CP-based scheme satisfies [c.f. (15)]: 51$\begin{array}{cc}\left(\ln 2\right){\left(\det \left(R_h\right)\right)}^{\frac{1}{r_h}}\frac{N_{\mathrm{sub}}{d}_{\min }^2}{\left(N_{\mathrm{sub}}+Q\right)N_t}\le G_c\le {\left(\det \left(R_h\right)\right)}^{\frac{1}{r_h}}\frac{N_{\mathrm{sub}}{d}_{\min }^2}{\left(N_{\mathrm{sub}}+Q\right)N_t},& \left(49\right)\end{array}$

[0137] where the upper bound is achieved when {overscore (N)}sub=Q+1 satisfies a certain algebraic property [24]. A proof for the diversity and coding gains for CP-based scheme is given in Appendix D. As Nsub increases (ρ→1), the coding gain of our CP-based design in (49) approaches the maximum coding gain in (15). The encoding and decoding processes of this CP-based approach are summarized in the block diagram of FIG. 4. And our results are collected in the following proposition:

[0138] Proposition 3 CP-based STDO block codes guarantee the maximum space-Doppler diversity Gd=rh, when {overscore (N)}sub≧Q+1, and offer low (FFT based) sphere decoding complexity at the receiver. When the channel correlation matrix Rh has full rank NtNr(Q+1), the CP-based design achieves the maximum coding gain Gcmax of linearly coded systems, asymptotically, as Nsub increases. The transmission rate of CP-based design is rstbcNsub/(Nsub+Q), where rstbe is the rate of block ST codes.

[0139] In the ZP-based approach, zero padding (ZP) replaces the CP guard. Similar to the CP-based design, there are two stages of the middle STDO codec. The first stage implements the GCOD, which is similar to (38), while the second eliminates IBI by padding zeros after each sub-block.

[0140] A scalar GCOD can be extended to the block-based GCOD: 52$\begin{array}{cc}{C}_{N_j{N}_{\mathrm{sub}}\times N_t}=\sum _{i=0}^{N_s-1}\left[A_i\otimes u_i+B_i\otimes \left(P_1{u}_i^{*}\right)\right],& \left(50\right)\end{array}$

[0141] where ui is defined as in (50); P1 is a time-reversal matrix with entries [P1]p,q=δ(p+q−Nsub−1); and the matrices {Ai ∈Nd×Nt, Bi ∈Nd×Nt, ∀i ∈[0, Ns−1]} are defined as in (39). As for the second stage of the middle STDO encoder, instead of inserting the CP as in Section V.A, we insert leading and trailing zeros in each sub-block. Based on the design of the inner codec in (36) and the middle STDO encoder, the input-output relationship from C to yν is: 53$\begin{array}{cc}{y}_v=\sum _{\mu =1}^{N_t}{H^{\left(v,\mu \right)}\left(I_{N_d}\otimes T_{\mathrm{zp}}\right)\left[C\right]}_{\mu }+{\eta }_v,& \left(51\right)\end{array}$

[0142] where Tzp :=[0Nsub×Q/2INsub 0Nsub×Q/2]T implements the ZP insertion. We can verify that

H(ν,μ)(INd{circle over (x)}Tzp)=INd{circle over (x)}({overscore (H)}(ν,μ)Tzp),  (52)

[0143] where the (Nsub+Q)×(Nsub+Q) circulant matrix {overscore (H)}(ν,μ) has the same structure as H(ν,μ). The outer STDO encoder Θ is selected here to be an identity matrix; i.e., u=s.

[0144] At the receiver, to decode the ST block code and combine the results from different receive antennas, we use the MRC matrix 54$\begin{array}{cc}{G}_i^{\left(v\right)}=\left[\sum _{\mu =1}^{N_t}{\left[A_i\right]}_{\mu }^T\otimes {\left({\stackrel{\_}{H}}^{\left(v,\mu \right)}\right)}^{\mathscr{H}}\sum _{\mu =1}^{N_t}{\left[B_i\right]}_{\mu }^T\otimes \left({\stackrel{\_}{H}}^{\left(v,\mu \right)}{P}_2\right)\right],\forall i\in \left[0,N_s-1\right],& \left(53\right)\end{array}$

[0145] where P2 is an (Nsub+Q)×(Nsub+Q) time reversal matrix. Two properties of time-reversal matrices will come handy at this point [26]:

[0146] P1) P2Tzp=TzpP1;

[0147] P2) P2=FNsub+QHFNsub+QH=FNsub+QFNsub+Q.

[0148] Using P1) and P2), we can verify by direct substitution that

P2({overscore (H)}(ν,μ)*TzpP1=FNsub+QHFNsub+QH({overscore (H)}(ν,μ))*FNsub+QFNsub+QTzp=({overscore (H)}(ν,μ))HTzp.  (54)

[0149] From the circulant structure of {overscore (H)}(ν,μ), it also follows that {overscore (H)}(ν,μ)({overscore (H)}(ν,μ))H=({overscore (H)}(ν,μ))H{overscore (H)}(ν,μ). Similar to A5) and A5′), before we define the matrix to combine the multiple receive antennas, we need the following assumption: 55${\mathrm{A5}}^{″})\mathrm{Channels}{H}^{\left(v,\mu \right)}\mathrm{are}\mathrm{coprime};i.e.,\det \left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{H}}^{\left(v,\mu \right)}\left({\stackrel{\_}{H}}^{\left(v,\mu \right)}\right)}^{*}\right)\ne 0.$

[0150] Based on this commutativity and (54), by defining 56$\begin{array}{cc}{G}_i=\sqrt{N_t}{\left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{H}}^{\left(v,\mu \right)}\left({\stackrel{\_}{H}}^{\left(v,\mu \right)}\right)}^{\mathscr{H}}\right)}^{-\frac{1}{2}}\left[G_i^{\left(1\right)},\dots ,G_i^{\left(N_r\right)}\right],\forall i\in \left[0,N_s-1\right],& \left(55\right)\end{array}$

[0151] and combining the multiple receive antennas, we obtain 57$\begin{array}{cc}\begin{array}{c}{z}_i={G_i\left[y_1^T,y_1^{\mathscr{H}},\dots ,y_{N_r}^T,y_{N_r}^{\mathscr{H}}\right]}^T\\ =\frac{1}{\sqrt{N_t}}{\left(\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{{\stackrel{\_}{H}}^{\left(v,\mu \right)}\left({\stackrel{\_}{H}}^{\left(v,\mu \right)}\right)}^{\mathscr{H}}\right)}^{\frac{1}{2}}{T}_{\mathrm{zp}}{s}_i+{\xi }_i,\forall i\in \left[0,N_s-1\right].\end{array}& \left(56\right)\end{array}$

[0152] To decode si from zi, (·) can again rely on sphere-decoding implemented on blocks of size Nsub.

[0153] The performance of the SP-based scheme can be analyzed. The Euclidean distance between distance between zi and z′i as: 58$\begin{array}{cc}{d}^2\left(z_i,z_i^{\prime }\right)=\frac{1}{N_t}\sum _{v=1}^{N_r}\sum _{\mu =1}^{N_t}{\&LeftDoubleBracketingBar;E_i{h}^{\left(v,\mu \right)}\&RightDoubleBracketingBar;}^2,& \left(57\right)\end{array}$

[0154] where Ei is a Toeplitz matrix generated by ei=si−s′i. When rh=NrNt(Q+1), we find that the coding gain in this is 59$\begin{array}{cc}{G}_c={\left(\det \left(R_h\right)\right)}^{\frac{1}{r_h}}\frac{d_{\min }^2}{N_t}.& \left(58\right)\end{array}$

[0155] The coding and decoding processes for the ZP-based scheme are summarized in FIG. 5.

[0156] Proposition 4 ZP-based STDO block codes guarantee the maximim space-Doppler diversity Gd=rh, ∀Nsub>0. When the channel correlation matrix Rh has full rank NtNr(Q+1), the ZP-based design achieves the maximum coding gain Gcmax of linearly coded systems. The transmission rate of ZP-based design is rstbcNsub/(Nsub+Q), where rstbc is the rate of the block ST codes.

[0157] Comparing our three STDO designs, we note that: i) all schemes guarantee the maximum diversity gain; ii) DPS and ZP-based schemes achieve also the maximum coding gain, while the CP-based scheme achieves the maximum coding gain asymptotically (as Nsub increases); iii) to guarantee the maximum diversity gain, the CP-based scheme provides the lowest decoding complexity; iv) to deal with IBI, CP- and ZP-based approaches rely on CP or ZP guards, which consume extra bandwidth compared with DPS scheme that does not require any guard. Furthermore, together with GCOD design benefits, our CP- and ZP-based STDO codecs inherit also its limitation in suffering up to 50% rate loss, when Nt>2 antennas are signaling with complex constellations. Notwithstanding, the DPS attains full rate for any Nt.

EXAMPLES

[0158] Test Case 1: To validate the BEM, we utilize the widely accepted Jakes' model, with a carrier frequency of f0=900 MHz, and a mobile speed νmax=96 km/hr. The transmitted block length is N=25. Two transmit-antennas (Nt=2) and one receive-antenna (Nr=1) is considered. Since we want to test the model matching and the diversity for different Q's, we select the symbol period Ts=Q/(2fmaxN). We generate all multiple channels using this Jakes's model. At the transmitter, we adopt our DPS design based on the knowledge of Q and Nt=2. At the receiver, we consider two different scenarios: one uses the parameters generated by Jakes' model (corresponding curves are marked by “Jakes” in FIG. 6); and the other one uses the BEM coefficients to approximate the Jakes' model. The channel generated by the Jakes' model has 200 paths, while BEM approximates Jakes' model by Q+1 BEM coefficients. MMSE equalization is used for both scenarios. FIG. 6 depicts the performance results. We observe that; i) our DPS design enables spatial diversity gains even for systems adhering to the Jakes' model; ii) when Q is smnall (Q=2), model mismatch between the BEM and the Jakes' model causes an error floor; and iii) when Q is large, the BEM matches the Jakes' model well, and the BER performance improves considerably.

[0159] Test Case 2 (comparisons among the three STDO codecs); We compare DPS, CP-Based, and ZP-Based schemes with Nt=2 transmit antennas, Q+1=3 bases per channel, and BEM parameters that are i.i.d., Gaussian, with mean zero, and variance 1/(Q+1). We choose QPSK modulation for all these schemes. The number of information symbols per block is P=K=24. For DPS, the transmitted block length N=24, while for CP- and ZP-based schemes, the block length N=28 because of the CP and ZP guards, respectively. The linear precoder with grouping is employed for DPS and CP-based schemes with group sizes Nsub=6 and {overscore (N)}sub=3, respectively. FIG. 7 depicts the BER performance of these three codecs. SD has been employed for all schemes. We observe that: i) from the slope of the BER curves for Nr=1, all three schemes guarantee the maximum diversity order Gdmax=Nt(Q+1)=6; ii) with either Nr=1 or 2, the ZP-based scheme exhibits the best performance among the three; iii) compared with CP, the performance of DPS incurs about 0.5 dB loss at high SNR for Nr=1; and iv) as Nr increases, the performance difference among three schemes diminishes at high SNR.

[0160] When Nt>2, because of the ST block codes, CP- and ZP-based schemes will lose up to 50% rate. In contrast, the DPS scheme attains the full rate for any Nt. FIG. 8 depicts the performance of our three STDO when Nt=4. For CP- and ZP-based schemes, we select the block ST code as in (38) which loses 50% rate. To maintain similar rates, we select QPSK for CP- and ZP-based schemes and BPSK for DPS with the same symbol power. The information block length is K=36. From FIG. 8, we observe that DPS outperforms both CP and ZP. Note that even in this case, CP- and ZP-based schemes have lower rate ((9/11) bit/sec/Hz) than DPS (1 bit/sec/Hz).

[0161] Test Case 3: In this example, we compare our DPS scheme with the smart-greedy code for (Nt, Nr)=(2,1). To maintain the same rate, we select BPSK for our DPS scheme. Each channel has Q+1=3 bases, and the channel coefficients are i.i.e. with mean zero and variance 1/(Q+1). First, we consider the uncoded setup. The information block length is K=P=30. The number of groups for DPS is Ng=5, so that these two schemes have comparable decoding complexity. FIG. 9 depicts the BER versus SNR comparison for the smart-greedy code, and our DPS (the solid lines). It is evident that DPS outperforms the “smart-greedy” coding because the former guarantees the full space-Doppler diversity.

[0162] Furthermore, we consider the coded case for both schemes. We select a (7,3) Reed-Solomon coder with block interleaving. The number of information bits is 90. Therefore, the length of the coded block of bits is 210. We select the depth of the block interleaver as 42. For the DPS design, we split the coded bits into 5 blocks. Each block is divided into 7 groups. The simulation results are shown in FIG. 9 (the dashed lines). Note that the DPS scheme still outperforms the smart-greedy codes remarkably.

[0163] Test Case 4 (correlated channels): In this example, we investigate the performance of our three schemes, when the channel coefficients are not i.i.d.. The carrier frequency is now f0=2 GHz, and the maximum mobile speed is νmax=160 km/hr. For these f0 and νmax values, we find that fmax≈296.3 Hz. The sampling period is defined as Ts=Q/(2fmaxN). We generate each channel correlation matrix E[h(ν,μ)(h(ν,μ))H]=UΛ, where U is a (Q+1)×(Qreal+1) unitary matrix, and Λ is a (Qreal+1) diagonal matrix. The ith entry of 60$\Lambda {\mathrm{is}\left[\Lambda \right]}_{i,i}={\gamma \left(\pi \sqrt{f_{\max }^2-{\left(\left(i-2Q_{\mathrm{real}}\right)/\left({\mathrm{KT}}_s\right)\right)}^2}\right)}^{-1},$

[0164] where γ is a normalizing factor. It is clear that the choice of entries of Λ reflects the underlying Doppler spectrum. We consider the channels corresponding to different antennas to be independent. Then we generate h(ν,μ)=UΛ{overscore (h)}(ν,μ), where the entries of {overscore (h)}(ν,μ) are i.i.d. with zero mean and unit variance. We consider Nt=2 and Nr=1 for all schemes. We select Q=4 and Qreal=2. For the CP-based scheme, we still use the GLCP method with group size {overscore (N)}sub=3, while for the DPS method, we select Nsub=6. The information block length for all schemes is K=36. Hence, for CP- and ZP-based schemes, the transmitted block length is N=44. We select BPSK modulation throughout this experiment. The rate for DPS and the smart-greedy code of [20] is 1 bit/sec/Hz, while the rate is (9/11) bits/sec/Hz for CP- and ZP-based schemes.

[0165] FIG. 10 depicts the BER performance for our three proposed schemes and the smart-greedy code. It can be seen that all our proposed schemes achieve full diversity (in this case it is Gd=6), and outperform the “smart-greedy” code. Compared with the CP-based and DPS schemes, the ZP-based scheme has about 1 dB gain at BER=10−4. However, note that both CP- and ZP-based schemes have lower rate than DPS.

[0166] Various embodiments of the invention have been described. We described a basis expansion model (BEM) to benchmark the performance of multi-antenna space-time coded transmissions over correlated time-selective fading MIMO channels. Specifically, we expressed in closed form the maximum achievable space-Doppler diversity gain in terms of the number of transmit-receive antennae and the number of bases. Furthermore, we quantified in closed-form the maximum possible coding gain for all linearly coded space-time transmissions, and found it to depend on the rank of the BEM coefficients' correlation matrix and the minimum Euclidean distance of the constellation used.

[0167] In addition to performance limits, the BEM enabled us to develop space-time-Doppler (STDO) coded designs capable of achieving (or approaching) these gains, using only knowledge of the maximum Doppler spread. We established two neat BEM properties that played an instrumental role in these designs: i) multiple (Nt) BEMs with Q+1 bases each, can be rendered mathematically equivalent to a single faster BEM with Nt(Q+1) bases, via a digital phase sweeping operation at the transmitters; and, ii) a BEM for time-selective channels is dual to a tapped delay line model for frequency-selective channels, which allows designs developed for one model to be used for the other, after incorporating appropriate FFT-based operations at the transmit-receive sides.

[0168] The first property led us to an STDO coded system based on a novel digital phase sweeping design, which collects the maximum joint space-Doppler diversity and large coding gains, while it facilitates application of SISO channel estimators, and affords a low-complexity modular implementation when working with linearly precoded small-size groups of symbols. Its unique feature is full rate (1 symbol/sec/Hz) operation, regardless of the constellation and the number of transmit-receive antennae. The second property showed us the way to adjust existing space-time coded designs maximizing space-multipath diversity over frequency-selective channels, to collect joint space-Doppler gains over our time-selective MIMO channel. Using the same property in the reverse direction, we established that the limits on coding gains we derived for the BEM, apply to space-time coded transmissions over frequency-selective MIMO channels as well. The multipath-inspired designs yielded space-time-Doppler coded block transmissions with cyclic prefix or zero padding guard intervals. The former system affords the lowest (FFT-based) complexity, while the latter exhibits the best performance. With two transmit-antennas they have full rate; but with more, they both suffer the same rate loss as space-time block orthogonal designs, with complex constellations.

[0169] All three designs were developed in a unifying framework that entails three-stages (outer-middle-inner) of encoding and decoding. Their relative strengths were delineated both analytically and with simulations that also compared them with an existing system. Both coded and uncoded transmissions were tested over i.i.d. and correlated channels, and confirmed that the proposed designs outperform existing alternatives as they exploit fully the joint space-Doppler diversity that becomes available with time-selective channels.

[0170] The described techniques can be embodied in a variety of receivers and transmitters including base stations, cell phones, laptop computers, handheld computing devices, personal digital assistants (PDA's), and the like. The devices may include a digital signal processor (DSP), field programmable gate array (FPGA), application specific integrated circuit (ASIC) or similar hardware, firmware and/or software for implementing the techniques. If implemented in software, a computer readable medium may store computer readable instructions, i.e., program code, that can be executed by a processor or DSP to carry out one of more of the techniques described above. For example, the computer readable medium may comprise random access memory (RAM), read-only memory (ROM), non-volatile random access memory (NVRAM), electrically erasable programmable read-only memory (EEPROM), flash memory, or the like. The computer readable medium may comprise computer readable instructions that when executed in a wireless communication device, cause the wireless communication device to carry out one or more of the techniques described herein. These and other embodiments are within the scope of the following claims.

Claims

1. A wireless communication device comprising: a first coder that linearly precodes a data stream to produce a precoded data stream; a second coder that processes the precoded data stream in accordance with a doppler spread of time-selective communication channels to produce a transformed data stream; and an antenna to output a waveform in accordance with the transformed data stream.

2. The wireless communication device of claim 1, wherein the second coder processes the precoded data stream in accordance with a Basis Expansion Model (BEM) for the channels to produce the transformed data stream.

3. The wireless communication device of claim 2, wherein the second coder samples the data stream to collect blocks of frequency-domain samples, processes the collected blocks of samples to transform the frequency-domain samples to time-domain samples for the transformed data stream.

4. The wireless communication device of claim 3, wherein the second coder collects blocks of Q+1 frequency-domain samples, wherein Q is a selected as a function of the doppler spread of the channels.

5. The wireless communication device of claim 3, further comprising a modulator to process blocks of N symbols within the transformed data stream at a symbol period Ts to produce the waveform for transmission with the antennas, and wherein the second coder samples the precoded data stream with a period of 1/N*Ts to collect the samples.

6. The wireless communication device of claim 5, wherein Q=2 * ┌fmax*Ts*N┐, where fmax represents the doppler spread of the channels.

7. The wireless communication device of claim 5, further comprising a power splitter to produce a plurality of mirrored data streams from the transformed data stream; and wherein the antenna comprises one of a plurality of antennas to output waveforms in accordance with the mirrored data streams.

8. The wireless communication device of claim 7, further comprising a digital phase sweeping module that computes a single channel vector from estimates of doppler offset for the channels, and process the mirrored data streams with the single channel vector to shift a basis of each of the channels so that the basis become consecutive within a Fast Fourier Transform (FFT) grid.

9. The wireless communication device of claim 7, further comprising a modulator to modulate each mirrored data stream and produce the waveforms for transmission with the respective antennas.

10. The wireless communication device of claim 1, the first coder comprises a linear constellation precoder.

11. The wireless communication device of claim 1, wherein the first coder applies a unitary matrix to blocks of M symbols of the data stream.

12. The wireless communication device of claim 1, wherein the waveform is a multi-carrier waveform.

13. The wireless communication device of claim 1, wherein the waveform is single-carrier waveform.

14. The wireless communication device of claim 1, wherein the first coder linearly precodes the data stream by applying a matrix to symbols blocks of the data stream, wherein the matrix has a size that is an integer function of the number of antennas and the doppler spread.

15. The wireless communication device of claim 14, wherein the blocks include N symbols, where N is selected as a function of the number of antennas Nt and the doppler spread of the channels.

16. The wireless communication device of claim 14, wherein the first coder divides each of the symbol blocks of the data stream into Ng groups, where each group has Nsub symbols, and Nsub is an integer function of the number of antennas and the doppler spread.

17. The wireless communication device of claim 16, wherein the matrix applied by the linear precoder has a size of Nsub×Nsub.

18. The wireless communication device of claim 1, wherein the wireless communication device comprises one of a base station and a mobile device.

19. A wireless communication device comprising: a demodulator that receives a waveform from a plurality of time-selective multipath communication channels and produces a demodulated data stream; and a decoder that decodes the demodulated data in accordance with a Basis Expansion Model (BEM) that parameterizes time-selective communication multi-path channels as a function of frequency-selective multi-path communication channels.

20. The wireless communication device of claim 19, wherein the decoder applies an N×N matrix to the demodulated data, wherein the N is selected as a function of a doppler spread of the frequency-selective multi-path communication channels.;

21. A method comprising: estimating a doppler spread for a plurality of channels from a transmitter to a receiver; applying a linear precoder to a data stream to form a precoded data stream; processing the precoded data stream in accordance with the doppler spread to produce a transformed data stream; and transmitting a waveform with an antenna in accordance with the transformed data stream.

22. The method of claim 21, wherein processing the precoded data stream comprise processing the precoded data stream in accordance with a Basis Expansion Model (BEM) for the channels to produce the transformed data stream.

23. The method of claim 21, wherein processing the precoded data stream comprises: sampling the data stream to collect blocks of frequency-domain samples; processing the collected blocks of samples to transform the frequency-domain samples to time-domain samples; and outputting the transformed data stream based on the time-domain samples.

24. The method of claim 23, wherein sampling the precoded data stream comprises collecting blocks of Q+1 frequency-domain samples, wherein Q is a selected as a function of the doppler spread of the channels.

25. The method of claim 23, further comprising modulating blocks of N symbols within the transformed data stream at a symbol period Ts to produce the waveform for transmission with the antennas.

26. The method of claim 25, wherein sampling the precoded data stream comprises sampling the precoded data stream with a period of 1/N*Ts to collect the samples.

27. The method of claim 25, wherein Q=2 * [fmax*Ts*N], where fmax represents the doppler spread of the channels.

28. The method of claim 21, further comprising: splitting the power of the transformed data stream to produce a plurality of mirrored data streams; and transmitting the mirrored data stream with respective antennas.

29. The method of claim 28, further comprising: computing a single channel vector from the estimated doppler offset; and processing the mirrored data streams with the single channel vector to shift a basis of each of the channels so that basis become consecutive within a Fast Fourier Transform (FFT) grid.

30. The method of claim 21, wherein applying a linear precoder comprises applying a linear constellation precoder.

31. The method of claim 21, wherein the waveform is a multi-carrier waveform.

32. The method of claim 21, wherein the waveform is a single-carrier waveform.

33. The method of claim 21, wherein applying a linear precoder comprises: selecting a block size N as a function of the number of antennas Nt and the doppler spread; and applying a matrix to symbol blocks of the data stream, wherein the matrix has a size that is selected as a function of the number of antennas and the doppler spread.

34. The method of claim 33, wherein applying a linear precoder comprises: dividing each of the blocks of the data stream into Ng groups, where each group has Nsub symbols, and Nsub is selected as a function of the number of antennas and the doppler spread; and applying the matrix to each of the symbol groups within the blocks, wherein the matrix has a dimension that is a function of the number of symbols Nsub within the groups.

35. The method of claim 34, wherein the size of the matrix is Nsub×Nsub.

36. A computer-readable medium comprising instructions to cause a programmable processor of a wireless communication device to: estimate a doppler spread for a plurality of channels from a transmitter to a receiver; apply a linear precoder to a data stream to form a precoded data stream; process the precoded data stream in accordance with the doppler spread to produce a transformed data stream; and transmit a waveform with an antenna in accordance with the transformed data stream.

37. The computer-readable medium of claim 36, further comprising instructions to cause the programmable processor to process the precoded data stream comprise processing the precoded data stream in accordance with a Basis Expansion Model (BEM) for the channels to produce the transformed data stream.

38. The computer-readable medium of claim 36, further comprising instructions to cause the programmable processor to: sample the data stream to collect blocks of frequency-domain samples; process the collected blocks of samples to transform the frequency-domain samples to time-domain samples; and output the transformed data stream based on the time-domain samples.