Patent Grant
12107495
This invention relates to power converters, and in particular, to dc-dc power converters.
It is known in the art that electrical devices require electric power to operate. However, some electrical devices are more omnivorous than others. For example, a tungsten filament light bulb will operate over a wide range of voltages. Although it may be dim at low voltages, and although it may burn out prematurely at high voltages, it does not simply stop operating.
Digital circuits, however, are more finicky in their requirements. A digital circuit demands power with particular characteristics. A processor that receives power falling short of these characteristics will not just compute more slowly. It will simply shut down.
Unfortunately, power is not always delivered in a form that a microprocessor-based system will find acceptable. For example, in a handheld device, the battery voltage ranges from fully charged to almost zero. Thus, most such systems require something that accepts power in raw form and delivers it to the system in a form that the system finds more palatable.
This critical but unglamorous task falls upon the power converter.
A variety of power converters are known. These include power converters described in U.S. Pat. Nos. 8,860,396, 8,743,553, 8,723,491, 8,503,203, 8,693,224, 8,724,353, 8,339,184, 8,619,445, 8,817,501, U.S. Patent Publ. 2015/0077175, and U.S. Pat. No. 9,041,459.
The contents of all the foregoing patents are herein incorporated by reference.
In one aspect, the invention features an apparatus for providing electric power to a load. Such an apparatus includes a power converter that accepts electric power in a first form and provides electric power in a second form. The power converter includes a control system, and first and second stages in series. The first stage accepts electric power in the first form. The control system controls operation of the first and second stages. The first stage is either a switching network or a regulating network. The second stage is a regulating network when the first stage is a switching network. On the other hand, the second stage is a switching network when the first stage is a regulating network.
Among the embodiments are those in which the control system controls at least in part based on a voltage measured between the first and second stages.
Also among the embodiments are those in which the first stage is a regulating network, those in which the first stage is a switching network, and those in which it is the second stage that is a switching network, such as a cascade multiplier. In either case, the switching network can be a cascade multiplier.
In some embodiments, at least one of the stages includes a switching network having first and second terminals. Among these are embodiments in which these terminals are isolated, embodiments in which they have a common ground, and embodiments in which they have separate grounds.
In other embodiments, at least one of the stages includes a switching network having a first and second switching circuits, each of which has first and second terminals. In these embodiments, the first terminal of the second switching circuit connects to the second terminal of the first switching circuit. Among these are embodiments in which the two switching circuits have different voltage-transformation ratios, and embodiments in which they have the same voltage-transformation ratios.
In some embodiments, the switching network includes first and second switching circuits in series, whereas in others, it includes first and second switching circuits in series-parallel.
Some embodiments of the power converter further include a third stage in series with the first stage and the second stage so that the second stage is between the first stage and the third stage. These embodiments include those in which the first and third stages are switching networks, those in which the first and third stages are regulating networks, and those in which the third stage is operated with a duty cycle that causes the third stage to become a magnetic filter.
In some embodiments, the switching network includes a cascade multiplier. Among these are those embodiments in which the cascade multiplier is a single-phase cascade multiplier, those in which it is asymmetric, those in which it is a step-down, multiplier, and any those in which it is any combination thereof. Also among these embodiments are those in which the cascade multiplier is a dual-phase cascade multiplier. In this case, the cascade multiplier could be a symmetric cascade multiplier, or one that includes parallel pumped capacitors, or one that lacks DC capacitors.
In some of the foregoing embodiments, the cascade multiplier creates an auxiliary voltage to drive an additional circuit. Among these are embodiments that include a level shifter connected to be driven by the auxiliary voltage, and those in which a gate driver is connected to be driven by the auxiliary voltage.
In some embodiments, the switching network includes first and second dual-phase cascade multipliers, and a phase node shared by both cascade multipliers. In these embodiments, the first cascade multiplier, which is stacked on the second, is asynchronous and the second cascade multiplier is synchronous. Among these embodiments are those in which the first and second cascade multipliers operate at the same frequency, and those in which the first and second cascade multipliers operate at different frequencies.
In some embodiments, the regulating network includes a buck converter. Among these are embodiments in which the buck converter includes first and second terminals with the same reference voltage. Examples include those in which the buck converter's first and second terminals are at different reference voltages, those in which the buck converter has three terminals, and those in which the buck converter has a floating node at a floating voltage. In embodiments that have a floating node, the floating node can be between two loads or between two sources.
A variety of other regulating networks are contemplated. These include a buck-boost converter, a boost converter, and even a four-terminal non-inverting buck-boost converter.
In some embodiments that use a boost converter as a regulating network, the switching network includes a step-down single-phase asymmetric cascade multiplier. In some of these embodiments, selection switches connected to the regulating network cause the switching network to output a fraction of its normal output voltage. In others, switches are oriented so that cathodes of parasitic diodes corresponding to the switches connect to each other. Among these embodiments are those in which the first stage is a regulating network.
Embodiments include those in which those in which the regulating network regulates plural wires, those in which it regulates at most one wire, and those in which it regulates a particular one of plural wires based on an input voltage to the regulating network.
Also among the embodiments are those in which the regulating network has plural output ports and those in which it is a multi-tap boost converter. Among these are embodiments in which the switching network includes a single-phase step-down switched-capacitor circuit.
In yet other embodiments, the power converter floats above ground.
In some embodiments, the switching network is reconfigurable. In other embodiments, it is the regulating network that is reconfigurable. In yet others, both are reconfigurable. In either case, there are embodiments in which a magnetic filter connects to whichever of the two are reconfigurable. Thus, a magnetic filter could be connected to either the reconfigurable switching network or the reconfigurable regulating network.
In some embodiments, the switching network includes a dual-phase switched capacitor circuit. Among these are embodiments in which the switched capacitor circuit includes pump capacitors in series and DC capacitors in series.
In some embodiments, the switching network includes a dual-phase switching circuit including DC capacitors that store charge from the regulating network only during a dead-time transition during which the switching network is between states.
In yet other embodiments, the regulating network includes an inductor that promotes adiabatic charge transfer within the switching network.
Some embodiments also include a magnetic filter connected to the switching network to promote adiabatic charge transfer within the switching network. Among these are embodiments in which the magnetic filter is connected between the switching network and a load, those in which the magnetic filter is connected between the switching network and a source, and those in which the regulating network and the magnetic filter cooperate to promote adiabatic charge transfer within the switching network.
Embodiments further include those that have a circuit connected to the switching network to constrain current flow out of the switching network, and those that have a circuit connected to the switching network to promote adiabatic charge transfer within the switching network.
In some embodiments, the switching network includes a two-phase step-down switching network and the regulating network is a step-down network. Among these are embodiments in which the switching network includes a cascade multiplier. In those embodiments that include a cascade multiplier, the regulating network can include a buck converter. Also among these embodiments are those in which regulating network promotes adiabatic charge transfer.
In still other embodiments, the switching network includes a step-down single-phase asymmetric cascade multiplier and the regulating network includes a converter that causes voltage to step down. In some of these embodiments, it is the first stage that is a switching network.
In some embodiments, the regulating network includes a multiple-tap buck converter configured to have two operating modes. Among these are embodiments in which the switching network provides first and second voltage rails that, in operation, are maintained at different voltages.
Yet other embodiments are those in which the regulating network includes a buck converter having multiple taps and configured have three operating modes. Among these are embodiments in which the switching network provides first, second, and third voltage rails that, in operation, are maintained at different voltages.
Other embodiments of the apparatus are those in which the switching network includes a two-phase switched-capacitor circuit and the regulating network is a buck converter.
Also among the embodiments are those in which the regulating network includes parallel first and second regulating circuits.
In some embodiments, the power converter includes first and second outputs. In operation, the first output and second outputs being maintained at corresponding first and second voltage differences. The first voltage different is a difference between a first voltage and a second voltage, and the second voltage difference is a difference between a third voltage and the second voltage.
In some embodiments, the regulating network includes first, second, and third regulating circuits in parallel.
In other embodiments, the power converter includes first second, and third outputs. In operation, the first, second, and third outputs are maintained at corresponding first second, and third voltage differences. The first voltage different is a difference between a first voltage and a second voltage. The second voltage difference is a difference between a third voltage and the second voltage. And the third voltage difference is a difference between a fourth voltage and the second voltage.
In some embodiments, the power converter has a first terminal and a second terminal such that, in operation, a first voltage different is maintained across the first terminal and a second voltage difference is maintained across the second terminal. The first voltage difference is a difference between a first voltage and a second voltage, and the second voltage difference is a difference between a third voltage and the second voltage, with the second voltage being variable. Some of these embodiments also have a third stage that provides the second voltage. Also among these are embodiments in which the third stage includes a switched-mode power converter, a switched capacitor converter, a buck converter, or a cascade multiplier.
In some embodiments, the power converter is configured to provide AC output with a non-zero DC offset.
In other embodiments, the switching network includes a reconfigurable asynchronous cascade multiplier, and the regulating network is connected to the switching network to enable the switching network to cause either a step up in voltage or a step down in voltage. In some cases, the regulating network includes a four-switch buck-boost converter.
In still other embodiments, the first stage is a switching network that includes a reconfigurable cascade multiplier that operates synchronously in a single-phase, and the regulating network includes a four switch buck boost converter. Among these are embodiments in which the regulating network connects to the switching network at a point that enables the switching network to step voltage up or step voltage down.
Embodiments also include those in which the switching network includes a cascade multiplier with a charge pump embedded therein. The charge pump can have a variety of characteristics. For example, the charge pump can be reconfigurable, or it can be a fractional charge pump. Alternatively, the embedded charge pump operates in multiple modes, each of which corresponds to a voltage transformation ratio. Or the cascade multiplier might include a reconfigurable two-phase asynchronous step-down cascade multiplier. In any of these embodiments, the regulating network could include a two-phase boost converter.
In still other embodiments, the power converter further includes a third stage in series with the first stage and the second stage, wherein the second stage is between the first stage and the third stage, both of which are switching networks. The regulating circuit includes a buck converter, and both switching networks include a single-phase asynchronous step-up cascade multiplier. These embodiments include those in that further include a stabilizing capacitor at an output of the regulating network.
In still other embodiments, the power converter further includes a third stage in series with the first stage and the second stage, with the second stage being between the first stage and the third stage. In these embodiments, the first stage and the third stage are switching networks, the regulating circuit includes a buck-boost converter, the first switching network includes a single-phase asynchronous step-up cascade multiplier, and the second switching network includes a single-phase synchronous step-up cascade multiplier. Among these embodiments are those that also have a stabilizing capacitor at an output of the regulating network.
In some embodiments, the power converter further includes a third stage in series with the first and second stage, with the second stage being between the first stage and the third stage. The first and third stage are both regulating networks. However, the first stage includes a boost converter, and the third stage includes a buck converter. The switching network includes first and second cascade multipliers having equal numbers of stages. Some of these embodiments also have a phase pump shared by the first and second cascade multipliers. In others, the first and second cascade multipliers operate 180 degrees out of phase. And in yet others, the cascade multipliers comprise corresponding first and second switch stacks, and an output of the switching network is a voltage difference between a top of the first switch stack and a top of the second switch stack.
In some embodiments, the power converter further includes a third stage in series with the first and second stages, with the second stage being between the first stage and the third stage. In these embodiments, the first stage and the third stage are regulating networks, the first stage includes a three-level boost converter, the third stage includes a buck converter, and the switching network includes first and second cascade multipliers having unequal numbers of stages.
In other embodiments, the switching network receives current that has a first portion and a second portion, wherein the first portion comes from the regulating network, and the second portion, which is greater than the first, bypasses the regulating network.
In some embodiments, the power converter further includes a third stage in series with the first stage and the second stage, the second stage being between the first stage and the third stage. The first stage is a first regulating network, the third stage is a second regulating network, and the first stage includes a boost converter. The third stage includes a buck converter. The switching network includes cascade multipliers having unequal numbers of stages. Among these embodiments are those in which the second stage includes an additional inductor connected to the first stage.
Yet other embodiments include a third stage. In these embodiments, the first stage includes a regulating network, the third stage includes a regulating network, the power converter provides a load with a first voltage difference, the first stage provides a second voltage difference to the second stage, the second stage provides a third voltage difference to the third stage, the first voltage difference is a voltage difference between a first voltage and a second voltage, the second voltage difference is a voltage difference between a third voltage and a fourth voltage, the third voltage difference is a voltage difference between a fifth voltage and a sixth voltage, the fourth voltage differs from the second voltage, and the sixth voltage differs from the second voltage. Among these embodiments are those in which the second stage includes a reconfigurable switching network.
In some embodiments, the first stage includes a switching network having a reconfigurable dual phase cascade multiplier with an embedded inductor configured to promote adiabatic charge transfer between capacitors in the cascade multiplier. In some embodiments, the inductor is embedded at a location through which a constant current passes. Also among these embodiments are those in which the second stage includes a zeta converter, and those in which the cascade multiplier includes a phase pump with the inductor being embedded therein. In others of these embodiments, the cascade multiplier includes pump capacitors, and the inductor is embedded at a location that maximizes the number of paths that pass between the inductor and a pump capacitor.
In some embodiments, the switching network includes a dual-phase cascade multiplier with pump capacitors in series. Among these are embodiments in which the switching network has a variable transfer function and embodiments in which the switching network includes a phase pump that includes an embedded charge pump. In this latter case, the embedded charge pump includes switch sets, pump capacitors, and a controller that operates the switch sets to cause transitions between a first operating mode and a second operating mode, each of which corresponds to a transfer function for the cascade multiplier. Among these cases are those in which the controller operates the switch sets so that the embedded controller causes the cascade multiplier to have a transfer function in which the cascade multiplier provides either a voltage gain or a voltage attenuation.
In another aspect, the invention features an apparatus for providing electric power to a load includes a power converter that accepts electric power in a first form and provides electric power in a second form. The power converter includes a control system, a first stage, and a second stage in series. The first stage accepts electric power in the first form. The control system controls operation of the first and second stage. The first stage is either a switching network or a regulating network. The second stage is a regulating circuit when the first stage is a switching network, and a switching network otherwise.
These and other features will be apparent from the following detailed description and the accompanying figures, in which:
Each stage is either a regulating network 16A or a switching network 12A. The illustrated power converter separates the function of voltage/current transformation from that of regulation. As shown in
A power source 14 and a load 18A are shown only for clarity. These components are not actually part of the power converter. They merely represent the source of the power to be converted, and the ultimate consumer of that power. Dashed lines between these components and the power converter indicate that they are optional. Other components shown connected with dashed lines in this and other figures are likewise optional. For example, a dashed wire between the regulating network 16A and the switching network 12A is also optional.
In
The power source 14 need not deliver a constant stream of power. In fact, if it did, the power converter would not be nearly as necessary. After all, among the tasks of a power converter is to deliver a constant stream of power with specific characteristics to the load 18A notwithstanding variations in the power stream provided by the power source 14. The power source 14 is merely a source of power, or equivalently, since power is the time-derivative of energy, a source of energy.
The load 18A can be any type of electrical load. What is essential, is that it be a net energy consumer. Examples of a load 18A include a microprocessor, LED, RF PA, or a DSP. In fact, the load 18A might even be another power converter.
The arrows shown in the figure represent power flow, and not magnitude of power flow. Hence, each stage can be bidirectional. In such cases, if the load 18A supplies power, then the load 18A acts as a power source 14 and the power source 14 acts as a load 18A. However, in some embodiments, one or more stages are unidirectional. In addition, embodiments exist in which a stage can be a step-up stage, a step-down stage or a step-up/down stage.
The illustrated regulating network 16A can itself comprise two or more constituent regulating circuits operating as a combination in order to regulate some electrical parameter. These regulating circuits can have different voltage ratings and connect to each other in different ways. In some embodiments, the regulating circuits connect in series. In others, they connect in parallel, in series-parallel, or in parallel-series.
The regulating circuits that comprise a regulating network 16A can be of different types. For example, a regulating network 16A may comprise a buck converter in combination with a linear regulator. Examples of suitable regulating circuits include a buck converter, a boost converter, a buck-boost converter, a fly-back converter, a push-pull converter, a forward converter, a full bridge converter, a half-bridge converter, a multi-level converter (buck or boost), a resonant converter, a Cuk converter, a SEPIC converter, a Zeta converter, and a linear regulator.
Like the regulating network 16A, the switching network 12A can also be made of a combination of cooperating switching circuits. These individual switching circuits can have different transformation ratios, the same transformation ratios, and different voltage ratings. They can also be different kinds of switching circuits, such as series parallel or cascade multiplier circuits.
A cascade multipliers includes a switch stack, a phase pump, pump capacitors, and, optionally, dc capacitors. The phase pump comprises a pair of switches that cooperate to create a pump signal Vclk. In general, the two states of the clock are separated by a brief dead-time to allow transients and the like to decay. In cascade multipliers that require a complement to the clock signal, the phase pump includes another pair of switches to generate the complement. The switch stack is a series of switches connected between the input and the output of the cascade multiplier.
In those cases in which the switching circuit is a cascade multiplier, it can be asymmetric, symmetric, series-pumped, or parallel-pumped. Additional types of switching circuits include series-parallel switching circuits, parallel-series switching circuits, voltage-doubling circuits, and Fibonacci circuits. These constituent switching circuits can connect to each other in series, in parallel, in series-parallel, or in parallel-series. Some configurations of the illustrated power converter permit adiabatic charge transfer into or out of a capacitor in the switching network 12A.
Other configurations feature a reconfigurable switching network 12A that transitions between two or more states in the course of transferring charge. This charge transfer depends on the voltage across the capacitor's terminals. Reconfiguring the switching network 12A involves causing switches in the network to change state to cause this voltage to change. Reconfiguration can occur, for example, when the voltage or current transformation between the ports of the switching network 12A is to be changed.
In some embodiments, the controller receives multiple sensor inputs from the power converter and provides control signals along first and second paths P1, P2. Examples of sensor signals provided to the sensor inputs are VO, VX, VIN, IIN, IX, and IO. Among the foregoing sensor inputs, the negative terminals of VIN, VX, and VO can be at ground, above ground, or below ground depending upon their corresponding regulating circuits and switching networks. In fact, since voltage reflects a difference in potential energy between two points, there is nothing particularly special about ground.
The controller's function is to control both the regulating network 16A and the switching network 12A in an effort to control VIN, IIN, VO, and IO. In carrying this out, the controller can use either feed-forward control or feedback control. Feed-forward control involves choosing an output control signal based on an input, whereas feedback control involves choosing an output control signal based on an output.
Additional control methods that are applicable include voltage-mode control, current-mode control, hysteretic control, PFM control, pulse-skipping control, and ripple-based control. In embodiments that rely upon voltage-mode control, control can be linear or non-linear. In embodiments that rely upon current-mode control, current can be based on both an average value of current or a peak value of current.
It is possible to interconnect the regulating network and the switching network in a variety of ways.
In particular,
As mentioned above, it is possible for a switching network 12A to comprise two or more switching circuits.
The embodiment of
Conversely, the embodiment of
The four building blocks described above combine in various ways. For example, combining the building block in
The model shown in
The model shown in
The model shown in
During normal operation, the switching network 12A alternates between a first and second state at a specific frequency and duty cycle, such as 50%.
During the first state, the first switch 1 is closed and the second switch 2 is open. During the second state, the first switch 1 is open and the second switch 2 is closed. The frequency at which the first and second switches 1, 2 both transition between states can be the same as or different from that at which the regulating network 16A switches between states. In cases where these frequencies are the same, they can, but need not be in phase.
The pump capacitors C5-C7 swing up and down as the pump signal Vclk alternates between zero volts and the output voltage VO. At each clock cycle, charge moves between a pump capacitor C5-C7 and a dc capacitor C1-C4. In the particular embodiment shown, charge from a dc capacitor C1 makes its way to the last dc capacitor C4 after three clock cycles. Overall, the switching network 12A can be modeled using the circuit model shown in
In an alternative embodiment, shown in
A particular feature of the embodiment shown in
In operation, the first switch 1A and the second switch 2A are never in a closed state at the same time. To as great an extent as is possible, the first switch 1A is synchronized with a first switch set 1 so that when the first switch 1A is open, so are all the switches in the first switch set 1, and when the first switch 1A is closed, so are all the switches in the first switch set 1. Similarly, to as great an extent as is possible, the second switch 2A is synchronized with the second switch set 2 so that when the second switch 2A is open, so are all the switches in a second switch set 2, and when the second switch 2A is closed, so are all the switches in the second switch set 2. As a result, when the pump signal Vclk is high, a pump capacitor C8 connects to the first dc capacitor C2. When the pump signal Vclk is low, a pump capacitor C5 is connected to the first dc capacitor C2.
An advantage of the embodiment shown in
Yet another advantage is the possibility for using charge that is stored in the second dc capacitor C3 to supply a linear regulator, thus creating a regulated voltage.
In yet another embodiment, shown in
Because the nodes are shared in the illustrated embodiment, both cascade multipliers are operated at the same frequency. However, sharing the phase pump is not required.
Additionally, in the particular embodiment shown, the transformation ratio of each stage is relatively low. However, there is no special constraint on transformation ratio. For example, it would be quite possible for the dual-phase asymmetric cascade multiplier to have a transformation ratio of 2:1, while the dual-phase symmetric cascade multiplier has a transformation ratio of 10:1.
An advantage of the structure shown in
The cascade multiplier includes a phase pump 32.
The phase pump 32 shown in
In operation, the phase pump 32 operates in either a first operating mode or a second operating mode. In the first operating mode, the pump signal Vclk alternates between 0 volts and VO/2 volts. In the second operating mode, the pump signal Vclk alternates between 0 volts and VO volts.
Operation in the first mode requires that the phase pump 32 transition between four states according to the following switching pattern in Table 1A:
Operation in the second mode requires that the phase pump 32 transition between two states according to the following switching pattern in Table 1B:
Switching between the first and second mode enables the phase pump 32 to change the transfer function of the cascade multiplier. For example, if the phase pump 32 in
The phase pump 32 shown in
In normal operation, the phase pump 32 transitions between a first and a second state. During the first state, the switches in the first set of switches 1 are closed while those in the second set of switches 2 are opened. During the second state, the switches in the first set of switches 1 are opened while those in the second set of switches 2 are closed.
Unlike the phase pump 32 shown in
For example, if the phase pump 32 shown in
In
The buck converter includes a first switch S1, a second switch S2, an inductor L1, and a driver circuit 20A. The driver circuit 20A receives a control signal VR and outputs suitable voltages for controlling the first and second switches S1, S2.
It is not actually necessary for the input terminal and the output terminal to share a common negative terminal. In fact, there are six possible configurations of non-isolated regulating circuits that have two switches S1, S2, an inductor L1, and four terminals. Two of them are buck converters. These are shown in
Examples of configurations other than those described above for regulating circuits are shown in the appendix. All of the exemplary regulating circuits shown in the appendix can be used within the regulating network 16A.
The buck converters shown in
In operation, the buck converters in
Each buck-boost converter has first and second switches S1, S2, an inductor L1, and a driver circuit 20A that receives a control signal VR and, in response, outputs voltage signals that control the switches S1, S2. The buck-boost converters transition between first and second states. In the first state, the first switch S1 is closed and the second switch S2 is open. Conversely, in the second state, the first switch S1 is open and the second switch S2 is closed. In a power converter that uses this regulating network 16A, the voltage +V2 at the positive output terminal can be higher or lower than the voltage +V1 at the positive input terminal.
Each boost converter features first and second switches S1, S2, an inductor L1, and a drive circuit 20A that receives a control signal VR and, in response, outputs voltages suitable for driving the first and second switches S1, S2.
In operation, each boost converter transitions between first and second states. In the first state, the first switch S1 is closed and the second switch S2 is open. In contrast, in the second state, the first switch S1 is open and the second switch S2 is closed. In the regulating network 16A implemented using the boost converter of
The regulating network 16A can also be implemented using non-isolating regulating circuits that have a first switch S1, a second switch S2, a third switch S3, a fourth switch S4, an inductor L1, and four ports. A variety of configurations are shown in the table in the appendix.
When operating in buck mode, the first switch S1 is closed and the second switch S2 is opened. The remaining switches are then operated to transition between first and second states. In the first state, the third switch S3 is closed and the fourth switch S4 is open. In the second state, the third switch S3 is open and the fourth switch S4 is closed. In buck mode, +V2<+V1 while −V2<+V1.
When operating in boost mode, the third switch S3 is closed while the fourth switch S4 is open. The remaining switches are then operated to transition between first and second states. In the first state, the first switch S1 is closed and the second switch S2 is opened. In the second state, the first switch S1 is open and the second switch S2 is closed. When operated in boost mode, +V2>+V1 while −V2<+V1.
A converter along the lines shown in
A disadvantage of the power converters shown in
The power converter shown in
The illustrated switching network 12A is an example of a reconfigurable switched-capacitor network. There are many ways to implement such a reconfigurable switched-capacitor network. In fact, in principle, if one can add any number of switches, there are an infinite number of ways to implement such a reconfigurable switched-capacitor network.
The regulating network 16A includes first and second active switches 3, 4 and an inductor L1. The first and second active switches 3, 4 cycle between first and second states at a particular duty cycle and frequency.
Depending on the upon the required offset voltage Voff, which is set by input voltage VIN and output voltage VO, the offset voltage Voff can be set to a fraction of the output voltage VO by selectively enabling and disabling the selection switches S3-S10. In particular, the state of each selection switch for the various offset voltages Voff is shown in Table 2:
Some of the disabled switches (i.e., OFF) will see a higher voltage than the active switches in the regulating network 16A.
For example, suppose V4 equals one volt. Then V3 equals two volts, V2 equals three volts, and V4 equals four volts. When operating in the mode described on the first line of Table 2, switches S3 and S7 are ON: switch S6 has three volts across it and switch S5 has two volts across it, while the active switches 3, 4 only have one volt across them. In general, the selection switches S3-S10 will either have to have a higher voltage rating than the active switches 3, 4 or they will need to be implemented as cascaded low-voltage switches.
Another issue with this circuit is that the voltage across the selection switches S3-S10 can change polarity. This poses a difficulty because a MOSFET has a parasitic diode in parallel with it, the polarity of which depends on where the body contact of the MOSFET is tied. For example, in one embodiment, the first active switch 3 of the regulating network 16A is a MOSFET having a parasitic diode D1 with its positive terminal connected to the inductor L1. As a result, the first active switch 3 can only block a voltage that is higher at the terminal on the output side of the active switch 3 than the voltage at the inductor side of the active switch 3. Hence, the selection switches S3-S10 must be able to block in both directions. One way to do this is to connect two switches back-to-back with their bodies tied such that the cathodes (negative terminals) of their corresponding parasitic diodes connect to each other. Another way to do this is to provide circuitry for changing polarity of the parasitic diode on the fly, for example by providing a body-snatcher circuit.
The illustrated power converter further includes a disconnect switch S11 to protect the low voltage switches in the power converter in the event of a fault (described in U.S. Pat. No. 8,619,445). The disconnect switch S11 must be a high-voltage switch to achieve this function. Since this switch is not routinely operated, it can be made large to reduce its resistance. However, doing so increases die cost.
In some practices, because a boost converter is practical only when its duty cycle is between about 5% and 95%, there will be gaps in the space of available output voltages.
There are at least two ways to fill these gaps. A first way is to press the disconnect switch S11 into service as a linear regulator at the input of the regulating network 16A. Another way is to place a linear regulator at the output of the regulating network 16A. These both come at the cost of efficiency. Of the two, placing the linear regulator at the input is preferable because doing so impairs efficiency less than placing the linear regulator at the output.
In principle, it is possible to reconfigure either the regulating network 16A or the switching network 12A.
Instead of using a series of selection switches to present different voltages to a regulating network 16A with two output ports, as shown
The power converter shown in
In the power converter shown in
During normal operation, only two of the active switches S1-S5 are opening and closing at some specific frequency. The remaining active switches are disabled. The boost converter connected to multiple taps on the switching network 12A. This allows the boost converter to regulate the voltage difference between two dc capacitors by controlling the time ratio of the enabled active switches. Such control results in a regulated output voltage VO. Since the voltage difference between two dc capacitors is the output voltage VO, the active switches S1-S5 only need to support the output voltage VO, which is low voltage. In the illustrated examples, each active switch S1-S5 only has to support one volt. However, the disabled switches would normally have to see a higher voltage. Selection switches S6-S10 within the switching network 12A block this voltage, thereby sparing the disabled active switches from having to endure it.
Table 3 below shows the proper configuration of the switches to achieve a particular LX signal VLX.
The disconnect switch S11 is rated to handle the highest voltage. Its function is to disconnect the input from the output. However, it can also be used during startup, during a power to ground short, and to function as a linear regulator in the manner already discussed in connection with
The power converter of
In both cases, selection switches must be able to block in both directions. Thus, both the power converter of
One disadvantage of the power converter of
The switching network 12A shown in
In the power converters described thus far, adiabatic charging and discharging of capacitors in the switching network 12A is made possible by an inductor in the regulating network 16A. However, it is possible to separate the function of enabling adiabatic charge transfer and regulation by providing a separate magnetic filter. In
It is also possible to incorporate a magnetic filter into the power converters shown in
In some embodiments, the regulating network 16B participates in enabling adiabatic charge transfer even when a magnetic filter is present. For example, the magnetic filter may cause a first capacitor to charge adiabatically while the regulating network 16A causes the same first capacitor to discharge adiabatically, or vice versa.
The use of a magnetic filter provides another way to span the gap that arises when, in order to meet a voltage requirement, a regulating network 16A would have to operate at a duty cycle outside its permissible range of duty cycles. In embodiments without a magnetic filter, these gaps were filled by using a switch as a linear regulator. However, linear regulators are inefficient.
When a magnetic filter is made available, one can avoid using an inefficient linear regulator to span the gap by chopping the output of the switching network 12A and passing it through the magnetic filter to produce a dc output. In some embodiments, switches in the switching network 12A carry out the chopping. In other embodiments, an additional switch S12 can be added to aid in chopping. Note that elements shown connected with dotted lines are optional. In those embodiments in which a buck-boost converter implements the regulating network 16A, neither a linear regulator nor voltage chopping at the switching network 12A is required.
Table 4 below summarizes operation of two embodiments, one with and one without an additional switch S12. Option 1 of the table shows how the switches in the switching network 12A transition between first and second states to carry out chopping. Option 2 shows how the use of the additional switch S12 effectively adds a third state between the first and second states. A benefit of Option 2 is that it avoids having two series-connected switches conduct, thus reducing losses. In addition, Option 2 provides for a body diode that can conduct when the switching network 12A is transiting between states.
During normal operation the regulating network 16A alternates between a first and second state at a specific frequency and duty cycle, with the duty cycle determining the transformation ratio. During the first state, the first switch 3 closes and the second switch 4 opens. During the first switch 3 opens and the second switch 4 closes.
Operation of the power converter in
Another way to obtain a wide voltage range is to use a step-down switching network 12A and to implement the regulating network 16A with a buck converter with multiple taps, as shown in
An advantage of the power converter shown in
To obtain an even wider output range than that given by the power converter shown in
In
The timing diagram for the three operating modes is shown in Table 6:
Switches labeled “ON” in Table 6 are closed during the complete switching cycle. Switches labeled “OFF” are open during the complete switching cycle. As is apparent, each switch only has to support at most two volts. In addition, the switches are properly configured such that body snatcher circuits are not required. In particular, switch pair 1B, 2A and switch pair 1C, 2B show two body diodes pointing at each other. These switch pairs can therefore block voltages of any polarity.
The buck converter, which has two modes of operation, alternates between the switch configurations shown in Table 5. In the first mode, the LX signal VLX alternates between zero volts and two volts during a switching cycle. In the second mode, the LX signal VLX alternates between two volts and four volts during a switching cycle. The switches labeled “ON” are closed during the complete switching cycle and the switches labeled “OFF” are open during the complete switching cycle. Each switch only needs to be able support two volts. During normal operation the switching network 12A alternates between the first and second states at a specific frequency and duty cycle. In some embodiments, the duty cycle is 50%.
The first regulator includes first and second switches 3, 4 and a first inductor L1. The second regulator includes third and fourth switches 5, 6 and a second inductor L2.
During normal operation the first regulator alternates between a first and second state at a specific frequency and duty cycle. This duty cycle determines the transformation ratio. During the first state, the first switch 3 is closed and the second switch 4 is opened. During the second state, the states are reversed. The second regulator works in the same way with the third switch 5 replacing the first switch 3 and the fourth switch 6 replacing the second switch 4. The switching network 12A, the first regulator, and the second regulator can operate at the same or difference frequency and with any phase difference between them.
An advantage of the configuration shown in
The regulating network 16A includes an inductor L1, a first switch 3, a second switch 4, a third switch 5, and a fourth switch 6.
When the regulating network 16A operates in its boost mode, the intermediate voltage VX is higher than the input voltage VIN. In this mode, the third switch 5 and the fourth switch 6 are active, the first switch 3 is closed, and the second switch 4 is open.
Conversely, when the regulating network 16A operates in its buck mode, the intermediate voltage VX is lower than the input voltage VIN. In this mode, the first switch 3 and the second switch 4 are active, the third switch 5 is closed, and the fourth switch 6 is open.
Meanwhile, the switching network 12A includes first and second switch sets 1, 2, four selection switches S1-S4, four dc capacitors C1-C4, and three pump capacitors C5-C7. The voltages on the four dc capacitors C1-C4 are 4/2VX, 3/2VX, 2/2VX, and 1/2VX, respectively.
In operation, the four selection switches S1-S4 select different dc capacitors C1-C4 within the switching network 12A for presentation to the load 18A. By properly choreographing the enabling and disabling of the selection switches S1-S4 according to a distinct pattern, one can produce an ac output with a dc offset. This is particularly useful for envelope tracking when providing power to an RF power amplifier.
The regulating network 16A is a dual-inductor buck converter (shown as configuration “C1” in the table in the appendix) having a first inductor L1, a second inductor L2, a first capacitor C0, a first switch 3, and a second switch 4. In some embodiments, the first and second inductors L1, L2 are uncoupled. In others, the first and second inductors L1, L2 are coupled. These include embodiments that use both positive and negative coupling.
Unlike in a single-inductor buck converter, the input current into the dual-inductor converter is relatively constant. This results in lower rms current through the switching network 12A. Both terminals connected to the switching network 12A draw a relatively constant current. Because of this behavior, and because a pump capacitor would always be available to feed each inductor, the dual-inductor buck converter is best used with a full-wave cascade multiplier. It is also possible to use a half-wave cascade multiplier. However, in that case, the pump capacitors would only be feeding the inductors half the time. This requires providing high capacitance dc capacitors.
The switching network 12A includes first and second switch sets 1, 2, eight selection switches S1-S8, four dc capacitors C1-C4, and six pump capacitors C5-C10.
A third inductor L3 that feeds the switching network 12A promotes adiabatic charge transfer within the switching network 12A. Because it is only filtering a voltage ripple on the capacitors seen at the input of the switching network 12A, and because it does not have a particularly large voltage across it, this third inductor L3 has a much smaller inductance than those required within the regulating network 16A.
Enabling and disabling different selection switches S1-S8 reconfigures the switching network 12A, thus enabling one to change the offset voltage Voff of the switching network 12A. Table 7 shows switching patterns used to achieve four different offset voltages.
In the particular example shown, the load connected to the output of the power converter comprises a plurality of light-emitting diodes connected in series with each other and with the current path through a transistor biased by a voltage VB. This permits control over the LED current, thus enabling the brightness of the LED, which is proportional to the LED current ILED, to be controlled. The combination of a power converter and a circuit to control the LED current ILED, which amounts to a current sink, is commonly called an LED driver. In most embodiments, the current sink is somewhat more complicated than a single transistor. However, the principles illustrated in
Instead of using a linear regulator to fill the gaps as discussed in reference to
In another embodiment of the power converter of
The two-phase boost converter includes a first inductor L1, a second inductor L2, a first switch 5, a second switch 6, a third switch 7 and a fourth switch 8. A circuit 20A within the regulating network 16A receives control signals from a controller via a first path P1. During normal operation, the circuit 20A provides first drive signals provided to the first and second switches 5, 6 and second drive signals to the third and fourth switches 7, 8. The first and second drive signals are in phase quadrature. Controlling the duty cycle of at which the switches 4-8 switch regulates the output voltage VO.
The switching network 12A includes six selection switches S1-S6, six pump capacitors C5-C10, four dc capacitors C1-C4, first and second switch sets 1, 2, and a circuit 20B that provides drive signals to the first and second switch sets 1, 2 and to the six selection switches S1-S6 based on control signals received along a path P2 from the controller.
The reconfigurable fractional charge pump 22 has multiple modes. In the particular example described herein, the modes are a 1:1 mode and a 3:2 mode.
The reconfigurable fractional charge pump 22 enables the parent cascade multiplier in which it is embedded to output half ratios. Table 8 shows the available transformation ratios (V4:VO) for the parent cascade multiplier, and both the switch states and the transformation ratio of the embedded reconfigurable fractional charge pump 22 that would be required to achieve those transformation ratios.
A particular benefit of providing half ratios is that the resulting power converter operates without the gaps described in connection with
For example, suppose that the output voltage VO is one volt and that the input voltage VIN is 3.5 volts. The transformation ratio is 4:1. This requires a duty cycle of 50%, which is well within the permissible range for the regulating circuit (i.e., the dual-phase boost converter).
Suppose now that the input voltage VIN drops to 3.05 volts. At this point, the required duty cycle at the regulating circuit would drop to 5%, which is below the acceptable limit. Ordinarily, this would result in a gap. But not for the circuit shown in
In addition, to eliminating gaps, the ability to provide half ratios enables the regulating circuit to run with a duty cycle between 25% and 75%. This has many benefits, including reducing the rms current and thus boosting the efficiency.
The reconfigurable fractional charge pump 22 operates in either a first mode or a second mode. When operating in the first mode, the reconfigurable fractional charge pump 22 provides a transformation ratio V1:V2 of 3:2. When operating in the second mode, the transformation ratio is 1:1.
To operate in the first mode, the selection switch S0 opens, and the first and second switch sets 3, 4 switch open and close at some specific frequency. In some embodiments, the switches open and close with a 50% duty cycle. The first switch set 3 and the second switch set 4 are always in opposite states.
To operate in the second mode, the selection switch S0 closes, and the first and second switch sets 3, 4 open.
In the embodiment shown in
The first switching network 12A includes four dc capacitors C1-C4, three pump capacitors C5-C7, and first and second switches 1, 2. In the configuration shown, the first switching network 12A is not adiabatically charged. Hence, it operates with a duty cycle of near 50%. However this is not required because stability is not an issue. In operation, the first switching network 12A provides a first voltage V1 that is four times the input voltage VIN. However, the intermediate voltage VX that the regulating network 16A receives from the first switching network 12A is twice the input voltage VIN.
The regulating network 16A includes a first switch 5, a second switch 6, and an inductor L1. The regulating network 16A controls the duty cycle of the first and second switches 5, 6 to regulate its output voltage VO.
The second switching network 12B includes first and second switch sets 3, 4, two dc capacitors C11-C12, and three pump capacitors CB-C10. Unlike the first switching network 12A, there is an inductor L1 that feeds the second switching network 12B. As a result, the second switching network 12B is adiabatically charged.
A dc capacitor C12 connected at the output of the regulating network 16A impedes adiabatic operation and is thus optional. This dc capacitor C12 is typically added only to maintain stability. As a result, its capacitance is much smaller than that of other capacitors in the network.
Since the second switching network 12B is adiabatically charged and its transformation ratio is 1:4, operating it at a duty cycle of 50% promotes stability. The overall output voltage VO of the power converter is given by VO=8VIN(D+1), where the duty cycle D is equal to the duty cycle D of the second switch 6 of the regulating network 16A.
The regulating network 16A includes an inductor L1, a first switch 5, and a second switch 6.
The first and second switching networks 12A, 12B are the same as those shown in
The differences between the power converter in
In the embodiments shown, the first regulating network 16A is implemented as a step-up converter while the second regulating network 16B is implemented as a step-down converter. However, this does not have to be the case. For example, the order could be reversed, with the first regulating network 16A causing a step-down in voltage, and the second regulating network 16B causing a step-up in voltage. Or, both the first and second regulating networks 16A, 16B could cause a step-up or a step-down in voltage. An advantage of the particular configuration shown in the figures is that if the switching network 12A were a reconfigurable switching network, such as that shown in
The illustrated embodiments feature a controller to control operation of switches in the first and second regulating networks 16A, 16B. Such a controller can be used to implement a variety of control techniques that can be used in connection with controlling the operation of the first and second regulating networks 16A, 16B. In some embodiments, the controller implements feed-forward control over the first regulating network 16A and feedback control over the second regulating network 16B. In other embodiments, the controller implements feedback over the first regulating network 16A and feed-forward control over the second regulating network 16B.
In the embodiment shown in
The switching network 12A includes first and second switch sets 1, 2, three dc capacitors C1-C3, and six pump capacitors C4-C9 spread across two symmetric step-up cascade-multipliers. The two cascade multipliers share a common phase pump and operate 180 degrees out of phase. The operation of this switching network 12A is similar to that of a dual-phase version, or full-wave, cascade multiplier. The main distinction arises from separation of the bottom and top of the switch stacks.
Because the stack switches are separate, it is possible to create an intermediate voltage VX3 that is a difference between the voltages present at the tops of the switch stacks. In the embodiment shown, the voltage at the top of a first switch stack is 4VX, whereas the voltage at the top of a second switch stack is 3VX+VIN. Thus, the intermediate voltage VX3 is the difference between these, which is VX−VIN.
In the embodiment shown in
The switching network 12A includes first and second switch sets 1, 2, three dc capacitors C1-C3, and nine pump capacitors C4-C12 spread across two symmetric step-up cascade-multipliers. The two cascade multipliers share a common phase pump and operate 180 degrees out of phase. Unlike the switching network 12A shown in
In operation, the three-level boost converter operates in two modes. In each mode, the boost converter cycles through first, second, third, and fourth states at a particular frequency. Each state corresponds to a particular configuration of switches. Table 9A shows the four states in the first mode, and Table 9B shows the four states in the second mode.
Within each mode, the three-level boost converter regulates its output by controlling its generalized duty cycle. The generalized duty cycle is equal to a first time interval divided by a second time interval. The first time interval is equal to the amount of time the three-level boost converter spends in either the first state or the third state. The second time interval is the amount of time the three-level boost converter spends in either the second state or the fourth state.
When the three-level buck converter operates in the first mode, the intermediate voltage VX is greater than two times the input voltage VIN. In contrast, when the three-level buck converter operates in the second mode, the intermediate voltage VX is less than two times the input voltage VIN.
An advantage over the power converter shown in
The power converter of
In the switching network 12A shown in
In operation, a first voltage V1 is equal to VX+5VIN, a second voltage V2 is equal to VX+3VIN, and a third intermediate voltage VX3 is equal to 2VIN.
An advantage of the power converter shown in
Additionally, since the additional inductor L3 only has to promote adiabatic charge transfer, it can have a smaller inductance that the inductor L1 in the boost converter. This, in turn, reduces resistive inductor losses.
However, a disadvantage of the power converter shown in
The first regulating network 16A includes a first switch 3, a second switch 4, and an inductor L1. The second regulating network 16B includes a first switch 5, a second switch 6, and an inductor L2. The switching network 12A includes a first switch set 1, a second switch set 2, four dc capacitors C1-C4, and six pump capacitors C5-C10.
In the power converter shown in
The regulating network 16A includes a first inductor L3, a second inductor L4, a capacitor C10, a first switch 3, and a second switch 4. Depending upon the duty cycle, a Zeta converter can step-up or step-down the voltage. However, a disadvantage of the Zeta converter is the requirement for more passive components. In addition, a Zeta converter is more difficult to stabilize because of the additional poles and zero introduced.
The switching network 12A includes first and second switch sets 1, 2; two selection switches S1-S2, three dc capacitors C1-C3, six pump capacitors C4-C9, a first inductor L1, and a second inductor L2.
In operation, the switching network 12A transitions between a first mode and a second mode. During the first mode, the first selections switch S1 is closed and the second selection switch S2 is open. The intermediate voltage VX is then VIN/2. During the second mode, the first selection switch S1 is open and the second selection switch S2 is closed. In the second mode, the intermediate voltage VX becomes the input voltage VIN.
One way to achieve adiabatic inter-capacitor charge transfer within the switching network 12A is to place a small inductor in series with the second switch S2. However, although this would promote adiabatic charge transfer during the first mode, it would not do so during the second mode.
Another way to achieve adiabatic inter-capacitor charge transfer within the switching network 12A is to embed the first inductor L1 within the charge pump and the second inductor L2 in series with the ground terminal of the charge pump.
Preferably the first inductor L1 is embedded at a location that carries a constant current and that connects to charging and discharging paths of as many pump capacitors C4-C9 as possible. A suitable location is therefore at the phase pump.
A charge pump typically has two nodes that carry constant current. As shown in
The present application is a continuation of U.S. patent application Ser. No. 16/883,872, filed on May 26, 2020, now issued as U.S. Pat. No. 11,664,727, which is a continuation of U.S. Patent application Ser. No. 15/742,660, filed on Jan. 8, 2018, which is a national stage entry of PCT Application Serial No. PCT/US2016/041448, filed on Jul. 8, 2016, which claims priority from U.S. Provisional Application No. 62/189,909, filed on Jul. 8, 2015. The contents of these applications are herein incorporated by reference in their entirety.
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| 20240014736 A1 | Jan 2024 | US |
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