SWITCHED-MODE POWER SUPPLY SYSTEM AND SPEED VARIATOR COMPRISING SUCH A SYSTEM

Abstract

The invention relates to a switched-mode power supply system comprising two buffer capacitors connected in series and connected between the two input terminals of a DC input voltage source, two switches connected in series, a primary inductive assembly connected in series with the two switches, and a secondary winding magnetically coupled to the primary inductive assembly in order to deliver a DC output voltage. A current injection module injects current at a mid-point of the two buffer capacitors in order to generate a current imbalance at this mid-point. A balancing circuit for balancing the leakage currents of the capacitors, one end of said balancing circuit being connected to the mid-point of the buffer capacitors, maintains this imbalance at a predetermined value. The invention also relates to a speed variator using such a power supply system.

Description

Other features and advantages will be apparent in the following detailed description, which refers to embodiments given by way of examples and depicted by the appended drawings, in which:

FIG. 1 shows a power supply system integrated into a speed variator;

FIG. 2 shows in detail an example of a power supply system according to the invention;

FIG. 3 shows a preferred embodiment of a control circuit for a switch of the power supply system;

FIG. 4 shows a second embodiment of this control circuit;

FIG. 5 shows a variant of a balancing circuit for the power supply system; and

FIG. 6 shows a simplified diagram for the connection of the balancing circuit.

Referring to FIG. 1, an industrial speed variator is supplied from a three-phase AC power supply mains 5, for example in the 400 V to 690 V voltage range. It is based on an AC/DC/AC architecture (AC: alternating current; DC: direct current). Such a variator includes a rectifier module 20 delivering a DC bus voltage between the positive terminal A2 and the negative terminal A1 from the input AC voltage, which may for example be a three-phase or single-phase voltage. It then includes an inverter module 30 delivering, from this DC bus a variable AC output voltage for supplying the electrical load driven by the variator, such as a three-phase or single-phase motor M. The bus voltage also serves for supplying a switched-mode power supply system 10.

FIG. 2 shows the details of one example of the architecture of the power supply system 10 based on a DC/DC switched-mode supply with an isolated transformer, of the flyback converter type. The invention could also apply to a DC/DC switched-mode power supply with an isolated transformer, of the forward converter type.

The power supply system 10 includes a capacitor buffer assembly which is connected between the input terminals A1 and A2 and comprises two electrolytic buffer capacitors CB1 and CB2 connected in series. A mid-point P_MC is positioned between the capacitors CB1 and CB2. The first buffer capacitor CB1 is connected between the negative input terminal A1 and the mid-point P_MC of the capacitors. The second buffer capacitor CB2 is connected between the mid-point P_MC and the positive input terminal A2. Preferably, the two buffer capacitors CB1, CB2 have the same values and are therefore designed to withstand around one half of the maximum bus voltage V_bus. One of the objectives of the present invention is to find a simple, easily implementable and inexpensive solution for balancing the voltages across the terminals of the two buffer capacitors CB1, CB2 without having to use dividing resistors.

The power supply system 10 includes a first switch SW1 and a second switch SW2 connected in series and driven by a first control circuit 11 and a second control circuit 12 respectively. The power supply system also includes a primary inductive assembly connected in series with the two switches SW1 and SW2.

Preferably, the two switches SW1 and SW2 are power transistors of the MOSFET or IGBT type, each possessing a gate G, the control of which allows a current to flow between a drain D and a source S. The source S1 of the first switch SW1 is connected to the negative input terminal A1. The drain D1 of the first switch SW1 is connected to the source S2 of the second switch SW2. The drain D2 of the second switch SW2 is connected to one end of the primary inductive assembly. The other end of the primary inductive assembly is connected to the positive input terminal A2. Thus, when the switches SW1 and SW2 are conducting, this means that a current flows through the primary inductive assembly. Moreover, a conventional clamping circuit 19 is added between the positive terminal A2 and the drain D2 of the second switch SW2 in order to protect the latter from overvoltages at the moment of opening it.

The gate G1 and the source S1 of the first switch SW1 are connected to the first control circuit 11. This control circuit 11 is a conventional circuit for opening (turning off) and closing (turning on) a MOSFET transistor. It is driven by an external control signal S, which for example may come from the control unit of the variator. The gate G2 and the source S2 of the second switch SW2 are connected to the second control circuit 12. The structure of the control circuit 12 will be detailed later. The fact of choosing two switches SW1 and SW2 in series makes it possible for the voltage withstood at the terminals of each of the switches to be divided by two, and therefore their cost and their size to be reduced compared with use of a single switch withstanding the entire DC bus voltage.

In the example shown in FIG. 2, the primary inductive assembly is made up of two primary windings Np1, Np2 which are connected directly in series between the positive terminal A2 and the drain D2 of switch SW2. A mid-point P_ML is positioned between the two windings Np1, Np2. This mid-point P_ML therefore constitutes a fixed voltage source V_CL, the value of which depends on the ratio of the two windings Np1, Np2. If the primary windings Np1, Np2 have identical inductance values, then the voltage at the mid-point P_ML is equal to one half of the bus voltage V_bus.

Moreover, the power supply system includes one or more secondary windings that are magnetically coupled to the primary inductive assembly Np1, Np2 for the purpose of delivering one or more corresponding DC output voltages. The example in FIG. 2 shows a single secondary winding Ns delivering an isolated secondary output voltage Us. These DC voltages serve to supply various components of the speed variator, especially the electronic circuits, including the control unit of the variator. It does not matter whether some of these output voltages are isolated and others not isolated from the input supply voltage for the DC bus. The advantage of having an isolated voltage is for example to be able to have a DC voltage available for the user as output of the variator.

According to the invention, the power supply system includes a current injection module, which serves to inject an additional current I_ADD at the mid-point P_MC of the two buffer capacitors so as to create a current imbalance at this mid-point P_MC. This additional current I_ADD will cause the voltage VC1 across the terminals of the first buffer capacitor CB1 to rise, to the detriment of VC2, in such a way that VC1 is greater than or equal to VC2 since VC1+VC2=V_bus.

The power supply system also includes a balancing circuit 15 intended to keep the imbalance of the leakage currents of the buffer capacitors CB1, CB2 at a predetermined value. The first end of the balancing circuit 15 is connected to the mid-point P_MC of the two buffer capacitors. The second end of the balancing circuit 15 is connected to a fixed voltage source V_CL. Preferably, the balancing circuit 15 is made up of a resistor R₀ in series with a diode D₀, the anode of the diode Do being directed towards the mid-point P_MC.

FIG. 6 shows a diagram of the general principle of the connection of the voltage source V_CL and the balancing circuit 15, and FIGS. 2 and 5 show two exemplary embodiments. The voltage source V_CL is connected between the positive input terminal A2 and the second end of the balancing circuit 15.

According to first preferred variant, since it is of very simple construction, indicated in FIG. 2, the second end of the balancing circuit 15 is connected directly to the mid-point P_ML of the two primary windings Np1, Np2, which therefore constitutes the voltage source V_CL. One of the windings Np2 is therefore connected between the positive terminal A2 and the balancing circuit 15. The balancing circuit 15 thus makes it possible to balance the voltage VC2 with the voltage V_CL, since the diode D₀ will conduct current as long as the voltage VC1 is above V_CL (or as long as the voltage VC2 is below V_CL). A current will therefore flow through the balancing circuit 15 until VC1 is equal to V_CL. If V_CL is chosen to be one half of the bus voltage V_bus (i.e. when the two primary windings Np1, Np2 are identical), then the voltages VC1 and VC2 are thus maintained at one half of the overall bus voltage V_bus (since VC1≧VC2) and therefore the desired balancing is achieved.

In a second variant indicated in FIG. 5, the voltage source V_CL is formed from a secondary balancing winding Ns2 magnetically coupled to the primary inductive assembly. The second end of the balancing circuit 15 is connected to one end of the secondary balancing winding Ns2, the other end of this balancing winding Ns2 being connected to the positive input terminal A2. The inductance of this specific balancing winding Ns2 is for example defined as one half of the primary inductance so as in this way to be able to keep the voltage VC2 at one half of V_bus. It should be pointed out that, in this second variant, the primary inductive assembly may then be made up of one or two primary windings, since the mid-point P_ML is no longer used.

The current injection module therefore intentionally creates a voltage imbalance at the mid-point P_MC by injecting the additional current I_ADD (see FIG. 6), thus making it possible for the balancing circuit 15 to regulate this imbalance about a value predetermined by the voltage source.

The module for injecting the additional current I_ADD could be produced by any additional current source. For example, this current source could be created using an additional resistor connected in parallel with the buffer capacitor CB2 or using an external converter circuit connected to the terminals A1, A2 and to the point P_MC. However, these solutions require the use of a large resistor, thus incurring compulsory heat losses.

This is why the invention advantageously proposes for this current injection module to be produced in the second control circuit 12, two embodiments of which are shown in detail in FIGS. 3 and 4. Thus, the role of the second control circuit 12 is the following: to synchronize the second switch SW2 with the first switch SW1 (i.e. to open SW2 when SW1 is open and close SW2 when SW1 is closed) and to inject an additional current I_ADD at the mid-point PMC.

FIG. 4 shows a minimal embodiment of the second control circuit 12 of the second switch SW2. In this embodiment, the control circuit 12 comprises a protection zener diode D2 connected between the gate G2 and the source S2 of the switch SW2, the anode of the protection diode D2 being directed towards the source S2. The control circuit 12 also includes a bypass diode D1 connected between the gate G2 of the switch SW2 and the mid-point P_MC of the capacitors CB1, CB2, the anode of the bypass diode D1 being directed towards the gate G2. Finally, the control circuit 12 includes a charging capacitor C4 in parallel with the bypass diode D1. The operation of such a control circuit 12 is as follows:

sequence for closing the switches SW1 and SW2. We assume initially that the two switches SW1 and SW2 are open. Control of the switches is initialized by a closing command for the signal S input into the first control circuit 11. The first switch SW1 is directly controlled, whereas the second switch SW2 is controlled via the voltage VDS1 between the source S1 and the drain D1 of the switch SW1. This voltage VDS1 drops when the switch SW1 closes. Consequently, a charging current -IC4 flows through the charging capacitor C4 towards the source-gate capacitor CSG2 of the switch SW2. Thus, the gate-source voltage VGS2 increases. At the moment when this voltage VGS2 reaches the breakdown voltage of the zener diode D2 (for example around 10 to 15 V), then a drain-source current IDS2 starts to flow and to discharge the drain-gate and drain-source capacitors of the switch SW2. Consequently, the drain-source voltage VDS2 of the switch SW2 drops. Because of the Miller effect, the voltage VGS2 remains approximately constant, whereas the voltage VDS2 drops. Once the voltage VDS2 reaches zero, the voltage VGS2 continues to rise and forces the MOSFET SW2 switch into the resistive linear region. The voltage VGS2 increases up to the breakdown voltage of the diode. The voltage VGS2 and the current IC4 then become zero. Shortly afterwards, the switch SW2 is fully supplied. The biased current needed to maintain a stable state is the leakage current of the bypass diode D1 which flows through the zener diode D2;

sequence for opening the switches SW1 and SW2. We now assume that the two switches SW1 and SW2 are closed. The voltage VGS2 is a maximum and the capacitor C4 is fully charged. Control of the switches is initialized by an opening command for the signal S input into the first control circuit 11. The first switch SW1 is controlled directly, whereas the switch SW2 is controlled by the voltage VDS1. When the switch SW1 opens, the voltage VDS1 starts to increase. The protection diode D2 is reverse-biased and the charging capacitor C4 can discharge with a discharge current IC4 flowing to the mid-point P_MC. This discharge current IC4 is approximately the inverse of the charging current -IC4 occurring during the closure phase of SW1, SW2. Once the voltage VDS1 has reached V_bus/2, the capacitor C4 is discharged and D1 starts to conduct a current ID1. For quite a short time, the switch SW1 is open while the switch SW2 is not yet open. Therefore during this time, the current flowing through SW2 is diverted towards the diodes D1 and D2 and the mid-point PMC. At the same time, the capacitor CSG2 has discharged and the voltage VGS2 has dropped. As long as the voltage VGS2 is still above the Miller voltage, the switch SW2 conducts. At the moment when the voltage VGS2 has reached the Miller voltage, the switch SW2 starts to open and the current ID1 then drops back to zero.

The situation as regards the main currents flowing through the second control circuit 12 involves the following:

a charging current -IC4, for charging the capacitor C4 during the closure phase;

a reverse current IC4, for discharging C4 during the opening phase; and

a current ID1 flowing to the mid-point P_MC through D1 and D2 when SW1 is open and SW2 is closed, during the opening phase.

As the phases of opening and closing SW1 and SW2 rapidly follow each other during normal operation of the power supply system, the sum of these currents delivers overall the additional current I_ADD that flows to the mid-point PMC.

FIG. 3 shows a preferred embodiment of the second control circuit 12 for the second switch SW2. In this embodiment, certain passive components have been added to the embodiment shown in FIG. 4, for greater efficiency of the power supply system.

This preferred embodiment comprises a filter made up from a filtering resistor R3, connected between the gate G2 of the switch SW2 and the cathode of the protection diode D2, and from a filtering capacitor C3 between the anode of the diode D2 and the source S2 of SW2. The main purpose of this filter is to delay the opening of the switch SW2, and therefore to increase the flow time of the current ID1 and therefore to ensure a higher current I_ADD. FIG. 3 also shows, in parallel with the resistor R3, an acceleration diode D3 with its cathode directed towards the gate G2. The diode D3 shunts the resistor R3 during closure of SW2, in order to accelerate switching. Finally, FIG. 3 shows two biased resistors R1 and R2. The resistor R1 is connected in parallel with the capacitor C4, and the resistor R2 is inserted between the capacitor C4 and the cathode of the diode D2. The purpose of R1 is to improve the conduction of the switch SW2 in the closed state. The purpose of R2 is to damp any oscillations due to any parasitic inductance of the second control circuit 12.

The switched-mode power supply system described in the present invention can obviously be used in any DC/DC converter and is not limited to a speed variator.

Of course, it is possible, without departing from the scope of the invention, to imagine other variants and detailed improvements and even to envisage the use of equivalent means.

Claims

1. Switched-mode power supply system (10) comprising: two input terminals, namely a negative input terminal (A1) and a positive input terminal (A2) respectively, which are intended to receive a DC input supply voltage;a capacitive buffer assembly connected between the input terminals and comprising two buffer capacitors (CB1, CB2) connected in series;a first switch (SW1) and a second switch (SW2) of the MOSFET or IGBT power transistor type, which are driven by a first and a second control circuit (11, 12) respectively;a primary inductive assembly (Np1, Np2); andat least one secondary winding (Ns) magnetically coupled to the primary inductive assembly (Np1, Np2) in order to deliver a DC output voltage (Us),

2. Power supply system according to claim 1, characterized in that the mid-point (PMC) of the two buffer capacitors is connected to the gate (G2) of the second switch (SW2).

3. Power supply system according to claim 1 or 2, characterized in that the balancing circuit (15) comprises a resistor (R0) and a diode (D0) connected in series, the anode of said diode being directed towards the mid-point (PMC) of the two buffer capacitors.

4. Power supply system according to claim 3, characterized in that the primary inductive assembly comprises two primary windings (Np1, Np2) connected in series and in that a mid-point (PML) of the two primary windings is connected to a second end of the balancing circuit (15).

5. Power supply system according to claim 3, characterized in that it comprises a balancing winding (Ns2) magnetically coupled to the primary inductive assembly (Np1, Np2), one end of which is connected to the positive input terminal (A2) and the other end of which is connected to a second end of the balancing circuit (15).

6. Power supply system according to claim 2, characterized in that the second control circuit (12) comprises a protection zener diode (D2), the cathode of which is connected to the gate of the second switch (SW2) and the anode of which is connected to the source of the second switch (SW2), a bypass diode (D1), the cathode of which is connected to the mid-point (PMC) of the buffer capacitors and the anode of which is connected to the cathode of the protection diode (D2), and a charging capacitor (C4) connected in parallel with the bypass diode (D1).

7. Power supply system according to claim 2, characterized in that the second control circuit (12) comprises: a protection zener diode (D2), the anode of which is connected to the source of the second switch (SW2) and the cathode of which is connected to the gate of the second switch (SW2) via a filtering resistor (R3);a bypass diode (D1), the cathode of which is connected to the mid-point (PMC) of the buffer capacitors and the anode of which is connected to the cathode of the protection diode (D2);an acceleration diode (D3) in parallel with the filtering resistor (R3), the cathode of which diode is directed to the gate of the second switch (SW2);a filtering capacitor (C3) connected between the gate and the source of the second switch (SW2); anda charging capacitor (C4) connected to the mid-point (PMC) of the buffer capacitors and to the cathode of the protection diode (D2) via a biasing resistor (R2).

8. Power supply system according to claim 1, characterized in that the system delivers several output voltages (Us), some of which are isolated and others not isolated from the input supply voltage.

9. Speed variator comprising a rectifier module (20) delivering a DC input voltage and an inverter module (30) delivering a variable output voltage to an electrical load, characterized in that the variator includes a power supply system (10) according to one of the preceding claims, which is supplied by the DC input voltage.