This application claims priority from Japanese Patent Application No. JP 2006-067730 filed in the Japanese Patent Office on Mar. 13, 2006, the entire content of which is incorporated herein by reference.
1. Field of the Invention
The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.
2. Description of the Related Art
Most of power supply circuits that rectify commercial alternating-current power and provide desired direct-current voltage have recently been switching type power supply circuits. A switching power supply circuit has a transformer and other devices miniaturized by increasing switching frequency, and is used as a power supply for various electronic devices as a high-power DC-to-DC converter.
The commercial alternating-current power is a sinusoidal alternating voltage. When a smoothing and rectifying circuit using a rectifying element and a smoothing capacitor rectifies and smoothes the commercial alternating-current power, due to a peak hold effect of the smoothing and rectifying circuit, a current flows from the commercial alternating-current power supply to the switching power supply circuit during a short period around a peak voltage of the alternating voltage, and the current flowing from the commercial alternating-current power supply to the switching power supply circuit has a distorted waveform that differs greatly from a sinusoidal wave. Then a power factor indicating efficiency of use of the power supply is deteriorated. In addition, a measure is necessary for suppressing harmonics, which result from such a distorted current waveform of the cycle of the commercial alternating-current power. A method using a so-called active filter as in Japanese Patent Laid-open No. Hei 6-327246, for example, to solve these problems is known as a technology in related art for improving the power factor.
A configuration for power factor improvement includes: the step-up type converter composed of an inductor L, a fast recovery type fast switching diode D, and a switching element Q; and a control section for the step-up type converter. The control section has a multiplier 111 as a main component. The inductor L and the fast switching diode D are inserted between the positive electrode output terminal of the primary side rectifying element Di and the positive electrode terminal of the smoothing capacitor Cout in a state of being connected in series with each other. A resistance Ri is inserted between the negative electrode output terminal of the primary side rectifying element Di (primary side ground) and the negative electrode terminal of the smoothing capacitor Cout. The switching element Q is a MOS-FET, for example. The switching element Q is inserted between a point of connection between the inductor L and the fast switching diode D and the primary side ground.
The multiplier 111 is connected with a current detection line LI, a waveform input line Lw, and a voltage detection line Lv. The multiplier 111 detects a signal corresponding to a rectified current Iin flowing through the negative electrode output terminal of the primary side rectifying element Di from across the resistance Ri. The signal is input from the current detection line LI. In addition, the multiplier 111 detects a signal corresponding to a rectified voltage Vin at the positive electrode output terminal of the primary side rectifying element Di. The signal is input from the waveform input line Lw. This rectified voltage Vin is obtained by converting the waveform of an alternating input voltage VAC from the commercial alternating-current power supply AC into an absolute value. Further, the multiplier 111 detects a variation difference of the direct-current input voltage (a signal obtained by amplifying a difference between a predetermined reference voltage and the direct-current voltage Vout will be referred to as a variation difference, which will hereinafter be used similarly) on the basis of the direct-current voltage Vout of the smoothing capacitor Cout. The direct-current voltage Vout is input from the voltage detection line Lv. Then, a drive signal for driving the switching element Q is output the multiplier 111.
The multiplier 111 multiplies together the signal, which is corresponding to the rectified current Iin and detected from the current detection line LI, and the variation difference, which is detected from the voltage detection line Lv, of the direct-current input voltage. The multiplier 111 detects an error between a result of the multiplication and the signal, which is corresponding to the rectified voltage Vin and is detected from the waveform input line Lw. After amplifying the error signal, the multiplier 111 performs a PWM (Pulse Width Modulation) conversion, and controls the switching element Q by a binary signal having a high level and a low level. Thus, a two-input feedback system is formed, the value of the direct-current voltage Vout is made to be a predetermined value, and the rectified current Iin is made to have a waveform similar to that of the rectified voltage Vin. As a result, the waveform of the alternating voltage applied from the commercial alternating-current power supply AC to the primary side rectifying element Di and the waveform of the alternating current flowing into the primary side rectifying element Di are also similar to each other, so that the power factor approaches substantially one. Thus power factor improvement is achieved.
The power supply circuit shown in
The positive electrode output terminal of the primary side rectifying element Di is connected to the positive electrode terminal of a smoothing capacitor Ci via a series connection of the inductor LN, a choke coil PCC (functioning as an inductor Lpc), and a fast recovery type fast switching diode D20. The smoothing capacitor Ci has the same function as the smoothing capacitor Cout in
A switching element Q103 corresponds to the switching element Q in
In addition, a series connection of a voltage dividing resistance R101 and a voltage dividing resistance R102 is provided between the positive electrode output terminal of the primary side rectifying element Di and a primary side ground. A point of connection between the voltage dividing resistance R101 and the voltage dividing resistance R102 is connected to a terminal T5. Thereby a voltage rectified by the primary side rectifying element Di is divided and then input to the terminal T5. The voltage of a resistance 103, that is, a voltage corresponding to the source current of the switching element Q103 is input to a terminal T2. The source current of the switching element Q103 is a current contributing to storing of magnetic energy, of a current I101 flowing through the choke coil PCC. Then, a second feedback control circuit is formed which makes the signal, which is input to the terminal T5 of the power factor and output voltage controlling IC 120, corresponding to the rectified voltage have a similar form to that of the signal corresponding to the envelope of the voltage input to the terminal T2 (that is, the envelope of the current I101).
In addition, a terminal T4 is supplied with operating power for the power factor and output voltage controlling IC 120. A half-wave rectifier circuit formed by a rectifier diode D11 and a series resonant capacitor C11 shown in
A drive signal (gate voltage) for driving the switching element is output from a terminal T3 to the gate of the switching element Q103. That is, the drive signal for operating the two feedback control circuits is input to the gate of the switching element Q103. The two feedback control circuits are the first feedback control circuit for making the value of the voltage divided by the above-mentioned voltage dividing resistances R5 and R6 a predetermined value and the second feedback control circuit for making the envelope of the current I101 have a similar form to that of the direct-current input voltage Ei. Thereby the waveform of an alternating input current IAC flowing in from the commercial alternating-current power supply AC is substantially the same as the waveform of the alternating input voltage VAC, so that the power factor is controlled to be substantially one. That is, the power factor is improved.
Returning to
A converter transformer PIT is provided to transmit the switching output of the switching element Q101 and the switching element Q102 from a primary side to a secondary side. One terminal part of a primary winding N1 of the converter transformer PIT is connected to a point of connection between the switching element Q101 and the switching element Q102 (switching output point) via a primary side series resonant capacitor C101. Another terminal part of the primary winding N1 is connected to the primary side ground. The primary side series resonant capacitor C101 and a primary side leakage inductance L1 form a series resonant circuit. The series resonant circuit performs a resonant operation by being supplied with the switching output by the switching element Q101 and the switching element Q102.
A secondary winding N2 is wound on the secondary side of the converter transformer PIT. The secondary winding N2 in this case has a secondary winding part N2A and a secondary winding part N2B provided with a center tap as shown in
First, as shown in
The power factor PF is substantially constant as the load power Po is varied, as shown in
As shown in
As is understood from the description so far, the power supply circuit shown in
However, the power supply circuit of the configuration shown in
In addition, the active filter circuit performs hard switching operation, and thus causes much noise. Therefore a strict noise suppressing measure is necessary. Thus, in the circuit shown in
Further, while the switching frequency of the switching element Q103 operated by the power factor and output voltage controlling IC 120 as a general-purpose IC is fixed at 60 kHz, the switching frequency of the current resonant converter in the following stage is varied in a range of 80 kHz to 200 kHz. The switching timing (clocks) of the switching element Q103 and the current resonant converter are thus different from each other and independent of each other. Therefore, due to the switching operations of the switching element Q103 and the current resonant converter operating on the respective clocks, ground potentials interfere with each other and become unstable, and for example an abnormal oscillation tends to occur. This also invites for example problems of more difficult circuit design and degradation in reliability.
The part of the current resonant converter in the power supply circuit having the configuration shown in
A switching power supply circuit according to an embodiment of the present invention may include a rectifying and smoothing section, a converter section, and a power factor improving section. The rectifying and smoothing section may convert an alternating input voltage from an alternating-current power supply into a primary side direct-current voltage. The converter section may convert the primary side direct-current voltage from the rectifying and smoothing section into an alternating voltage and further converts the alternating voltage into a secondary side direct-current voltage. The power factor improving section may improve a power factor. The rectifying and smoothing section may include a primary side rectifying element for rectifying the alternating input voltage and a smoothing capacitor for smoothing rectified output from the primary side rectifying element. The converter section may include a choke coil; a converter transformer; a main switching element; an oscillating and driving circuit; a primary side series resonant capacitor; a primary side parallel resonant capacitor; and an active clamping circuit. The choke coil may have one terminal connected to the smoothing capacitor. The converter transformer has a primary winding on a primary side and a secondary winding on a secondary side. The primary winding may be connected to another terminal of the choke coil, and the secondary winding may be magnetically loosely coupled with the primary winding. The main switching element may switch the primary side direct-current voltage supplied via the primary winding of the converter transformer. The oscillating and driving circuit drives on-off of the main switching element. The control circuit may supply the oscillating and driving circuit with a control signal to make a value of the secondary side direct-current voltage output by a secondary side rectifier circuit connected to the secondary winding of the converter transformer a predetermined value. The primary side series resonant capacitor has a predetermined capacitance. The primary side series resonant capacitor may have one terminal connected to a point of connection between the choke coil and the primary winding, whereby the primary side series resonant capacitor may be connected in series with one of the choke coil and the primary winding. The predetermined capacitance may be set such that a primary side first series resonance frequency governed by an inductance of the choke coil and the predetermined capacitance is substantially twice a primary side second series resonance frequency governed by a leakage inductance occurring at the primary winding and the predetermined capacitance. The primary side parallel resonant capacitor has a predetermined capacitance. The primary side parallel resonant capacitor may be connected in parallel with the main switching element. The predetermined capacitance may be set such that a primary side parallel resonance frequency governed by the inductance of the choke coil, the leakage inductance occurring at the primary winding, and the predetermined capacitance is substantially twice the primary side first series resonance frequency. The active clamping circuit may include a voltage clamping capacitor and an auxiliary switching element connected in series with the voltage clamping capacitor. The auxiliary switching element is turned on and off complementarily to the main switching element, whereby a voltage applied to the main switching element is clamped.
The power factor improving section may add and pass a current corresponding to a voltage generated in the primary side series resonant capacitor to the smoothing capacitor via the primary side rectifying element. Thereby the power factor of the switching power supply circuit as a load for the alternating-current power supply is improved.
The power factor improving section of the switching power supply circuit may have an inductor connected between the primary side rectifying element and the smoothing capacitor, and another terminal of the primary side series resonant capacitor is connected to a point of connection between the inductor and the primary side rectifying element.
The choke coil may be used as the inductor of the power factor improving section, whereby the number of parts is reduced.
The choke coil may be formed as a leakage inductance occurring at a primary winding of a choke transformer. The choke transformer may be formed with the primary winding and a secondary winding magnetically loosely coupled with each other, and the current corresponding to the voltage generated in the primary side series resonant capacitor flows via the secondary winding of the choke transformer. Thereby the power factor of the switching power supply circuit as a load for the alternating-current power supply is improved.
The switching power supply circuit according to one embodiment of the present invention can be provided with a power factor improving function without the active filter. By omitting the active filter, the power conversion efficiency characteristics of the switching power supply circuit are improved. Then, a radiator and the like can be omitted or reduced in size. In addition, as compared with the configuration including the active filter, the number of parts is also reduced greatly. Further, one switching element is used to deal with high power, so that the circuit is reduced in size, weight, and cost. In addition, while the active filter performs hard switching operation, the switching converter according to one embodiment of the present invention, which may be based on a resonant converter, may perform soft switching operation. This greatly reduces switching noise, and thus contributes to reductions in size, weight, and cost of a noise filter. Further, because a plurality of clocks of different frequencies may not exist, a problem of mutual interference due to the plurality of clock frequencies does not occur, reliability is improved, and circuit board pattern design and the like are made easier. Further, the withstand voltage of the switching element can be lowered.
Prior to description of the best mode for carrying out the present invention (hereinafter referred to as an embodiment), a fundamental configuration of a switching converter that performs a switching operation of a class E resonance type (hereinafter referred to also as a class E switching converter) will first be described with reference to
The class E switching converter shown in
The drain of the switching element Q1 is connected to the positive electrode of a direct-current input voltage Ein via a series connection of a choke coil L10. The source of the switching element Q1 is connected to the negative electrode of the direct-current input voltage Ein. In addition, the drain of the switching element Q1 is connected with one terminal of a choke coil L11. A series resonant capacitor C11 is connected in series with another terminal of the choke coil L11. An impedance Z as a load is inserted between the series resonant capacitor C11 and the negative electrode of the direct-current input voltage Ein. The impedance Z in this case is obtained by converting a load on a secondary side to a load on a primary side.
Since the inductance of the choke coil L10 is set considerably higher than the inductance of the choke coil L11, the class E switching converter having such a configuration can be considered to be a form of a complex resonant converter having a parallel resonant circuit formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonant capacitor Cr and a series resonant circuit formed by the inductance of the choke coil L11 and the capacitance of the series resonant capacitor C11. Also, the class E switching converter having such a configuration can be said to be the same as a voltage resonant converter of a single-ended type in that the switching converter has one switching element.
A switching voltage V1 is obtained across the switching element Q1. The switching voltage V1 is at a zero level during an on period TON during which the switching element Q1 is on, and forms a sinusoidal pulse waveform during an off period TOFF during which the switching element Q1 is off. This switching pulse waveform is obtained by the resonant operation (voltage resonant operation) of the above parallel resonant circuit.
A switching current IQ1 flows through the switching element Q1 and the body diode DD. The switching current IQ1 is at a zero level during the period TOFF. For a certain period from a start of the on period TON, the switching current IQ1 first flows through the body diode DD, and is therefore of negative polarity. Then the switching current IQ1 is inverted to be of positive polarity, and flows from the drain to the source of the switching element Q1.
A current I2 flowing through the class E switching converter transformer is obtained by combining the switching current IQ1 flowing through the switching element Q1 (and the body diode DD) and a current flowing through the primary side parallel resonant capacitor Cr. The current I2 has a waveform including a sinusoidal wave component.
A relation between the switching current IQ1 and the switching voltage V1 indicates that ZVS operation is obtained in timing of turning off the switching element Q1, and that ZVS operation and ZCS operation are obtained in timing of turning on the switching element Q1.
Since the inductance of the choke coil L10 is set higher than the inductance of the choke coil L11 as described above, a current I1 flowing into the class E switching converter so as to flow from the positive electrode terminal of the direct-current input voltage Ein through the choke coil L10 forms a pulsating current waveform having a predetermined average level as shown in
The inductance of the choke coil L10 is set higher than the inductance of the choke coil L11, a stable ZVS operation is obtained in timing of turning off the switching element Q1, and stable ZVS and ZCS operations are obtained in timing of turning on the switching element Q1, as described above. With the fact as a starting point, in order to reduce device size and price, the inventor listed in the present application (abbreviated as the present inventor) created a modified circuit of a class E switching converter in which the inductance of the choke coil L10 is reduced, and obtained a circuit configuration in which stable ZVS operation can be performed in timings of turning off and turning on the switching element Q1.
Specifically, a converter part (switching converter) according to the creation of the present inventor has a voltage resonant circuit and two current resonant circuits on the primary side. One of the current resonant circuits is a primary side first series resonant circuit whose primary side first series resonance frequency is governed by the inductance of a choke coil and a primary side series resonant capacitor. The other is a primary side second series resonant circuit whose primary side second series resonance frequency is governed by a leakage inductance occurring at the primary winding of a converter transformer and the primary side series resonant capacitor. The primary side first series resonance frequency is set at substantially twice the primary side second series resonance frequency. Thereby excellent stable ZVS operation is obtained even when the value of the inductance of the choke coil is low. In this case, substantially twice a frequency includes a range of 20 percent with twice the frequency as a center.
Further, the present inventor combined a power factor improving part with the converter part, and thereby achieved a switching power supply circuit also exhibiting an excellent power factor improving characteristic. Specifically, the power factor improving part may be configured to pass a part of resonance current flowing though the primary side first series resonant circuit and a part of resonance current flowing though the primary side second series resonant circuit from an alternating-current power supply via a primary side rectifying element. Alternatively, the power factor improving part may pass a current corresponding to a voltage generated in the primary side series resonant capacitor from the alternating-current power supply via the primary side rectifying element. Concrete circuit configurations of these parts will hereinafter be described as embodiments.
As the present embodiment, a modification of the above-described class E switching converter is applied to a power supply circuit. An outline of a switching power supply circuit according to a first embodiment shown in
The rectifying and smoothing section is formed with the primary side rectifying element Di, which is supplied with the input alternating voltage from the alternating-current power supply AC and rectifies the input alternating voltage, and the smoothing capacitor Ci. The input alternating voltage from the alternating-current power supply AC is input to the input side of the primary side rectifying element Di, and one terminal of the output side of the primary side rectifying element Di is connected to the smoothing capacitor Ci, whereby the primary side direct-current voltage is generated.
While the primary side has the configuration of a voltage and current resonant converter performing class E switching operation, the primary side has different connections from those of the class E switching converter shown in
More specifically, when viewed as a resonant converter, this multiple resonant converter section includes the choke coil PCC having one terminal connected to one terminal of the smoothing capacitor Ci and the converter transformer PIT wound with the primary winding N1 and the secondary winding N2 loosely coupled with each other. One terminal of the primary winding N1 is connected to another terminal of the choke coil PCC. Another terminal of the primary winding N1 of the converter transformer PIT (a simple abbreviation of the primary winding N1 will also be used hereinafter) is connected to one terminal of the switching element Q1, whereby alternating-current power is supplied to the converter transformer PIT. Then, the multiple resonant converter section includes: the primary side first series resonant circuit of the current resonant type whose primary side first series resonance frequency as resonance frequency is governed by the inductance L3 possessed by the choke coil PCC and the capacitance of the primary side series resonant capacitor C2 connected to the other terminal of the choke coil PCC and the one terminal of the primary winding N1; the primary side second series resonant circuit of the current resonant type whose primary side second series resonance frequency as resonance frequency is governed by the leakage inductance L1 occurring at the primary winding N1 and the capacitance of the primary side series resonant capacitor C2; and a primary side parallel resonant circuit of the voltage resonant type whose resonance frequency is governed by the leakage inductance L1 occurring at the primary winding N1, the inductance L3 possessed by the choke coil PCC, and a primary side parallel resonant capacitor Cr connected in parallel with the switching element Q1.
In this case, a ratio of the primary side first series resonance frequency to the primary side second series resonance frequency is set at substantially 2:1. That is, when the primary side second series resonance frequency is a reference frequency, the primary side first series resonance frequency is set at substantially twice the primary side second series resonance frequency. When the primary side first series resonance frequency is a reference frequency, the primary side second series resonance frequency is set at substantially ½ of the primary side first series resonance frequency. The present inventor has found that this value is important in relation to the effect of ZVS operation in timings of turning off and turning on the switching element Q1, and that a load power variable range where ZVS operation is obtained is narrowed as the ratio of the primary side first series resonance frequency to the primary side second series resonance frequency deviates from the above-mentioned value. In this case, substantially twice a frequency and substantially ½ of a frequency include a range of 20 percent with twice the frequency and ½ of the frequency as a center. Incidentally, the frequency of the primary side parallel resonant circuit is substantially twice the primary side first series resonance frequency.
In addition, the multiple resonant converter section includes the oscillating and driving circuit 2 for on-off driving the switching element Q1 and the control circuit 1. The control circuit 1 supplies the oscillating and driving circuit 2 with a control signal to make the value of the secondary side direct-current output voltage Eo output by the secondary side rectifier circuit connected to the secondary winding N2 of the converter transformer PIT (a simple abbreviation of the secondary winding N2 will also be used hereinafter) a predetermined value. The secondary side rectifier circuit connected to the secondary winding N2 has a secondary side series resonant capacitor C4 to form a secondary side series resonant circuit.
The control circuit 1 supplies the oscillating and driving circuit 2 with a detection output corresponding to a difference between the input secondary side direct-current output voltage Eo and a predetermined reference voltage value. The oscillating and driving circuit 2 drives the switching element Q1 so as to mainly change switching frequency according to the input detection output of the control circuit 1. In addition to the switching frequency, the oscillating and driving circuit 2 may change a time ratio as a ratio of the on period of the switching element Q1 in one cycle.
Thus variably controlling the switching frequency of the switching element Q1 changes the resonant impedance of the primary side and the secondary side in the power supply circuit, and changes an amount of power transmitted from the primary winding N1 to the secondary winding N2 side of the converter transformer PIT and an amount of power to be supplied from the secondary side rectifier circuit to the load. Thereby an operation of matching the magnitude of the secondary side direct-current output voltage Eo with the reference voltage is obtained. That is, the secondary side direct-current output voltage Eo is stabilized.
In addition, the active clamping circuit is formed by the series circuit of the voltage clamping capacitor Cc and the auxiliary switching element Q2. In this case, the auxiliary switching element Q2 is turned on complementarily to the switching element Q1. That is, the auxiliary switching element Q2 is not turned on while the switching element Q1 is on (conducting), and the switching element Q1 is not turned on while the auxiliary switching element Q2 is on.
Further, the switching power supply circuit according to the present embodiment has the power factor improving section. The power factor improving section includes: the primary side rectifying element Di functioning as a rectifying element for making a current obtained by adding together the resonance currents flowing through the primary side first series resonant circuit and the primary side second series resonant circuit flow in one direction from the alternating-current power supply AC to the smoothing capacitor Ci; and the power factor improving inductor Lo.
One terminal of the power factor improving inductor Lo is connected to one terminal of the output side of the primary side rectifying element Di and the primary side series resonant capacitor C2. Another terminal of the power factor improving inductor Lo is connected to the smoothing capacitor Ci.
In addition, a filter capacitor CN is connected to the input side of the primary side rectifying element Di. This filter capacitor CN is to suppress normal mode noise, thereby making it possible to prevent a radiation component occurring according to the switching of the switching element Q1 from flowing out to the alternating-current power supply AC side.
The secondary side rectifier circuit of the switching power supply circuit according to the present embodiment is formed as a full-wave rectifier circuit by connecting the secondary side rectifying element Do operating at high speed and the smoothing capacitor Co to the secondary winding N2 with which the secondary side series resonant capacitor C4 is connected in series. That is, a positive current and a negative current flow through the secondary side series resonant capacitor C4 in a switching cycle, and the secondary side series resonant capacitor C4 functions as a part of a resonant circuit without being charged with a charge of either polarity. That is, the secondary side rectifier circuit forms a secondary side series resonant circuit whose series resonance frequency is governed by the leakage inductance L2 of the secondary winding N2 and the secondary side series resonant capacitor C4. Incidentally, the secondary side rectifier circuit may be not only a rectifier circuit generating a voltage once the voltage generated in the secondary winding N2 but also a voltage doubler rectifier circuit generating a voltage twice the voltage generated in the secondary winding N2. Further, as for the secondary side resonant circuit, not only may a series resonant circuit be formed to make a multiple converter, but also a partial voltage resonant circuit or a parallel resonant circuit may be formed to make a multiple converter. Further, a secondary side rectifier circuit may be provided without a resonant circuit being formed on the secondary side. The various examples of modification of these secondary side rectifier circuits will be described later.
The switching power supply circuit according to the embodiment shown in
Alternating voltage passed through the common mode noise filter is supplied to the input side of the primary side rectifying element Di formed by bridge connection of four fast type rectifying elements (diodes), and then rectified by the primary side rectifying element Di, whereby a pulsating voltage is generated. The pulsating voltage is applied to the smoothing capacitor Ci via the power factor improving inductor Lo. A direct-current input voltage Ei as direct-current voltage having a voltage value around the peak value of the pulsating voltage is maintained across the smoothing capacitor Ci. A reason of forming the primary side rectifying element Di by the bridge connection of the fast type rectifying elements (diodes) is for circuit simplification. The primary side rectifying element Di functions as a rectifying element for passing the resonance currents flowing through the primary side first series resonant circuit and the primary side second series resonant circuit in one direction to form a part of the power factor improving section. That is, the power factor improving section passes a component in one direction as part of the resonance currents flowing in both directions from the commercial alternating-current power supply AC via the primary side rectifying element Di, and thereby improves the power factor.
The direct-current input voltage Ei has a level corresponding to once an alternating input voltage VAC. The direct-current input voltage Ei is a direct-current input voltage for the class E switching converter in a subsequent stage.
The multiple resonant converter section functions as modified class E switching converter in substantially the same manner as the class E switching converter. The multiple resonant converter section is formed with the choke coil PCC, the converter transformer PIT, the primary side series resonant capacitor C2, the primary side parallel resonant capacitor Cr, and the switching element Q1 as main parts. Correspondences between parts of the class E switching converter whose principles have been described with reference to
That is, in the first embodiment shown in
The primary winding N1 and the secondary winding N2 of the converter transformer PIT are loosely coupled with each other at a coupling coefficient of 0.82. Therefore the primary winding N1 has the leakage inductance L1. The primary side first series resonant circuit whose primary side first series resonance frequency is governed by the inductance L3 of the choke coil PCC and the capacitance of the primary side series resonant capacitor C2 is formed. In addition, the primary side second series resonant circuit whose primary side second series resonance frequency is governed by the leakage inductance L1 and the capacitance of the primary side series resonant capacitor C2 is formed. In addition, the primary side parallel resonant circuit whose primary side parallel resonance frequency is governed by the leakage inductance L1, the inductance L3 of the choke coil PCC, and the capacitance of the primary side parallel resonant capacitor Cr is formed.
The resonance frequency being “governed” means that the resonance frequency is determined mainly by these elements. For example, although the primary side first series resonance frequency, the primary side second series resonance frequency, and the primary side parallel resonance frequency are affected by the inductance component of the power factor improving inductor Lo, the smoothing capacitor Ci, and the like, the inductance component of the power factor improving inductor Lo, the smoothing capacitor Ci, and the like produce less effect on the primary side series resonance frequencies and the primary side parallel resonance frequency.
Specifically, the primary side first series resonant circuit forms a current path in one direction from a ground point of the primary side rectifying element Di through anodes and cathodes of each of two sets of two fast rectifying elements connected in series with each other to the primary side series resonant capacitor C2 to the choke coil PCC to the smoothing capacitor Ci. The primary side first series resonant circuit forms a current path in another direction from the primary side series resonant capacitor C2 to the power factor improving inductor Lo to the choke coil PCC. The primary side second series resonant circuit forms a current path in one direction from the ground point of the primary side rectifying element Di through the anodes and cathodes of each of the two sets of two fast rectifying elements connected in series with each other to the primary side series resonant capacitor C2 to the primary winding N1 and then from the drain to the source of the switching element Q1. The primary side second series resonant circuit forms a current path in another direction from the primary side series resonant capacitor C2 to the power factor improving inductor to the smoothing capacitor Ci to a body diode DD1 to the primary winding N1.
In addition, as described above, the secondary winding N2 of the converter transformer is connected to the secondary side series resonant capacitor C4. The secondary side series resonant circuit whose resonance frequency is governed by the leakage inductance component (represented by the inductance L2 in
The switching element Q1 supplying alternating voltage to the primary side series resonant circuits and the primary side parallel resonant circuit is connected to the other terminal of the primary winding N1. In this case, the oscillating and driving circuit 2 drives the switching element Q1. Thus, the primary side operates as a converter performing modified class E switching operation and has the configuration of a voltage and current resonant converter, and the secondary side has a current resonant converter, whereby, as a whole, a multiple resonant converter that holds the value of secondary side direct-current output voltage Eo constant is formed. That is, the switching power supply circuit according to the present embodiment has the multiple resonant converter section including the primary side series resonant circuits, the primary side parallel resonant circuit, and the secondary side series resonant circuit, as viewed in terms of alternating current.
The active clamping circuit will next be described. The active clamping circuit is inserted between the switching element Q1 and the smoothing capacitor Ci. The active clamping circuit clamps a voltage across the switching element Q1.
The active clamping circuit is formed by a series connection circuit of the voltage clamping capacitor Cc and the auxiliary switching element Q2. A MOS-FET having a body diode DD2 is used as the auxiliary switching element Q2. The gate of the MOS-FET is supplied with a voltage obtained by dividing a voltage from a control winding Ng by a resistance Rg1 and a resistance Rg2. The control winding Ng is wound in the converter transformer PIT in a direction such that the switching element Q1 and the auxiliary switching element Q2 are turned on complementarily to each other. The value of the resistance Rg1 is 220Ω, and the value of the resistance Rg2 is 100Ω. A voltage generated in the control winding Ng has a near-sinusoidal waveform. Therefore, by changing a ratio between the resistance Rg1 and the resistance Rg2, it is possible to adjust a voltage between the gate and the source of the auxiliary switching element Q2, and adjust the length of a time during which the voltage generated in the switching element Q1 is clamped. That is, as the value of the resistance Rg1 is decreased as compared with the value of the resistance Rg2, the time during which the auxiliary switching element Q2 is on is lengthened.
The capacitance of the voltage clamping capacitor Cc is sufficiently large so that a voltage across the voltage clamping capacitor Cc is scarcely changed by an amount of charge of a current flowing through the auxiliary switching element. Thus, while the auxiliary switching element Q2 is on, that is, while a current is flowing through the auxiliary switching element, the voltage applied to the switching element Q1 is clamped. Therefore, the peak part of the voltage which should inherently form a sinusoidal wave shape is flattened. Then, the withstand voltage of the switching element Q1 can be lowered.
Further, it has been confirmed by experiment that a range where ZVS characteristics are exhibited is expanded by passing a current through the auxiliary switching element Q2. That is, the active clamping circuit can widen the range of ZVS characteristics.
In addition, the auxiliary switching element Q2 is controlled by the control winding Ng, and the voltage generated in the control winding Ng is controlled by the switching element Q1. Thus the auxiliary switching element Q2 operates in synchronism with the switching element Q1. Therefore a problem of for example occurrence of an undesired beat resulting from a plurality of switching frequencies being mixed (interference between a plurality of frequencies) does not occur.
The action of the power factor improving section will next be described. A power factor improving effect is produced by connecting the power factor improving inductor Lo and the primary side rectifying element Di to the primary side series resonant capacitor C2. The resonance current flowing through the primary side first series resonant circuit and the primary side second series resonant circuit flows through the primary side series resonant capacitor C2. A current in one direction of the resonance current flowing through the primary side series resonant capacitor C2 flows from the alternating-current power supply AC via the cathodes of two rectifying elements forming the primary side rectifying element Di, is added to rectified current, and then flows as current I1. In this case, the power factor improving inductor Lo having a high impedance for this resonance current prevents resonance current from flowing from the smoothing capacitor Ci to the primary side series resonant capacitor C2.
A current in another direction of the resonance current is difficult to flow through the cathodes of the two rectifying elements forming the primary side rectifying element Di, but is added, as current I2 flowing to the smoothing capacitor Ci, to the current from the primary side rectifying element Di. As described above, in order to have an effect of changing a path through which the resonance current passes according to the flowing direction of the current, the primary side rectifying element Di needs to be formed by a fast rectifying element having a sufficient switching capability for a component having the cycle of the resonance current flowing through the primary side first series resonant circuit and the primary side second series resonant circuit and a harmonic having an integral multiple of the cycle. Otherwise, there occur not only an increase in switching loss and a decrease in efficiency of the switching power supply circuit but also destruction of the primary side rectifying element Di due to heat loss in some cases.
Thus, the current in one direction as part of the resonance current which is included in the resonance current flows through the primary side rectifying element Di, whereby the flow angle of an alternating input current IAC is increased, and the power factor is improved. That is, when the primary side series resonant capacitor C2 is not connected to the primary side rectifying element Di, the current I1 has a waveform in the shape of a pulse flowing around the peak of a voltage V2. However, the resonance current in one direction included in the resonance current also flows outside the vicinity of the peak of the voltage V2, whereby the conduction angle is increased. Thus, in the present embodiment, the resonance current flowing through the primary side first series resonant circuit and the primary side second series resonant circuit is fed back to the smoothing capacitor Ci, and thereby a power regenerative type power factor improving section is formed.
Further, description will be made of concrete constitutions of details of important parts in the switching power supply circuit according to the embodiment shown in
Details of the converter transformer PIT will first be described. The converter transformer PIT has functions of insulating the primary side and the secondary side from each other and performing voltage conversion. Further, the converter transformer PIT also functions as the inductance L1 forming a part of the resonant circuits for making the modified class E switching converter of the multiple resonant type function. The inductance L1 is the leakage inductance component formed by the converter transformer PIT. A concrete structure of the converter transformer PIT will be described with reference to
The converter transformer PIT has an EE type core (EE-shaped core) formed by combining an E-type core CR1 and an E-type core CR2 of ferrite material with each other such that magnetic legs of the E-type core CR1 are opposed to magnetic legs of the E-type core CR2. A primary side winding part and a secondary side winding part are divided from each other so as to be independent of each other, and a bobbin B formed by a resin, for example, is provided. Then, the bobbin B wound with the primary winding N1, the control winding Ng, and the secondary winding N2 is attached to the EE-shaped core. Thereby the primary winding N1, the control winding Ng, and the secondary winding N2 are wound around the central magnetic leg of the EE-shaped core with the primary winding N1 and the control winding Ng wound in one region and with the secondary winding N2 separated in a winding region different from the one region. The structure of the converter transformer PIT as a whole is thus obtained.
A gap G of 1.2 mm is formed in the central magnetic leg of the EE-shaped core. Thereby 0.82 is obtained as the value of a coupling coefficient k between the primary side and the secondary side. Thus, the leakage inductance L1 having a high inductance value is obtained. Incidentally, the gap G is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs of the E-type core CR1 and the E-type core CR2. The number of turns of the primary winding N1 is 45 T (turns). The number of turns of the secondary winding N2 is 30 T (turns). The number of turns of the control winding Ng is 1 T (turn). A core material is EER-35 (the name of the core material).
The value of the primary side parallel resonant capacitor Cr is 1000 pF. The value of the primary side series resonant capacitor C2 is 0.1 μF. The value of the voltage clamping capacitor Cc is 0.1 μF. The value of the secondary side series resonant capacitor C4 is 0.056 μF. The value of the filter capacitor CN is 1 μF.
Both the choke coil PCC and the power factor improving inductor Lo can employ substantially the same configuration as the converter transformer PIT. The value of the inductance L3 of the choke coil PCC is 82 μH. The value of the inductance of the power factor improving inductor Lo is 82 μH.
A rectifying element meeting 3 A/600 V withstand voltage specifications is used as the primary side rectifying element Di. A rectifying element meeting 5 A/200 V withstand voltage specifications is used as the secondary side rectifying element Do. Both are formed by a fast rectifying element.
On the secondary side of the converter transformer PIT, a voltage waveform similar to an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. The secondary side series resonant capacitor C4 is connected in series with the secondary winding N2. Thereby the secondary side series resonant circuit is formed by the leakage inductance L2 as viewed from the secondary winding N2 side and the secondary side series resonant capacitor C4. In the present embodiment, the resonance frequency of the secondary side series resonant circuit is set substantially equal to the primary side series resonance frequency governed by the primary side series resonant capacitor C2, the leakage inductance L1, and the inductance L3 of the choke coil PCC described above. However, the resonance frequency of the secondary side series resonant circuit can be set suitably in relation to the primary side series resonance frequency. Further, a partial voltage resonant circuit may be provided on the secondary side without the secondary side series resonant circuit being provided.
A MOS-FET is selected as the switching element Q1, as described above. The switching element Q1 includes the body diode DD1 in parallel with the source and drain of the switching element Q1.
The control circuit 1 supplies the oscillating and driving circuit 2 with a detection output corresponding to a difference between the input secondary side direct-current output voltage Eo and a predetermined reference voltage value. The oscillating and driving circuit 2 drives the switching element Q1 so as to mainly change switching frequency according to the input detection output of the control circuit 1. In addition to the switching frequency, the oscillating and driving circuit 2 may change a time ratio as a ratio of the on period of the switching element Q1 in one cycle.
Thus variably controlling the switching frequency of the switching element Q1 changes the resonant impedance of the primary side and the secondary side in the power supply circuit, and changes an amount of power transmitted from the primary winding N1 to the secondary winding N2 side of the converter transformer PIT and an amount of power to be supplied from the secondary side rectifier circuit to the load. Thereby an operation of matching the magnitude of the secondary side direct-current output voltage Eo with the reference voltage is obtained. That is, the secondary side direct-current output voltage Eo is stabilized. In this case, the value of the secondary side direct-current output voltage Eo is 175 V.
The configuration and action of the switching power supply circuit according to the present embodiment has been described above.
Introducing a part of representative characteristics that can be read from
When the value of the alternating input voltage VAC is 230 V, the value of the power factor PF at a load power Po of 300 W is 0.930, and the value of the power factor PF at a load power Po of 75 W is 0.76, these values representing a high power factor. When the value of the alternating input voltage VAC is 230 V, the value of the power conversion efficiency ηAC→DC at a load power Po of 300 W is high at 93.5%. In the range where the load power Po is 300 W to 0 W, the value of the direct-current input voltage Ei is 353 V to 374 V. The above values of the power factor in both cases where the value of the alternating input voltage VAC is 100 V and 230 V exceed a power factor of 0.75 as a standard value for Class D of televisions and the like as defined in IEC 61000-3-2, which is an international standard related to harmonic current suppression.
Such a switching power supply circuit according to the present embodiment has improved the power conversion efficiency ηAC→DC as compared with the switching power supply circuit shown as background art in
In addition, in the switching power supply circuit according to the embodiment, the operation of the multiple resonant converter section and the power factor improving section is a so-called soft switching operation, and therefore the level of switching noise is greatly reduced as compared with the circuit using the active filter shown in
Further, the switching circuit according to the embodiment has the primary side series resonant circuits and the primary side parallel resonant circuit as well as the secondary side series resonant circuit. Therefore, the secondary side direct-current output voltage Eo can be maintained at a predetermined voltage level by very slight changes in frequency, and the noise filter can be designed easily. For this reason, the noise filter in one stage composed of one common mode choke coil CMC and two across capacitors CL suffices to meet power supply interference standards. In addition, a sufficient measure against the normal mode noise of the rectification output line can be taken by the single filter capacitor CN.
Further, rectifier diodes Do1 to Do4 on the secondary side and the like operate in synchronism with the switching element Q1. Therefore ground potential interference between the active filter side and the switching converter in the subsequent stage as in the power supply circuit of
Further, by providing the active clamping circuit, it is possible to lower the withstand voltage of the switching element Q1 and reduce a range of variation of the ratio TON/TOFF between the on period TON and the off period TOFF. Further, it is possible to lower the withstand voltage of the switching element Q1, expand a ZVS range, and achieve so-called range widening that allows the value of the alternating input voltage VAC to range widely from 85 V to 264 V.
In a switching power supply circuit according to a second embodiment shown in
With such a configuration, the number of fast rectifying elements used for power factor improvement can be decreased from four in the first embodiment to one, thereby reducing cost. In addition, the filter capacitor CN is disposed in a stage subsequent to the primary side rectifying element Di as viewed from the side of an alternating-current power supply AC, so that an ordinary part not approved by a safety standard can be used as the filter capacitor CN. The constitutions of the other parts do not differ from those of the first embodiment, and produce actions and effects that are substantially the same as in the first embodiment.
In a switching power supply circuit according to a third embodiment shown in
Specifically, the switching power supply circuit according to the third embodiment includes a rectifying and smoothing section, a converter section, and a power factor improving section. The rectifying and smoothing section converts input alternating-current power from an alternating-current power supply AC into primary side direct-current power. The converter section converts the primary side direct-current power from the rectifying and smoothing section into alternating-current power and further converts the alternating-current power into secondary side direct-current power. The power factor improving section improves a power factor. The rectifying and smoothing section includes a primary side rectifying element Di, which is supplied with the input alternating-current power from the alternating-current power supply AC and rectifies the input alternating-current power, and a smoothing capacitor Ci. The converter section includes: a choke coil PCC; a converter transformer PIT; a switching element Q1; an oscillating and driving circuit 2; a control circuit 1; a primary side first series resonant circuit; a primary side second series resonant circuit; and an active clamping circuit. The choke coil PCC is supplied with the primary side direct-current power from the smoothing capacitor Ci. The converter transformer PIT has a primary winding N1 supplied with the power from the choke coil PCC and a secondary winding N2 magnetically loosely coupled with the primary winding N1. The switching element Q1 supplies the alternating-current power to the primary winding N1. The oscillating and driving circuit 2 on-off drives the switching element Q1. The control circuit 1 supplies the oscillating and driving circuit 2 with a control signal to make the value of a secondary side direct-current output voltage Eo output by a secondary side rectifying element Do and a smoothing capacitor Co forming a secondary side rectifier circuit connected to the secondary winding N2 a predetermined value. The primary side first series resonant circuit is governed primary side first series resonance frequency by an inductance L3 possessed by the choke coil PCC and a primary side series resonant capacitor C2. The primary side second series resonant circuit is governed primary side second series resonance frequency by a leakage inductance L1 occurring at the primary winding N1 and the primary side series resonant capacitor C2. The primary side first series resonance frequency is set at substantially twice the primary side second series resonance frequency. The active clamping circuit clamps a voltage applied to the switching element Q1. The active clamping circuit is a series circuit of a voltage clamping capacitor Cc and an auxiliary switching element Q2. In this case, the auxiliary switching element Q2 is turned on complementarily to the switching element Q1. For the power factor improving section to pass a current corresponding to a voltage generated in the primary side series resonant capacitor C2 from the alternating-current power supply via the primary side rectifying element Di (fast rectifying element), the above primary side rectifying element Di is formed by a fast rectifying element.
In the following, description will be made of characteristic parts of the third embodiment which are different from the first embodiment and the second embodiment, and description of similar parts to those of the first embodiment and the second embodiment will be omitted.
Description will first be made of the primary side first series resonant circuit and the primary side second series resonant circuit. The primary side first series resonant circuit forms a current path from the choke coil PCC through the smoothing capacitor Ci and a ground point to the primary side series resonant capacitor C2. The primary side second series resonant circuit forms a current path in one direction from the primary side series resonant capacitor C2 through the primary winding N1 and the drain and the source of the switching element Q1 to the ground point. The primary side second series resonant circuit forms a current path in another direction from the primary side series resonant capacitor C2 through a body diode DD1 to the primary winding N1.
A current flowing through the power factor improving section for power factor improvement is a part of resonance current flowing through the primary side first series resonant circuit and the primary side second series resonant circuit, which part is superimposed on a current I1. That is, by connecting the primary side series resonant capacitor C2 to the primary side rectifying element Di, a resonance current corresponding to a resonance voltage occurring in the primary side series resonant capacitor C2 is passed from the alternating-current power supply AC via the anodes and cathodes of two fast rectifying elements. A current resulting from adding together the resonance current for power factor improvement and a current obtained by rectifying commercial alternating-current power from the alternating-current power supply AC flows as the current I1.
That is, when the power factor improving section is not provided, the current I1 has a waveform in the shape of a pulse flowing around the peak of a voltage V2. However, the resonance current corresponding to the resonance voltage occurring in the primary side series resonant capacitor C2 is also passed outside the vicinity of the peak of the voltage V2, whereby a conduction angle is increased. Thus, in the present embodiment, the resonance voltage occurring in the primary side series resonant capacitor C2 is fed back to the smoothing capacitor Ci via the choke coil PCC, and thereby a voltage feedback type power factor improving section is formed.
The concrete constants of parts in the third embodiment are set as follows. The secondary side direct-current output voltage Eo is 175 V. The period TOFF of the switching element Q1 is changed according to variation in load power Po, the period TON of the switching element Q1 is reduced with decrease in load power Po and increase in alternating input voltage VAC, and the value of the secondary side direct-current output voltage Eo is made constant by increasing switching frequency.
A ferrite material of the converter transformer PIT is EER-35. A gap of the converter transformer PIT is 1.2 mm. The coupling coefficient of the converter transformer PIT is 0.82. The number of turns of the primary winding N1 is 45 T (turns). The number of turns of the secondary winding N2 is 30 T (turns). The number of turns of a control winding Ng is 1 T (turn).
The value of a primary side parallel resonant capacitor Cr is 1000 pF. The value of the primary side series resonant capacitor C2 is 0.1 μF. The value of the voltage clamping capacitor Cc is 0.1 μF. The value of the secondary side series resonant capacitor C4 is 0.056 μF. The value of the filter capacitor CN is 1 μF.
Both the choke coil PCC and the power factor improving inductor Lo can employ substantially the same configuration as the converter transformer PIT. The value of the inductance L3 of the choke coil PCC is 82 μH.
A rectifying element meeting 3 A/600 V withstand voltage specifications is used as the primary side rectifying element Di. A rectifying element meeting 5 A/200 V withstand voltage specifications is used as the secondary side rectifying element Do. Both are formed by a fast rectifying element.
The waveforms of principal parts in the third embodiment are substantially the same as in
Introducing a part of representative characteristics that can be read from
When the value of the alternating input voltage VAC is 230 V, the value of the power factor PF at a load power Po of 300 W is 0.945, and the value of the power factor PF at a load power Po of 75 W is 0.77, these values representing a high power factor. When the value of the alternating input voltage VAC is 230 V, the value of the power conversion efficiency ηAC→DC at a load power Po of 300 W is high at 91.7%. In the range where the load power Po is 300 W to 0 W, the value of the direct-current input voltage Ei is 360 V to 375 V. The above values of the power factor in both cases where the value of the alternating input voltage VAC is 100 V and 230 V exceed a power factor of 0.75 as a standard value for Class D defined in IEC 61000-3-2.
Such a switching power supply circuit according to the present embodiment has improved the power conversion efficiency ηAC→DC as compared with the switching power supply circuit shown as background art in
In addition, in the switching power supply circuit according to the embodiment, the operation of the multiple resonant converter section and the power factor improving section is a so-called soft switching operation, and therefore the level of switching noise is greatly reduced as compared with the circuit using the active filter shown in
Further, the switching circuit according to the embodiment has the primary side series resonant circuits and the primary side parallel resonant circuit as well as the secondary side series resonant circuit. Therefore, the secondary side direct-current output voltage Eo can be maintained at a predetermined voltage level by very slight changes in frequency, and the noise filter can be designed easily. For this reason, the noise filter in one stage composed of one common mode choke coil CMC and two across capacitors CL suffices to meet power supply interference standards. In addition, a sufficient measure against the normal mode noise of the rectification output line can be taken by the single filter capacitor CN.
Further, rectifier diodes Do1 to Do4 on the secondary side and the like operate in synchronism with the switching element Q1. Therefore ground potential interference between the active filter side and the switching converter in the subsequent stage as in the power supply circuit of
Further, by employing the active clamping circuit, a range of variation of the ratio TON/TOFF between the on period TON and the off period TOFF is reduced as compared with a case where the active clamping circuit is not provided. Further, it is possible to lower the withstand voltage of the switching element Q1, expand a ZVS range, and achieve so-called range widening that allows the value of the alternating input voltage VAC to range widely from 85 V to 264 V.
In a switching power supply circuit according to a fourth embodiment shown in
The converter section in the fourth embodiment is the same as in the first to third embodiments in that ZVS operation is performed with the resonance frequency of the primary side first series resonant circuit set at substantially twice the resonance frequency of the primary side second series resonant circuit.
In a switching power supply circuit according to a fifth embodiment shown in
Specifically, the switching power supply circuit according to the fifth embodiment includes a rectifying and smoothing section, a converter section, and a power factor improving section. The rectifying and smoothing section converts input alternating-current power from an alternating-current power supply AC into primary side direct-current power. The converter section converts the primary side direct-current power from the rectifying and smoothing section into alternating-current power and further converts the alternating-current power into secondary side direct-current power. The power factor improving section improves a power factor. The rectifying and smoothing section includes a primary side rectifying element Di, which is supplied with the input alternating-current power from the alternating-current power supply AC and rectifies the input alternating-current power, and a smoothing capacitor Ci. The converter section includes the choke coil primary winding NC1, a converter transformer PIT, an oscillating and driving circuit 2, a control circuit 1, a primary side first series resonant circuit, a primary side second series resonant circuit, and an active clamping circuit. The choke coil primary winding NC1 of the choke transformer VFT is supplied with the primary side direct-current power from the smoothing capacitor Ci. The converter transformer PIT has a primary winding Ni supplied with the power from the choke coil primary winding NC1 and a secondary winding N2 magnetically loosely coupled with the primary winding N1. The switching element Q1 supplies the alternating-current power to the primary winding N1. The oscillating and driving circuit 2 on-off drives the switching element Q1. The control circuit 1 supplies the oscillating and driving circuit 2 with a control signal to make the value of a secondary side direct-current output voltage Eo output by a secondary side rectifying element Do and a smoothing capacitor Co forming a secondary side rectifier circuit connected to the secondary winding N2 a predetermined value. The primary side first series resonant circuit is governed primary side first series resonance frequency by an inductance L3 occurring at the choke coil primary winding NC1 and a primary side series resonant capacitor C2. The primary side second series resonant circuit is governed primary side second series resonance frequency by a leakage inductance L1 occurring at the primary winding N1 and the primary side series resonant capacitor C2. The primary side first series resonance frequency is set at substantially twice the primary side second series resonance frequency. The active clamping circuit clamps a voltage applied to the switching element Q1. The active clamping circuit is a series circuit of a voltage clamping capacitor Cc and an auxiliary switching element Q2. In this case, the auxiliary switching element Q2 is turned on complementarily to the switching element Q1. For the power factor improving section to pass a current corresponding to a voltage generated in the primary side series resonant capacitor C2 from the alternating-current power supply via the primary side rectifying element Di (fast rectifying element), the above primary side rectifying element Di is formed by a fast rectifying element. The voltage occurring in the primary side series resonant capacitor C2 is applied to the choke coil primary winding NC1, a voltage similar to the voltage occurring in the primary side series resonant capacitor C2 is generated in the choke coil secondary winding NC2. The current corresponding to the voltage occurring in the primary side series resonant capacitor C2 is made to flow from the alternating-current power supply. Incidentally, the frequency of a primary side parallel resonant circuit is substantially twice the primary side first series resonance frequency.
In the following, description will be made of characteristic parts of the fifth embodiment which parts are different from the first to fourth embodiments, and description of similar parts to those of the first to fourth embodiments will be omitted.
Description will first be made of the primary side first series resonant circuit and the primary side second series resonant circuit. The primary side first series resonant circuit forms a current path from the leakage inductance L3 occurring at the choke coil primary winding NC1 through the smoothing capacitor Ci and a ground point to the primary side series resonant capacitor C2. The primary side second series resonant circuit forms a current path in one direction from the primary side series resonant capacitor C2 to the primary winding N1 and then from the drain to the source of the switching element Q1 to the ground point. The primary side second series resonant circuit forms a current path in another direction from the primary side series resonant capacitor C2 through a body diode DD1 to the primary winding N1.
The choke transformer VFT has substantially the same structure as the converter transformer PIT shown in
The current flowing through the power factor improving section for power factor improvement is a part of resonance current flowing through the primary side first series resonant circuit and the primary side second series resonant circuit, which part is superimposed on a current I1. That is, by connecting the primary side series resonant capacitor C2 to the choke coil primary winding NC1 and connecting the choke coil secondary winding NC2 to the primary side rectifying element Di, the resonance current corresponding to the resonance voltage occurring in the primary side series resonant capacitor C2 is passed from the alternating-current power supply AC via the anodes and cathodes of two fast rectifying elements. A current resulting from adding together the resonance current for power factor improvement and a current obtained by rectifying commercial alternating-current power from the alternating-current power supply AC flows as the current I1.
That is, when the power factor improving section is not provided, the current I1 has a waveform in the shape of a pulse flowing around the peak of a voltage V2. However, the resonance current corresponding to the resonance voltage occurring in the primary side series resonant capacitor C2 is also passed outside the vicinity of the peak of the voltage V2, whereby a conduction angle is increased. On the other hand, the primary side direct-current power from the smoothing capacitor Ci is supplied to the primary winding N1 of the converter transformer PIT via the choke coil primary winding NC1. Thus, in the present embodiment, the resonance voltage occurring in the primary side series resonant capacitor C2 is fed back to the smoothing capacitor Ci via the choke coil secondary winding NC2 of the choke transformer VFT, and thereby a voltage feedback type power factor improving section is formed.
The concrete constants of parts in the fifth embodiment are set as follows. The secondary side direct-current output voltage Eo is 175 V. The period TOFF of the switching element Q1 is changed according to variation in load power Po, the period TON of the switching element Q1 is reduced with decrease in load power Po and increase in alternating input voltage VAC, and the value of the secondary side direct-current output voltage Eo is made constant by increasing switching frequency.
A ferrite material of the converter transformer PIT is EER-35. A gap of the converter transformer PIT is 1.2 mm. The coupling coefficient of the converter transformer PIT is 0.82. The primary winding N1 is set at 45 T. The secondary winding N2 is set at 30 T. The value of a primary side parallel resonant capacitor Cr is 6800 pF. The value of the primary side series resonant capacitor C2 is 0.1 μF. The value of the voltage clamping capacitor Cc is 0.1 μF. The value of a secondary side series resonant capacitor C4 is 0.056 μF. The value of a filter capacitor CN is 1 μF. The value of the leakage inductance L3 occurring at the choke coil primary winding NC1 is 82 μH. The value of the leakage inductance L3′ occurring at the choke coil secondary winding NC2 is 82 μH. The specifications of the primary side rectifying element Di are 3 A/600 V specifications. The specifications of the secondary side rectifying element Do are 5 A/200 V specifications. Each of the primary side rectifying element Di and the secondary side rectifying element Do is a fast rectifying element.
The waveforms of principal parts in the fifth embodiment are substantially the same as in
Introducing a part of representative characteristics that can be read from
When the value of the alternating input voltage VAC is 230 V, the value of the power factor PF at a load power Po of 300 W is 0.935, and the value of the power factor PF at a load power Po of 75 W is 0.76, these values representing a high power factor. When the value of the alternating input voltage VAC is 230 V, the value of the power conversion efficiency ηAC→DC at a load power Po of 300 W is high at 92.1%. In the range where the load power Po is 300 W to 0 W, the value of the direct-current input voltage Ei is 356 V to 376 V. The above values of the power factor in both cases where the value of the alternating input voltage VAC is 100 V and 230 V exceed a power factor of 0.75 as a standard value for Class D defined in IEC 61000-3-2.
Such a switching power supply circuit according to the present embodiment has improved the power conversion efficiency ηAC→DC as compared with the switching power supply circuit shown as background art in
In this case, the leakage inductance L3 occurring at the choke coil primary winding NC1 is a low value of 82 μH. Further, it suffices to have one switching element Q1 as switching element. Therefore device size and weight can be reduced. Further, it is possible to optimize the power factor improving section (approximate the power factor to one) while making the value of the leakage inductance L3 an optimum value (the value of the leakage inductance L3 that makes the primary side first series resonance frequency the primary side second series resonance frequency) by adjusting the ratio between the choke coil primary winding NC1 and the choke coil secondary winding NC2 of the choke transformer VFT.
In addition, in the switching power supply circuit according to the embodiment, the operation of the multiple resonant converter section and the power factor improving section is a so-called soft switching operation, and therefore the level of switching noise is greatly reduced as compared with the circuit using the active filter shown in
Further, the switching circuit according to the embodiment has the primary side series resonant circuits and the primary side parallel resonant circuit as well as a secondary side series resonant circuit. Therefore, the secondary side direct-current output voltage Eo can be maintained at a predetermined voltage level by very slight changes in frequency, and the noise filter can be designed easily. For this reason, the noise filter in one stage composed of one common mode choke coil CMC and two across capacitors CL suffices to meet power supply interference standards. In addition, a sufficient measure against the normal mode noise of the rectification output line can be taken by the single filter capacitor CN.
Further, rectifier diodes Do1 to Do4 on the secondary side and the like operate in synchronism with the switching element Q1. Therefore ground potential interference between the active filter side and the switching converter in the subsequent stage as in the power supply circuit of
Further, by employing the active clamping circuit, a range of variation of the ratio TON/TOFF between the on period TON and the off period TOFF is reduced as compared with a case where the active clamping circuit is not provided. Further, it is possible to lower the withstand voltage of the switching element Q1, expand a ZVS range, and achieve so-called range widening that allows the value of the alternating input voltage VAC to range widely from 85 V to 264 V.
In a switching power supply circuit according to a sixth embodiment shown in
The converter section in the sixth embodiment is the same as in the first to fifth embodiments in that ZVS operation is performed with the resonance frequency of the primary side first series resonant circuit set at substantially twice the resonance frequency of the primary side second series resonant circuit. The power factor improving section in the sixth embodiment is the same as in the fourth embodiment in that the current corresponding to the voltage occurring in the primary side series resonant capacitor C2 is passed from the alternating-current power supply AC via the fast rectifying element D1.
FIGS. 13 to 17 show examples of modification of the secondary side circuits that are replaceable in the switching power supply circuits according to the first to sixth embodiments.
A secondary side rectifier circuit shown in
The terminal part on the secondary winding part N2A side of the secondary winding N2 is connected to a point of connection between the anode of a rectifier diode Do1 and the cathode of a rectifier diode Do2 via the series connection of the secondary side series resonant capacitor C4A. In addition, the terminal part on the secondary winding part N2B side of the secondary winding N2 is connected to a point of connection between the anode of a rectifier diode Do3 and the cathode of a rectifier diode Do4 via the series connection of the secondary side series resonant capacitor C4B. The respective cathodes of the rectifier diode Do1 and the rectifier diode Do3 are connected to the positive electrode terminal of a smoothing capacitor Co. The negative electrode terminal of the smoothing capacitor Co is connected to the secondary side ground. A point of connection between the respective anodes of the rectifier diode Do2 and the rectifier diode Do4 is connected to the secondary side ground.
Thus, the secondary winding part N2A, the secondary side series resonant capacitor C4A, the rectifier diode Do1, the rectifier diode Do2, and the smoothing capacitor Co form a first voltage doubler half-wave rectifier circuit including the first secondary side series resonant circuit. The secondary winding part N2B, the secondary side series resonant capacitor C4B, the rectifier diode Do1, the rectifier diode Do2, and the smoothing capacitor Co form a second voltage doubler half-wave rectifier circuit including the second secondary side series resonant circuit. Thus the smoothing capacitor Co is charged with a rectified current by a potential obtained by superimposing a voltage induced in the secondary winding part N2B on a voltage across the secondary side series resonant capacitor C4B in a half period in which the alternating voltage of the secondary winding N2 is of one polarity. The smoothing capacitor Co is charged with a rectified current by a potential obtained by superimposing a voltage induced in the secondary winding part N2A on a voltage across the secondary side series resonant capacitor C4A in a half period in which the alternating voltage of the secondary winding N2 is of another polarity. Thereby a secondary side direct-current output voltage Eo having a level corresponding to twice the level of the voltage induced in the secondary winding part N2A and the secondary winding part N2B is obtained as voltage across the smoothing capacitor Co. That is, a voltage doubler full-wave rectifier circuit is obtained.
A secondary side rectifier circuit shown in
A secondary side rectifier circuit shown in
A secondary side rectifier circuit shown in
A secondary side rectifier circuit shown in
Incidentally, these partial resonant circuits resonate in timings of turning on and off the rectifier diode Do1 and the like.
It is to be noted that while the concrete examples of design of the power supply circuits according to the embodiments described thus far assume that commercial alternating-current power is input at an alternating input voltage VAC of 100 V, the embodiment of the present invention is not particularly limited to the value of the alternating input voltage VAC. For example, also in a case of designing a power supply circuit ready for input of 200-V commercial alternating-current power, similar effects can be obtained by using a configuration based on the present invention. In addition, as for circuit forms of details of the primary side voltage resonant converter and the configuration of the secondary side rectifier circuit including the secondary side series resonant circuit, for example, other circuit forms and other configurations are conceivable. Further, as the switching element, an element other than a MOS-FET, such for example as an IGBT (Insulated Gate Bipolar Transistor) and a bipolar transistor may be selected. In addition, while each of the above-described embodiments employs the externally excited switching converter, the embodiment of the present invention is applicable to cases where the switching converter is formed as a self-excited switching converter.
It should be understood by those skilled in the art that various modifications, combinations, sub-combinations and alterations may occur depending on design requirements and other factors insofar as they are within the scope of the appended claims or the equivalents thereof.
Number | Date | Country | Kind |
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P2006-067730 | Mar 2006 | JP | national |