Patent Application
20180309372
The present invention relates generally to an electronic circuit, and, in particular embodiments, to a system and method for a switched mode converter.
Power supply systems are pervasive in many electronic applications from computers to automobiles. Generally, voltages within a power supply system are generated by performing a DC-DC, DC-AC, and/or AC-DC conversion by operating a switch loaded with an inductor or transformer. One class of such systems includes switch-mode power supply (SMPS). An SMPS is usually more efficient than other types of power conversion systems because power conversion is performed by controlled charging and discharging of the inductor or transformer and reduces energy lost due to power dissipation caused by resistive voltage drops.
Specific topologies of SMPS include buck converters, boost converters, and buck-boost converters, among others. Depending on the topology selected and the needs of a particular system, the SMPS may be implemented using a half-bridge architecture, a full bridge architecture, or with any other implementation known in the art.
A transformer may be used in some converters, in part, to provide galvanic isolation between input and output of the converter. For example, galvanically isolating an alternating current (AC) power source from the output of the converter may help protect against electrical shocks.
Converters may be implemented with resonant topologies. Resonant topologies typically exhibit high efficiency and high power density. Resonant topologies may be implemented by resonating a combination of inductors and capacitors. For example, an LLC converter is a resonant converter that includes two inductors and one capacitor.
A particular type of power supply that is widely used is the AC adapter. AC adapters are external AC/DC power supplies typically used to provide DC power from a standard AC power source. AC adapters may receive their power from an AC power source. The two most common types of AC power sources (also referred to as mains power) are the 120 Vrms, 60 Hz power source, also known as low-line power source or low-line power, and the 230 Vrms, 50 Hz power source, also known as high-line power source or high-line power. The root-mean-square (RMS) voltage may not be exactly 120 Vrms and 230 Vrms for low-line and high-line, respectively. For example, the mains voltage of a low-line input may vary between 85 Vrms and 140 Vrms. Similarly, the mains voltage of a high-line input may vary between 200 Vrms and 270 Vrms. The AC signal produced by a low-line power source may be referred to as a low-line AC signal, low-line signal or low-line voltage. Similarly, the AC signal produced by a high-line power source may be referred to as a high-line AC signal, high-line signal or high-line voltage.
Universal adapters are AC adapters that are configured to operate with either low-line power or high-line power. Some universal adapters automatically adjust to the type input power received. Other universal adapters may allow for manual selection of the mode of operation.
Converters may also be used in systems that comply with a particular standard. For example, the USB Power Delivery (USB-PD) specification describes the standard related to power delivery in USB applications.
In accordance with an embodiment, a converter includes: a rectifying stage having a first supply terminal and a second supply terminal, the first supply terminal and the second supply terminal configured to receive a bipolar ac signal from an AC power source, the rectifying stage including a half-bridge circuit coupled between the first supply terminal and the second supply terminal, a transformer, and a resonant tank coupled between an output of the half-bridge circuit and a primary winding of the transformer; and a DC-DC converter stage coupled between the rectifying stage and an output terminal.
For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:
Corresponding numerals and symbols in different figures generally refer to corresponding parts unless otherwise indicated. The figures are drawn to clearly illustrate the relevant aspects of the preferred embodiments and are not necessarily drawn to scale. To more clearly illustrate certain embodiments, a letter indicating variations of the same structure, material, or process step may follow a figure number.
The making and using of the presently preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention.
The present invention will be described with respect to preferred embodiments in a specific context, a converter having a resonant converter stage cascaded with a DC-DC converter stage in various configurations, voltage and power levels. Embodiments of the present invention may be used with other configurations, and other voltage and power levels.
In an embodiment of the present invention, a converter provides a regulated DC output to a load by using a resonant converter stage that receives energy from an AC power source, and a DC-DC converter stage that regulates the output voltage. The resonant converter may also provide galvanic isolation between the AC power source and the load. The DC-DC converter may be implemented to comply with industry standards, such as USB-PD, and may support a wide range of voltage and power levels. Some embodiments may be implemented with power factor correction (PFC). Other embodiments may be implemented without PFC. The resonant converter stage with zero voltage switching (ZVS) or quasi-ZVS (QZVS) techniques. The DC-DC converter stage may also be implemented with ZVS or QZVS.
In some embodiments, the resonant converter stage is implemented with a traditional LLC topology that uses a bridge rectifier coupled between the AC power source and the LLC converter. The LLC converter may operate with a constant frequency and duty cycle Other embodiments may implement the resonant converter stage with an ACX topology configured to receive an AC signal from the AC power source and produce a rectified signal. Embodiments implementing the resonant converter stage with an ACX topology may operate without a bridge rectifier. The ACX converter may be implemented with bidirectional switches that may switch with constant frequency and duty cycle.
Some applications may benefit from AC/DC conversion. For example, the USB-PD specification version 1.1, revision 3.0, makes it possible for a monitor with a supply from the wall to simultaneously charge a laptop through a USB cable while operating as a display. Some embodiments of the present invention are configured to receive an AC signal from an AC power source and provide power to a load while complying with the USB-PD standard. A resonant stage implemented with an LLC converter may be used to transfer energy from the AC power source to a DC-DC converter while providing galvanic isolation. A diode-bridge may be used to provide a rectified signal to the LLC converter.
During normal operation, diode bridge 106 may rectify an AC signals received from AC power source 102 and provide a rectified voltage to node Vin_LLC. Capacitor Cin may provide energy storage, in part, to reduce the voltage ripple of node Vin_LLC. LLC converter no may receive the rectified voltage and deliver power to energy storage stage 112. LLC converter 110 may also provide galvanic isolation from AC power source 102 by using a transformer. DC-DC converter 122 may be used to deliver and regulate power to load Rload. EMI filter 104 may be used to reduce or eliminate EMI generated by converter 100.
Diode bridge 106 is configured to rectify an AC signal from AC power source 102 and produce a DC voltage at node Vin_LLC. Diode bridge 106 may be implemented according to various ways known in the art. For example, some embodiments may implement diode bridge 106 with four diodes. Other embodiments may use synchronous rectification techniques.
LLC converter no may receive a rectified signal from diode bridge 106 and produce a DC voltage at node Vout_LLC. LLC converter no may be implemented as a conventional LLC converter. For example, the switching frequency of LLC converter no may be modulated to produce a regulated voltage at node Vout_LLC. Alternatively, LLC converter no may be implemented with fixed frequency techniques. For example, since DC-DC converter 122 is coupled between LLC converter no and output node Vout, LLC converter no may switch at a constant frequency and constant duty cycle, and the voltage of node Vout may be regulated by DC-DC converter 122. The switching frequency of LLC converter 122 may be, for example, higher than 20 kHz. Implementations with frequencies of 100 kHz or higher are also possible. In some embodiments, LLC converter 110 may implement ZVS or QZVS.
DC-DC converter 122 may be implemented according to various ways known in the art. For example, DC-DC converter 122 may be implemented as a buck converter, boost converter, buck-boost converter with inverting and non-inverting topologies.
EMI filter 104 may be implemented according to various ways known in the art. EMI filter 104 may be configured to filter out frequencies in the range of frequencies that LLC converter no switches. Since LLC converter no may switch at frequencies higher than mains frequency, EMI filter 104 may be implemented with smaller inductors. In some embodiments, EMI filter 104 may be implemented as a notch filter to filter out a single frequency. For example, such may be the case for embodiments implementing LLC converter no with fixed frequency operation.
During normal operation, LLC converter 110 receives a DC signal at node Vin_LLC and produces a step down voltage at node Vout_LLC. Energy storage stage 112 stores energy and may also reduce the voltage ripple of node Vout_LLC. DC-DC converter 122 receives the step down voltage of node Vout_LLC and produces a regulated voltage at node Vout.
LLC converter no may operate as a conventional LLC converter. For example, half-bridge 129 may switch according to switching techniques of a conventional LLC converter to transfer energy to the secondary side of transformer 116. For example, the switching frequency of LLC converter 110 may be modulated to control the voltage of node Vout_LLC. LLC converter no may switch at frequencies higher than 20 kHz. LLC converter no may switch at frequencies around 100 kHz. Other frequencies may be used.
LLC secondary circuit 107 may be implemented according to various ways known in the art. For example, as shown in
DC-DC converter 122 may produce a regulated voltage at node Vout. Since DC-DC converter 122 is implemented as a buck-boost converter, DC-DC converter 122 may operate as a buck converter when the voltage of node Vout_LLC is higher than the desired voltage at Vout, and may operate as a boost converter when the voltage of node Vout_LLC is lower than the desired voltage at Vout. When DC-DC 122 operates as a buck converter, transistor 176 is off and transistor 174 is on, and transistors 170 and 172 switch on and off according to a typical buck converter. When DC-DC 122 operates as a boost converter, transistor 170 is on, transistor 172 is off, and transistors 174 and 176 switch on and off according to a typical boost converter.
Since DC-DC converter 122 is implemented as a non-inverted buck-boost converter, DC-DC converter 122 may produce a regulated output irrespective of whether AC power source 102 produces a high-line signal or a low-line signal. For example, when AC power source 102 produces a high-line voltage, DC-DC converter 122 may operate as a buck converter for the majority of the time. When AC power source 102 produces a low-line voltage, DC-DC converter 122 may operate as a boost converter for the majority of the time.
DC-DC converter 122 may regulate the voltage of node Vout to, for example, 20 V, 18 V, 12 V, 10 V, 5 V, 3.3 V, 1.8 V, 1.2 V, or 1 V. Other values may be used. DC-DC converter 122 may be implemented according to various ways known in the art and may be configured to regulate the voltage while complying with a particular standard such as, for example, USB-PD. For example, as shown in
Controller 145 is configured to produce signals S130, S134, S138, S140, S170, S172, S174, and S176, to drive transistors 130, 134, 138, 140, 170, 172, 174, and 176, respectively. Coupling controller 145 to transistors 130, 134, 138, 140, 170, 172, 174, and 176 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 145 from other parts of the circuit. Coupling between controller 145 and other components of converter 100 may also be achieved in other ways known in the art.
Controller 145 may be implemented as a single chip. For example, controller 145 may be implemented in a monolithic substrate. Alternatively, controller 145 may be implemented as a collection of controllers, such as, for example, a controller for controlling LLC converter 110, and a controller for controlling DC-DC converter 122. Other implementations known in the art are also possible.
Transformer 116 may include primary winding 118, upper secondary winding 121, and lower secondary winding 122. Other transformer implementations are possible. For example, transformer 116 may be implemented with a single secondary winding. The selection of the transformer may depend on the particular application. Converter 100 may be modified to accommodate a particular transformer implementation. For example, LLC converter no and controller 145, may be modified to accommodate a particular transformer selection. In some embodiments, resonant inductors 126 and 124 may be incorporated into transformer 116. Alternatively, resonant capacitor 128 and resonant inductors 126 and 124 may be implemented with discrete components. Other implementations are also possible.
As shown in
Advantages of some embodiments of the present invention include that a LLC converter may be implemented with two transistors on the primary side of the transformer. Since transformer size is typically inversely related to the switching frequency, using an LLC topology with a switching frequency substantially higher than the switching frequency of mains power may result in a physically small transformer.
In an embodiment of the present invention, an ACX converter receives an AC signal from an AC power source and produces a rectified signal while providing galvanic isolation between the AC power source and a load. The ACX converter is implemented with a half-bridge including two bidirectional switches that switch at a constant frequency and duty cycle. A DC-DC converter coupled to the ACX converter regulates the output voltage delivered to the load.
During normal operation, ACX converter 208 receives an AC signal from AC power source 102 and delivers a rectified signal to energy storage stage 212 and DC-DC converter 222. ACX converter 208 also provides galvanic isolation from AC power source 102 by the use of a transformer. DC-DC converter 222 regulates and delivers power to load Rload. EMI filter 204 may be used to reduce or eliminate EMI generated by converter 200.
As shown in
ACX converter 208 may be implemented with bidirectional switches switching at a constant frequency and duty cycle. The switching frequency may depend on the particular application and may be, for example, 100 kHz. In some embodiments, ACX converter 208 may implement ZVS or QZVS. The switching duty cycle of the bidirectional switches of ACX converter 208 may be, for example, 50%. A smaller duty cycle may be used depending on the application. For example, a duty cycle smaller than 50% may be used to accommodate for ZVS or QZVS.
DC-DC converter 222 may be implemented according to various ways known in the art. For example, DC-DC converter 222 may be implemented as a buck converter, boost converter, buck-boost converter, and with inverting and non-inverting topologies. In some embodiments, DC-DC converter 222 may be combined with ACX secondary circuit 203.
EMI filter 204 may be implemented according to various ways known in the art. Since ACX converter 208 may switch at a constant frequency, EMI 204 may be implemented, for example, as a notch filter configured to remove the switching frequency of ACX converter 208.
During normal operation, ACX converter 208 receives an AC signal at node Vin_ACX and delivers a rectified output at node Vout_ACX. In particular, half-bridge 229 receives an AC signal from node Vin_ACX and bidirectional switches 230 and 234 switch at a constant frequency and duty cycle to transfer energy to the secondary sides of transformer 216. Transistors 238, 240, 242, and 244 operate as a rectifying bridge that produces a rectified output at node Vout_ACX.
Transistors 238, 240, 242, and 244 may switch to produce a rectified voltage of node Vout_ACX according to synchronous rectification techniques. For example, transistors 238, 240, 242 and 244 may switch with ZVS or QZVS according to synchronous rectification techniques known in the art. As can be seen in
Controller 245 is configured to produce signals S230, S234, S238, S240, S242, and S244, to drive bidirectional switches 230 and 234, and transistors 238, 240, 242, and 244, respectively. As described below with respect to
Coupling controller 245 to bidirectional switches 230 and 234, and transistors 238, 240, 242, and 244 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 245 from other parts of the circuit. Coupling between controller 245 and other components of converter 200 may also be achieved in other ways known in the art.
Controller 245 may be implemented as a single chip. For example, controller 245 may be implemented in a monolithic substrate. Alternatively, controller 245 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX primary circuit 201, and a controller for controlling ACX secondary circuit 203. Other implementations known in the art are also possible.
ACX secondary circuit 203 may be implemented as a full-bridge synchronous rectifier. Alternatively, other implementations, such as a center-tap configuration or a voltage doubler may be used. For example,
Transformer 216 may include primary winding 218, and secondary winding 220. Other transformer implementations are possible. For example, transformer 216 may be implemented with a center-tapped configuration. The selection of the transformer may depend on the particular application. Converter 200 may be modified to accommodate a particular transformer implementation. For example, ACX converter 208 and controller 245, may be modified to accommodate a particular transformer selection.
In some embodiments, resonant inductors 226 and 224 may be incorporated into transformer 216. Alternatively, resonant capacitor 128 and resonant inductors 226 and 224 may be implemented with discrete components. Other implementations are also possible.
Bidirectional switches 230 and 234 may switch at a fixed frequency above the frequency of the AC voltage of node Vin_ACX. The particular switching frequency of bidirectional switches 230 and 234 may depend on the particular application. For example, bidirectional switches 230 and 234 may switch at 100 kHz. Other frequencies may be used.
Bidirectional switches 230 and 234 may be implemented according to various ways known in the art. For example,
Alternatively, bidirectional switches 230 and 234 may be implemented with other transistor technologies and in other configurations. For example,
After a resonant period, current 246 may change polarity and ACX primary circuit 201 transitions to a second state with bidirectional switch 230 open and bidirectional switch 234 closed, as show in
When the voltage of node Vin_ACX is negative, ACX primary circuit 201 may be in the first state with bidirectional switch 230 closed and bidirectional switch 234 open, as shown in
After a resonant period, current 246 may change polarity and ACX primary circuit 201 transitions to a second state with bidirectional switch 230 open and bidirectional switch 234 closed, as show in
As shown in
As shown in
A similar behavior is observed when the voltage of node Vin_ACX is negative. As shown in
The ACX converter receives an AC signal from an AC power source, such as AC power source 202 during step 273. The AC signal may be, for example, a high-line AC signal, also refereed as a high-line input voltage or high-line input, or a low-line AC signal, also referred as a low-line input voltage, or low-line input. During step 275, a half-bridge receiving the AC signal, such as half-bridge 229, switches with a constant frequency and a constant duty cycle. In particular, an upper bidirectional switch and a lower bidirectional switch of the half-bridge may switch with opposite phases at the constant frequency and constant duty cycle. The constant duty cycle may be 50% or lower. The duty cycle may be adjusted such that ZVS or QZVS is achieved. The constant frequency may be adjusted to be at or near a resonant frequency of a resonant tank coupled to the half-bridge. The resonant tank includes a resonant capacitor, such as resonant capacitor 228, and a first and second resonant inductors, such as resonant inductors 226 and 224 respectively. The resonant tank may be coupled to a primary winding of a transformer, such as primary winding 218 of transformer 216.
During step 277, the resonant tank is activated. In other words, the resonant tank is activated such that it resonates. Specifically, when the first bidirectional switch is closed and the second bidirectional switch is open, the resonant tank is exposed to the voltage of a first supply node, such as node Vin_ACX, thereby inducing the flow of current on a first direction, and when the first bidirectional switch is open and the second bidirectional switch is closed, the resonant tank is exposed to the voltage of a second supply node, such as primary ground 209, thereby inducting current flowing in a second direction opposite the first direction.
When the voltage of the first supply node is higher than the voltage of the second supply node, the first bidirectional switch may be closed and the second bidirectional switch may be open, and current flows from the first supply node, through the resonant tank, and through the primary winding of the transformer, such as shown in
When the voltage of the first supply node is lower than the voltage of the second supply node, the first bidirectional switch may be closed and the second bidirectional switch may be open and current flows from the resonant tank towards the first supply node, such as shown in
As the current flowing through the primary winding of the transformer changes polarity, a current induced in a secondary winding of the transformer, such as secondary winding 220, also changes polarity, thereby producing an alternating voltage across the secondary winding. The alternating voltage of the secondary winding may be rectified with a rectifying circuit, such as, for example, ACX secondary circuit 203. The rectifying circuit may switch according to synchronous rectification techniques to produce a rectified voltage of an output node of the ACX converter, such as node Vout_ACX. For example, ACX secondary circuit 203 may switch as shown in
Advantages of embodiments of the present invention include that since the ACX converter is configured to operate with an AC signal, the ACX converter may operate without a rectifying bridge between the AC power source and the input of ACX converter. As an additional benefit, small input capacitor Cin may be used since ACX converter may operate without controlling a ripple of the input voltage. The energy storage having capacitors with higher capacitance, therefore, may be implemented in energy storage stage 212. Since energy storage stage 212 is typically exposed to lower peak voltages than node Vin_ACX, lower-rated capacitors may be used. Since capacitors rated at low voltages are generally smaller than capacitors rated at high voltages, the physical volume of converters implementing the ACX converter may be reduced. Some embodiments of the present invention, therefore, may have a smaller physical volume than systems that use a rectifying bridge where the energy storage is on the primary side.
Other advantages of embodiments of the present invention include that since bidirectional switches 230 and 234 switch at a constant frequency and duty cycle irrespective of the polarity of the voltage of node Vin_ACX, the implementation of controller 245 may be simplified. The use of a fixed frequency may have the additional advantage of simplifying the implementation of EMI filter 204, which may be implemented to filter out the particular switching frequency of ACX converter 208. The relative high switching frequency of ACX converter 208 may also result in a smaller transformer implementation.
ACX converters may also be implemented with ZVS and QZVS. For example,
As shown in
During the time between t1 and t2, the voltage across bidirectional switch 330 increases while the voltage across bidirectional switch 334 decreases, as shown by curves 350 and 352, respectively. At time t2, therefore, transistor 336 may be turned on with ZVS, which corresponds to
During the time between t2 and t3, the voltage across bidirectional switch 334 is low, close to 0 V, while the voltage across bidirectional switch 330 is high, as shown by curves 352 and 350, respectively. During the time between t2 and t3, current flowing through bidirectional switch 334 increases, then peaks and then decreases according to a resonant period, as shown by curve 353, while there is no current flowing through bidirectional switch 330, as shown by curve 351. At time t3, transistor 336 is turned off, which corresponds to
During the time between t3 and t4, the voltage across bidirectional switch 330 decreases while the voltage across bidirectional switch 334 increases, as shown by curves 350 and 352, respectively. At time t4, therefore, transistor 332 may be turned on with ZVS, which corresponds to
As shown in
During the time between t1 and t2, the voltage across bidirectional switch 330 decreases while the voltage across bidirectional switch 334 increases, as shown by curves 350 and 352, respectively. At time t2, therefore, transistor 331 may be turned on with ZVS, which corresponds to
During the time between t2 and t3, the voltage across bidirectional switch 330 is low, close to 0 V, while the voltage across bidirectional switch 334 is high, as shown by curves 350 and 352, respectively. During the time between t2 and t3, current flowing through bidirectional switch 330 increases, then peaks and then decreases according to a resonant period, as shown by curve 351, while there is no current flowing through bidirectional switch 334, as shown by curve 353. At time t3, transistor 331 is turned off, which corresponds to
During the time between t3 and t4, the voltage across bidirectional switch 334 decreases while the voltage across bidirectional switch 330 increases, as shown by curves 352 and 350, respectively. At time t4, therefore, transistor 335 may be turned on with ZVS, which corresponds to
While ACX primary circuit 301 includes bidirectional switches 330 and 334 implemented with NMOS transistors in a back-to-back, common-drain configuration, it is understood that other transistor types and configurations are possible. For example, ACX primary circuit may be implemented with ZVS with any of the bidirectional switches shown in
The ACX primary circuit receives an AC signal from an AC power source, such as AC power source 202 during step 372. The AC signal may be, for example, a high-line AC signal or a low-line AC signal. The polarity of the AC signal is determined during step 374. If the AC signal is positive, non-blocking transistors, such as transistors 331 and 335, are turned on during step 376. During step 378, a first blocking transistor, such as transistor 332, is turned on. As a result, current may flow through the first blocking transistor and a resonant tank, such as a resonant tank including resonant capacitor 228 and resonant inductor 226. During step 380 and a first time after the first blocking transistor is turned on, the first blocking transistor is turned off. The first time may be a time substantially similar to the resonant period of the resonant tank. Turning off the first blocking transistor may discharge a drain capacitance of a second blocking transistor, such as transistors 336 as well as cause the current flowing through the resonant tank to change polarity. During step 382 and a second time after turning off the first blocking transistor, the second blocking transistor may be turned on. Since the drain capacitance of the second transistor is reduced, for example, to 0 V, the second blocking transistor may turn on with ZVS during step 382. During step 384 and a third time after turning on the second blocking transistor, the second blocking transistor may be turned off. The third time may be substantially similar to the first time. Turning off the second blocking transistor may discharge a drain capacitance of the first blocking transistor as well as cause the current flowing through the resonant tank to change polarity. The polarity of the AC signal is checked during step 374. If the polarity of the AC signal continues to be positive, step 376 may be skipped and the first blocking transistor may be turned on during step 378, repeating the sequence. Since the drain capacitance of the first blocking transistor is reduced, for example, to 0 V, the first blocking transistor may be turned on with ZVS during step 378. The sequence of steps including 378, 380, 382, and 384 correspond to loop 385. Since the switching frequency of the blocking transistors is higher than the mains frequency, it is understood that loop 385 may be executed several times consecutively.
The determination of which transistors are non-blocking transistors may depend on the polarity of the AC signal as well as on the configuration of the bidirectional switch. For example, for a positive AC signal, the non-blocking transistors of ACX primary circuit 301 are transistors 331 and 335 and the blocking transistors of ACX primary circuit 301 are transistors 332 and 336. For a negative AC signal, the non-blocking transistors of ACX primary circuit 301 are transistors are 332 and 336 and the blocking transistors of ACX primary circuit 301 are transistors 331 and 335. A person skilled in the art would be able to determine which transistors of the bidirectional switch are the blocking and non-blocking transistors depending on the polarity of the AC signal and the implementation of the bidirectional switch.
If the AC signal is negative, non-blocking transistors, such as transistors 332 and 336, are turned on during step 376. During step 386, a third blocking transistor, such as transistor 335, is turned on. As a result, current may flow through the third blocking transistor and the resonant tank. During step 388 and a fourth time after the third blocking transistor is turned on, the fourth blocking transistor is turned off. The fourth time may be substantially similar to the first time. Turning off the third blocking transistor may discharge a drain capacitance of a fourth blocking transistor, such as transistors 331 as well as cause the current flowing through the resonant tank to change polarity. During step 390 and a fifth time after turning off the third blocking transistor, the fourth blocking transistor may be turned on. Since the drain capacitance of the fourth transistor is reduced, for example, to 0 V, the fourth blocking transistor may turn on with ZVS during step 382. During step 392 and a sixth time after turning on the fourth blocking transistor, the fourth blocking transistor may be turned off. The sixth time may be substantially similar to the first time. Turning off the fourth blocking transistor may discharge a drain capacitance of the third blocking transistor as well as cause the current flowing through the resonant tank to change polarity. The polarity of the AC signal is checked during step 374. If the polarity of the AC signal continues to be negative, step 386 may be skipped and the third blocking transistor may be turned on during step 388, repeating the sequence. Since the drain capacitance of the third blocking transistor is reduced, for example, to 0 V, the third blocking transistor may be turned on with ZVS during step 378. The sequence of steps including 388, 390, 392, and 394 correspond to loop 387. Since the switching frequency of the blocking transistors is higher than the mains frequency, it is understood that loop 387 may be executed several times consecutively.
Advantages of some embodiments of the present invention include an increase efficiency resulting from the ACX converter switching with ZVS or QZVS in both the ACX primary circuit and the ACX secondary circuit. Since the non-blocking transistor may not switch during a first polarity of the AC signal, the switching of ACX primary circuit may be simplified, with, for example, only two transistor switching at the ACX switching frequency.
In addition to an ACX converter operating at fixed frequency and duty cycle, control of the ACX converter may be further simplified. For example,
ACX converter 408 may operate in a similar manner as ACX converter 208. ACX converter 408, however, uses diodes 438, 440, 442, and 444 instead of transistors 238, 240, 242, and 244 for rectification purposes.
In the embodiment of
|Vin_ACX|>2·n·Vout_ACX (1)
where n is the turn ratio of transformer 216. In some embodiments, n is equal to 2. Other values of n may be used. Equation 1 is also referred to as the forward energy transfer rule. When Equation 1 is true, the forward energy transfer condition is satisfied and energy is transferred from the primary side of the transformer to the secondary side of the transformer. When Equation 1 is false, the forward energy transfer condition is not satisfied. When the forward energy transfer condition is not satisfied, diodes 438, 440, 442, and 444 may prevent transfer of energy from the secondary side of the transformer back to the primary side of the transformer.
Controller 445 is configured to produce signals S230, S234, to drive bidirectional switches 230 and 234, respectively. As described above with respect to ACX converters 208 and 308, signals S230 and S240 may include additional signals for driving internal transistors of the bidirectional switches and may be configured to switch bidirectional switches 230 and 234 with ZVS. Controller 445, therefore, may produce signals S230 and S234 in open loop. In other words, controller 445 may control ACX converter 408 without sensing signals of ACX converter 408.
When the ACX secondary circuit is implemented with transistors, the transistors of the ACX secondary circuit may be turned on to reduce conduction losses when the forward energy transfer condition is satisfied. The transistors of the ACX secondary circuit may be turned off to prevent energy from transferring from the secondary side of the transformer to the primary side of the transformer when the forward energy transfer condition is not satisfied. For example,
ACX converter 508 may operate in a similar manner as ACX converter 408. ACX converter 508, however, uses transistors 238, 240, 242, and 244 instead of diodes 438, 440, 442, and 444 for rectification purposes.
To prevent energy transfer from the secondary side of transformer 216 to the primary side of transformer 216, controller 545 may turn off transistors 238, 240, 242, and 244 when the forward energy transfer condition is not satisfied. In other words, controller 545 may start switching transistors 238, 240, 242, and 244 according to synchronous rectification techniques when the forward energy transfer condition is satisfied and turn off transistors 238, 240, 242, and 244 when the forward energy transfer condition is not satisfied.
To determine when to start switching transistors 238, 240, 242, and 244, controller 545 may sense when the current flowing through the body diodes of transistors 238, 240, 242, or 244 becomes positive. One way to detect when the current flowing through the body diodes of transistors 238, 240, 242, or 244 becomes positive is to monitor a current flowing towards node Vout_ACX with current sensor 543. Controller 545, therefore, may begin switching transistors 238, 240, 242, and 244 according to synchronous rectification techniques when the current flowing through current sensor 543 becomes positive.
To determine when to stop switching transistors 238, 240, 242, and 244, controller 545 may sense the voltage at node Vout_ACX and turn off transistors 238, 240, 242, and 244 when the voltage of node Vout_ACX reaches a peak value. The determination of the peak value may be performed with a peak detector (not shown).
Current sensor 543 may be implemented according to various ways known in the art. For example, an analog-to-digital converter (ADC) may be used to sense a voltage across a sense resistor to determine the current. Other circuits and methods may be used to implement current sensor 543.
Controller 545 may be coupled to current sensor 543 and node Vout_ACX according to various ways known in the art. For example, opto-couplers may be used for coupling purposes to electrically isolate the controller from other parts of the circuit. Alternatively, controller 545 may be electrically isolated in other ways known in art. Other embodiments may couple controller 545 to current sensor 543 and node Vout_ACX with a direct electrical connection.
The waveforms of
As shown in
As shown in
By avoiding switching bidirectional switches during times where there is no energy transfer from the primary side of transformer 216 to the secondary since of transformer 216, efficiency of the ACX converter may be increased. Specifically, some of the switching losses associated with the switching of the bidirectional switches may be avoided without substantially impacting the energy delivery.
Detecting the start time and stop time for driving bidirectional switches 230 and 234 may be performed according to various ways known in the art. For example, the start time may be detected by monitoring the voltage of node Vin_ACX and detecting the zero crossing. The stop time may be determined by monitoring the voltage of node Vin_ACX and detecting the peak voltage of node Vin_ACX. Alternatively, an ACX converter may determine the frequency of the AC signal of node Vin_ACX and use a timer that counts from the time the zero crossing is detected to determine when to stop switching bidirectional switches 230 and 234. Other methods known in the art may be used.
As shown in
Detecting the start time and stop time for driving bidirectional switches 230 and 234 may be performed according to various ways known in the art. For example, the start time may be detected by monitoring the voltage of node Vin_ACX and comparing it with the voltage of node Vout_ACX to determine when the forward energy transfer condition is satisfied. Other methods known in the art may be used.
As shown in
The ACX secondary circuit may be implemented in various topologies. For example,
ACX converter 908 may operate in a similar manner as ACX converter 208 and may implement method 271 of operating an ACX converter. ACX converter 908 may also implement ZVS and method 370 of operating an ACX primary circuit with ZVS. ACX converter 908, however, implements ACX secondary circuit 908 with a center-tap topology instead of the full-bridge topology of ACX secondary circuit 208. Controller 945 may be adapted accordingly.
As shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 is open and bidirectional switch 234 is closed, as show in
When the voltage of node Vin_ACX is negative, bidirectional switch 230 is closed and bidirectional switch 234 is open, as shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 is open and bidirectional switch 234 is closed, as show in
ACX secondary circuit 903 may implement ZVS and may switch according to known synchronous rectification techniques. Some embodiments may implement ACX secondary circuit 903 with diodes instead of transistors 938 and 940. Other implementations and modifications are also possible.
ACX converter 1008 may operate in a similar manner as ACX converter 208 and may implement method 271 of operating an ACX converter. ACX converter 1008 may also implement ZVS and method 370 of operating an ACX primary circuit with ZVS. ACX converter 1008, however, implements ACX secondary circuit 1008 with a half-bridge voltage doubler topology instead of the full-bridge topology of ACX secondary circuit 208. Controller 1045 may be adapted accordingly.
As shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 may be open and bidirectional switch 234 may be closed, as show in
When the voltage of node Vin_ACX is negative, bidirectional switch 230 may be closed and bidirectional switch 234 may be open, as shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 may be open and bidirectional switch 234 may be closed, as show in
ACX secondary circuit 1003 may implement ZVS and may switch according to known synchronous rectification techniques. Some embodiments may implement ACX secondary circuit 1003 with diodes instead of transistors 1038 and 1040. Other implementations and modifications are also possible.
Referring back to
During normal operation, ACX converter 1108 receives an AC signal at node Vin_ACX and produces a rectified voltage at node Vout_ACX. Energy storage stage 1112 stores energy and may also reduce the voltage ripple of node Vout_ACX. DC-DC converter 1122 receives the rectified voltage of node Vout_ACX and produces a regulated voltage at node Vout. Since DC-DC converter 1122 is operating as a buck converter, the voltage of node Vout may be lower than the voltage of node Vout_ACX.
More particularly, ACX converter 1100 may be configured such that ACX converter 1100 operates with ACX secondary circuit 1103 switching in a full-bridge configuration when the AC signal of node Vin_ACX is a high-line signal and in a voltage doubler configuration when the AC signal of node Vin_ACX is a low-line signal. For example, when the AC signal of node Vin_ACX is a high-line signal, bidirectional switch 1149 may be closed and bidirectional switch 1151 may be open. When bidirectional switch 1149 is closed and bidirectional switch 1151 is open, transistor 1138, 1140, 1142 and 144 may switch in a similar as ACX converter 208.
When the AC signal of node Vin_ACX is a low-line signal, bidirectional switch 1149 may be open and bidirectional switch 1151 may be closed and transistors 1140 and 1144 may be off. When bidirectional switch 1149 is open, bidirectional switch 1151 is closed, and transistors 1140 and 144 are off, transistor 1138 and 1142 may switch in a similar manner as ACX converter 1008.
When the AC input is a high-line signal and bidirectional switch 1149 is closed and bidirectional switch 1151 is open, ACX converter 1108 charges capacitor 1114 in series with capacitor 1115. When the AC input is a low-line signal and bidirectional switch 1149 is open and bidirectional switch 1151 is closed, ACX converter 1108 charges capacitor 1114 and capacitor 1115 alternatively. When the AC input is a low-line signal, therefore, the voltage at node Vout_ACX is the sum of the voltages across capacitors 1114 and 1115. DC-DC converter 1122, therefore, may receive similar voltage levels irrespective of whether the AC signal of node Vin_ACX is a high-line signal or a low-line signal.
DC-DC converter 1122 may regulate the voltage of node Vout to, for example, 20 V, 18 V, 12 V, 10 V, 5 V, 3.3 V, 1.8 V, 1.2 V, or 1 V. Other values may be used. DC-DC converter 1122 may be implemented according to various ways known in the art and may be configured to regulate the voltage while complying with a particular standard such as, for example, USB-PD.
Bidirectional switches 1149 and 1151 may be implemented according to various ways known in the art. For example, bidirectional switches 1149 and 1151 may be implemented with the topologies shown in
Controller 1145 is configured to produce signals S230, S234, S1138, S1140, S1142, S1144, S1153, S1155, S1149, and S1151 to drive bidirectional switches 230 and 234, transistors 1138, 1140, 1142, 1144, 1153, and 1155, and bidirectional switches 1149 and 1151, respectively. Coupling controller 1145 to bidirectional switches 230 and 234, transistors 1138, 1140, 1142, 1144, 1153, and 1155, and bidirectional switches 1149 and 1151 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 1145 from other parts of the circuit. Coupling between controller 1145 and other components of converter 1100 may also be achieved in other ways known in the art.
Controller 1145 may be implemented as a single chip. For example, controller 1145 may be implemented in a monolithic substrate. Alternatively, controller 1145 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX converter 1108, and a controller for controlling DC-DC converter 1122. Other implementations known in the art are also possible.
As shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 may be open and bidirectional switch 234 may be closed, as show in
When the voltage of node Vin_ACX is a positive low-line voltage, bidirectional switch 230 may be closed and bidirectional switch may be open, bidirectional switch 1149 may be open and bidirectional switch 1151 may be closed, and transistors 1140 and 1144 may be off, as shown in
After a resonant period, current 246 may change polarity and bidirectional switch 230 may be open and bidirectional switch 234 may be closed, as show in
ACX secondary circuit 1103 may implement ZVS and may switch according to known synchronous rectification techniques. Some embodiments may implement ACX secondary circuit 1103 with diodes instead of transistors 1138, 1140, 1142 and 1144. Other implementations and modifications are also possible.
As shown in
Since transistors 1153 and 1155 of DC-DC converter 1122 operate continuously to deliver energy to load Rload at a regulated voltage, as shown by curve 1165, part of the energy stored in energy storage stage 1112 is delivered to the load. The voltage of node Vout_ACX, therefore, may decrease during times when energy is not being transferred to the secondary side of transformer 216, such as between time t1 and t2, as shown by curve 1164. For example, the voltage of node Vout_ACX may decrease from a peak voltage of about 43 V to a voltage of about 21 V.
When the AC signal is a low-line signal, ACX converter 1108 operates in low-line mode with bidirectional switch 1149 open, bidirectional switch 1151 closed and transistors 1140 and 1144 off, as shown in
The capacitance of energy storage stage 1112 may be given by
where Pout is the maximum power of the converter, td is the discharge time, for example, as shown in
Since when node Vin_ACX node receives a low-line signal capacitors 1114 and 1115 are independently charged and when during high-line signals capacitors 1114 and 1115 are charged in series, the energy stored by capacitors 1114 and 1115 may be higher in low-line mode than in high-line mode. The minimum peak voltage of node Vout_ACX when node Vin_ACX node receives a low-line signal may be higher than when node Vin_ACX receives a high-line signal. An additional benefit of operating in low-line mode is that switching losses may be lower than during high-line mode since transistors 1140 and 1144 do not switch during low-line mode.
Advantages of some embodiments of the present invention include that the DC-DC converter may be optimized for a particular DC-DC input voltage irrespective of the mains voltage. Other advantages includes that operating with low-line input signal may result in an increase in efficiency.
During normal operation, ACX converter 1008 receives an AC signal at node Vin_ACX and produces a rectified voltage at node Vout_ACX. Energy storage stage 1212 stores energy and may also reduce the voltage ripple of node Vout_ACX. DC-DC converter 1222 receives the rectified voltage of node Vout_ACX and produces a regulated voltage at node Vout. ACX converter 1008 may operate, for example, as described with respect to
Since the amount of energy stored in a capacitor is proportional to the voltage across the capacitor, energy storage stage 1212 may turn on transistors 1215 during a low-line input mode to increase the amount of capacitance available to, for example, double the amount. Alternatively, energy storage stage 1212 may be implemented without transistor 1215 and capacitor 1214.
DC-DC converter 1222 may be configured to switch in a high-line mode or low-line mode depending on the input that ACX converter 1008 receives. For example, when ACX converter 1008 receives a low-line input, the voltage of node Vout_ACX may be, for example, about 35 V. DC-DC converter 1222, therefore, may transfer energy from capacitors 1014 and 1015, simultaneously, to load Rload. When ACX converter 1008 receives a high-line input, the voltage of node Vout_ACX may be twice the voltage compared to the voltage when the ACX converter 1008 receives a low-line input. Therefore, DC-DC converter 1222 may transfer energy from either capacitor 1014 or 1015 and alternating cycle to cycle.
DC-DC converter 1222 may regulate the voltage of node Vout, for example, to 20 V, 18 V, 12 V, 10 V, 5 V, 3.3 V, 1.8 V, 1.2 V, or 1V. Other values may be used. DC-DC converter 1222 may be implemented according to various ways known in the art and may be configured to regulate the voltage while complying with a particular standard such as, for example, USB-PD.
Controller 1245 is configured to produce signals S230, S234, S1038, S1040, S1270, S1272, S1274, S1276, and S1215 to drive bidirectional switches 230 and 234, transistors 1238, 1240, 1270, 1272, 1274, 1276, and 1215, respectively. Coupling controller 1245 to bidirectional switches 230 and 234, and transistors 1238, 1240, 1270, 1272, 1274, 1276 and 1215 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 1245 from other parts of the circuit. Coupling between controller 1245 and other components of converter 1200 may also be achieved in other ways known in the art.
Controller 1245 may be implemented as a single chip. For example, controller 1245 may be implemented in a monolithic substrate. Alternatively, controller 1245 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX converter 1008 and energy storage stage 1212, and a controller for controlling DC-DC converter 1222. Other implementations known in the art are also possible.
As shown in
As shown in
As shown in
As shown in
As shown by
As shown in
During high-line signal, transistor 1215 is off and transistors 1270, 1272, 1274 and 1276 transition between the third state, second state, fourth state, second state and back to the third state, repeating the sequence. The delays between the switching signals as DC-DC converter 1222 transitions between states are used to allow for ZVS switching.
Advantages of some embodiments of the present invention include that the ACX secondary circuit may conduct a current through one switch at any time. Conduction losses, therefore, may be smaller than in other embodiments. Additionally, since the DC-DC converter operates with either a high input voltage or a low input voltage, the ACX converter may operate without being configured based on the whether the input is high-line or low-line.
During normal operation, ACX converter 908 receives an AC signal at node Vin_ACX and produces a rectified voltage at node Vout_ACX. ACX converter 908 may operate, for example, as described with respect to
Since the amount of energy stored in a capacitor is proportional to the voltage across the capacitor, energy storage stage 1312 may turn on transistors 1315 during a low-line input mode to increase the amount of capacitance available to, for example, double the amount. Alternatively, energy storage stage 1312 may be implemented without transistor 1315 and capacitor 1314.
Since DC-DC converter 1322 is implemented as an inverted buck-boost converter, DC-DC converter 1322 may produce a regulated output irrespective of whether the input is a high-line input or a low-line input. For example, when the voltage of node Vin_ACX is a high-line voltage, DC-DC converter 1322 may step down the voltage for the majority of the time. When the voltage of node Vin_ACX is a low-line voltage, DC-DC converter 1322 may step down the voltage during some times and step up the voltage during other times.
DC-DC converter 1322 may regulate the voltage across Rload to, for example, 20 V, 18 V, 12 V, 10 V, 5 V, 3.3 V, 1.8 V, 1.2 V, or 1 V. Other values may be used. The voltage at node Vout may be referred to as a negative voltage. DC-DC converter 1322 may be implemented according to various ways known in the art and may be configured to regulate the voltage while complying with a particular standard such as, for example, USB-PD.
Controller 1345 is configured to produce signals S230, S234, S938, S940, S1370, S1372, and S1315 to drive bidirectional switches 230 and 234, and transistors 938, 940, 1370, 1372, and 1315, respectively. Coupling controller 1345 to bidirectional switches 230 and 234, and transistors 938, 940, 1370, 1372, and 1315 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 1345 from other parts of the circuit. Coupling between controller 1345 and other components of converter 1300 may also be achieved in other ways known in the art.
Controller 1345 may be implemented as a single chip. For example, controller 1345 may be implemented in a monolithic substrate. Alternatively, controller 1345 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX converter 908 and energy storage stage 1312, and a controller for controlling DC-DC converter 1322. Other implementations known in the art are also possible.
As shown in
Advantages of some embodiments of the present invention include operating the ACX converter without configuring the ACX converter based on the whether the input is high-line or low-line. Other advantages include that a converter may be implemented with two bidirectional switches and five transistors.
During normal operation, ACX converter 908 receives an AC signal at node Vin_ACX and produces a rectified voltage at node Vout_ACX. ACX converter 908 may operate, for example, as described with respect to
Since DC-DC converter 1422 is implemented as a non-inverted buck-boost converter, DC-DC converter 1422 may produce a regulated output irrespective of whether the ACX converter 1408 receives a high-line voltage or a low-line voltage. For example, when ACX converter 1408 receives a high-line voltage, DC-DC converter 1422 may step down the voltage for the majority of the time. When ACX converter 1408 receives a low-line voltage, DC-DC converter 1422 may step up the voltage for the majority of the time.
DC-DC converter 1422 may regulate the voltage of node Vout to, for example, 20 V, 18 V, 12 V, 10 V, 5 V, 3.3 V, 1.8 V, 1.2 V, or 1 V. Other values may be used. DC-DC converter 1422 may be implemented according to various ways known in the art and may be configured to regulate the voltage while complying with a particular standard such as, for example, USB-PD.
Controller 1445 is configured to produce signals S230, S234, S938, S940, S1470, S1472, S1474, S1476 and S1315 to drive bidirectional switches 230 and 234, and transistors 938, 940, 1470, 1472, 1474, 1476, and 1315, respectively. Coupling controller 1445 to bidirectional switches 230 and 234, and transistors 938, 940, 1470, 1472, 1474, 1476, and 1315 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 1345 from other parts of the circuit. Coupling between controller 1445 and other components of converter 1400 may also be achieved in other ways known in the art.
Controller 1445 may be implemented as a single chip. For example, controller 1445 may be implemented in a monolithic substrate. Alternatively, controller 1445 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX converter 908 and energy storage stage 1312, and a controller for controlling DC-DC converter 1422. Other implementations known in the art are also possible.
As shown in
As shown in
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As shown in
As shown in
When DC-DC converter 1422 steps up the voltage with a low-line input, transistor 1315 is on and transistors 1470, 1472, 1474 and 1476 alternate between the third state and the fourth state, as shown in
As shown in
During normal operation, ACX converter 1508 receives an AC signal at node Vin_ACX and produces a rectified voltage at node Vout_ACX. Energy storage stage 912 stores energy and may also reduce the voltage ripple of node Vout_ACX. DC-DC converter 1122 receives the rectified voltage of node Vout_ACX and produces a regulated voltage at node Vout.
More particularly, the switching and operation of DC-DC converter 1122 may be similar to that of DC-DC converter 1122, as illustrated in
Transformer 1516 may be configured in a first state with primary winding 1518 in series with primary winding 1519 by closing bidirectional switch 1523 and opening bidirectional switches 1525 and 1527. Alternatively, transformer 1516 may be configured in a second state with primary winding 1518 in parallel with primary winding 1519 by opening bidirectional switch 1523 and closing bidirectional switches 1525 and 1527. When transformer 1516 is configured in the first state, transformer 1516 may have a turn ratio of 2n to 1. When transformer 1516 is configured in the second state, transformer 1516 may have a turn ratio of n to 1.
ACX converter 1508 may configure transformer 1516 to the first state when the voltage of node Vin_ACX is a high-line voltage and to the second state when the voltage of node Vin_ACX is a low-line voltage. By configuring ACX converter 1508 in a first and second state when the voltage node Vin_ACX is a high-line voltage or a low-line voltage, respectively, ACX converter 1508 produces a voltage at node Vout_ACX with a peak amplitude that does not substantially change based on whether the input voltage is a high-line voltage or a low-line voltage. Energy storage stage 912, therefore, may be implemented with capacitor 914, without using additional transistors.
Since the peak amplitude of voltage of node Vout_ACX does not substantially vary based on whether the input voltage of ACX converter 1508, DC-DC converter 1122 may be implemented as a buck converter, as shown in
Bidirectional switches 1523, 1525 and 1527 may be implemented according to various ways known in the art. For example, bidirectional switches 1523, 1525 and 1527 may be implemented with the topologies shown in
Controller 1545 is configured to produce signals S230, S234, S1523, S1525, S1527, S938, S940, S1153 and S1155 to drive bidirectional switches 230, 234, 1523, 1525, 1527, and transistors 938, 940, 1153 and 1155, respectively. Coupling controller 1545 to bidirectional switches 230, 234, 1523, 1525, 1527, and transistors 938, 940, 1153 and 1155 may be achieved through direct electrical connection or indirect electrical connections. For example, opto-couplers may be used to electrically isolate controller 1145 from other parts of the circuit. Coupling between controller 1545 and other components of converter 1500 may also be achieved in other ways known in the art.
Controller 1545 may be implemented as a single chip. For example, controller 1545 may be implemented in a monolithic substrate. Alternatively, controller 1545 may be implemented as a collection of controllers, such as, for example, a controller for controlling ACX converter 1508, and a controller for controlling DC-DC converter 1122. Other implementations known in the art are also possible.
As shown in
Advantages of some embodiments of the present invention include simplifying the energy storage state by implementing a transformer with a configurable turn ratio based on the input voltage. Other advantages include implementing a converter with five bidirectional switches and four transistors.
Converters using an ACX converter stage may also be implemented with PFC. For example,
During normal operation, converter 1600 may operate in a similar manner as converter 200. Converter 1600, however, operates ACX converter 1608 with PFC, instead of without PFC.
ACX converter 1608 may achieve PFC by operating with a fifth mode of control. When ACX converter 1608 is operated with the fifth mode of control, bidirectional switches 230 and 234 continuously switch at a constant frequency and a constant duty cycle. Similarly, the transistors of the secondary circuit of ACX converter 1608 continuously switch. In other words, ACX converter 1608 may transfer energy from the primary side of the transformer of ACX converter 1608 to the secondary side of the transformer and vice-versa. The forward energy transfer rule, as given by Equation 1, may not be followed in the fifth mode of control.
ACX converter 1608 may implement DC-DC converter 1622 with PFC, as opposed to without PFC. The implementation of DC-DC converters with PFC are known in the art, and any DC-DC converter implementation with PFC may be used.
Since ACX converter 1608 is configured to receive an AC signal, ACX converter 1608 may operate with a small input capacitor Cin. The main energy storage, however, may be implemented in output capacitor Cout rather than in energy storage stage 1612. Therefore, the capacitors of energy storage stage 1612 may also be small. As illustrated in
The switching and operation of ACX converter 908 may be similar to that of ACX converter 908 as illustrated in
As shown in
Advantages of some embodiments of the present invention include that converters utilizing an ACX converter may be implemented with PFC and without PFC. ACX converters, therefore, may be useful for implementing power supplies in a wide power delivery range. For example, embodiments of the present invention may be configured to deliver power levels of 1 W or less. Other embodiments may be configured to deliver power levels of 65 W, 100 W or higher. Other power delivery levels may be used.
Some converters may exhibit output ripple in the output voltage. For example, output ripple at twice the mains frequency may be present in the output voltage. Some converters having a converter stage using an ACX converter with PFC may reduce output ripple by using various techniques. For example,
During normal operation, converter 1700 may operate in a similar manner as converter 1600. Converter 1700, however, has buffer capacitor Cbuf in series with load Rload. To maintain a regulated output, series-power-pulsation buffer 1701 may control the voltage across buffer capacitor Cbuf such that Vout=V0+Vb is constant.
Series-power-pulsation buffer 1701 may be implemented according to various ways known in the art. For example, series-power-pulsation buffer 1701 may include a buck or buck-boost converter coupled from auxiliary capacitor Caux to buffer capacitor Cbuf. Other implementations are also possible.
During normal operation, converter 1800 may operate in a similar manner as converter 1600. Converter 1800, however, has compensation stage 1801 coupled in parallel to load Rload. To maintain a regulated output, compensation stage 1801 may transfer energy from auxiliary capacitor Caux to output capacitor Cout and transfer energy from output capacitor Cout to auxiliary capacitor Caux.
Compensation stage 1801 may be implemented according to various ways known in the art. For example, compensation stage 1801 may include a buck or boost converter coupled between auxiliary capacitor Caux and output capacitor Cout. Other implementations are also possible.
A converter including: a rectifying stage having a first supply terminal and a second supply terminal, the first supply terminal and the second supply terminal configured to receive a bipolar AC signal from an AC power source, the rectifying stage including a half-bridge circuit coupled between the first supply terminal and the second supply terminal, a transformer, and a resonant tank coupled between an output of the half-bridge circuit and a primary winding of the transformer; and a DC-DC converter stage coupled between the rectifying stage and an output terminal.
The converter of example 1, where the resonant tank includes a resonant capacitor, a first resonant inductor and a second resonant inductor.
The converter of one of examples 1 or 2, where an output of the DC-DC converter stage is configured to provide power to a USB power delivery (USB-PD) interface.
The converter of one of examples 1 to 3, where the half-bridge circuit includes: a first bidirectional switch coupled between the first supply terminal and the output of the half-bridge circuit; and a second bidirectional switch coupled between the output of the half-bridge circuit and the second supply terminal.
The converter of one of examples 1 to 4, where the first bidirectional switch is turned off and the second bidirectional switch is turned on when a voltage between the first and second supply terminal of the rectifying stage is lower than an output of the rectifying stage multiplied by a first factor.
The converter of one of examples 1 to 5, further including a controller configured to turn on and off the first bidirectional switch and the second bidirectional switch with a constant frequency and a constant duty cycle.
The converter of one of examples 1 to 6, where the controller turns on the first bidirectional switch with zero voltage switching (ZVS) or quasi-ZVS (QZVS).
The converter of one of examples 1 to 7, where the rectifying stage further includes a switching network coupled to a first secondary winding of the transformer.
The converter of one of examples 1 to 8, further including a controller configured to turn on and off transistors of the switching network when a voltage between the first and second supply terminal of the rectifying stage is lower than an output of the rectifying stage multiplied by a first factor.
The converter of one of examples 1 to 8, further including a controller configured to turn off transistors of the switching network when a voltage between the first and second supply terminal of the rectifying stage is lower than an output of the rectifying stage multiplied by a first factor.
The converter of one of examples 1 to 10, where the first factor is based on a turning ratio of the transformer.
The converter of one of examples 1 to 11, where the switching network includes a first transistor coupled between a first terminal of the first secondary winding and a first switching terminal, a second transistor coupled between the first terminal of the first secondary winding and a second switching terminal; and the DC-DC converter stage is coupled between the first switching terminal and the second switching terminal.
The converter of one of examples 1 to 12, where the switching network further includes: a first capacitor coupled between the first switching terminal and a second terminal of the first secondary winding; and a second capacitor coupled between the second terminal of the first secondary winding and the second switching terminal.
The converter of one of examples 1 to 13, where the switching network further includes: a third transistor coupled between the first switching terminal and a second terminal of the first secondary winding; a fourth transistor coupled between the second terminal of the first secondary winding and the second switching terminal; and a first capacitor coupled between the first switching terminal and the second switching terminal.
The converter of one of examples 1 to 14, where the switching network further includes: a first bidirectional switch coupled between the fourth transistor and the second terminal of the first secondary winding.
The converter one of examples 1 to 11, where the switching network includes a first transistor coupled between a first terminal of the first secondary winding and a first switching terminal; a second transistor coupled between a second terminal of a second secondary winding and the first switching terminal; and a first capacitor coupled between the first switching terminal and a second switching terminal, the second switching terminal coupled to a second terminal of the first secondary winding and a first terminal of the second secondary winding; and the DC-DC converter stage coupled between the first switching terminal and the second switching terminal.
The converter one of examples 1 to 11 and 16, where the primary winding of the transformer includes a first portion of the primary winding coupled to a second portion of the primary winding via a first switch.
The converter one of examples 1 to 11 and 16 to 17, where the first switch includes a mechanical relay.
The converter one of examples 1 to 11 and 16 to 18, where the DC-DC converter stage includes a non-inverted buck-boost converter.
The converter one of examples 1 to 11 and 16 to 18, where the DC-DC converter stage includes a boost converter.
The converter one of examples 1 to 4, 6 to 18, where the DC-DC converter stage includes a boost converter with power factor correction (PFC).
A method of operating a converter including: receiving a bipolar AC signal from an AC power source with a half-bridge circuit coupled to a resonant tank, where the resonant tank includes a first resonant capacitor, a first resonant inductor and a second resonant inductor; activating the resonant tank; rectifying the bipolar AC signal with a switching network to produce a rectified signal; galvanically isolating the half-bridge circuit from the switching network; and converting the rectified signal to a first voltage with a DC-DC converter.
The method of example 22, where activating the resonant tank includes: turning on and off a first bidirectional switch of the half-bridge circuit at a constant frequency and a constant duty cycle; and turning on and off a second bidirectional switch of the half-bridge circuit at a constant frequency and a constant duty cycle.
The method of one of examples 22 or 23, where gavanically isolating the half-bridge circuit from the switching network includes using a transformer coupled between the half-bridge circuit and the switching network; and the rectifying the bipolar AC signal further includes turning on and off transistors of the switching network.
The method of one of examples 22 to 24, where the rectifying the bipolar AC signal further includes turning off transistors of the switching network when the bipolar AC signal is lower than the rectified signal multiplied by a first factor.
The method of one of examples 22 to 25, where the rectifying the bipolar AC signal further includes turning off the first bidirectional switch and turning on the second bidirectional switch when a voltage across a secondary winding of the transformer is larger than a voltage across a primary winding of the transformer multiplied by a first factor.
The method of one of examples 22 to 26, where the bipolar AC signal includes a root-mean-square (RMS) voltage between 85 V and 140 V and the first voltage includes a DC level between 3 V and 20 V.
The method of one of examples 22 to 26, where the bipolar AC signal includes an RMS voltage between 200 V and 270 V and the first voltage includes a DC level larger than 3 V.
A resonant converter including: a half-bridge circuit configured to receive a bipolar AC signal, the half-bridge circuit including a first bidirectional switch coupled between a first supply terminal and a second supply terminal; a second bidirectional switch coupled between the first bidirectional switch and the second supply terminal; and a resonant tank coupled between the half-bridge circuit and a primary winding of a transformer, where the first bidirectional switch and the second bidirectional switch turn on and off at a constant frequency and a constant duty cycle.
The resonant converter of example 29, where the resonant tank includes a resonant capacitor, a first resonant inductor, and a second resonant inductor.
The resonant converter of one of examples 29 or 30, where the transformer includes the first resonant inductor.
The resonant converter of one of examples 29-31, further including a switching network coupled between a secondary winding of the transformer and an output terminal.
The resonant converter of one of examples 29-32, where switches of the switching network are configured to turn off when a voltage of the bipolar AC signal is lower than a voltage of the output terminal multiplied by a first factor.
While this invention has been described with reference to illustrative embodiments, this description is not intended to be construed in a limiting sense. Various modifications and combinations of the illustrative embodiments, as well as other embodiments of the invention, will be apparent to persons skilled in the art upon reference to the description. It is therefore intended that the appended claims encompass any such modifications or embodiments.
