The present invention is directed to an antenna (spatial) processing useful in wireless communication applications, such as short-range wireless applications.
Composite Beamforming (CBE) is an antenna processing technique in which a first communication device, having a plurality of antennas, weights a signal to be transmitted by its antennas to a second communication device also having a plurality of antennas. Similarly, the second communication device weights and combines the received signals received by its antennas. A multiple-input/multiple-output (MIMO) optimized communication system is defined by CBF. The transmit weights and receive weights are determined to optimize the link margin between the devices, thereby significantly extending the range of communication between the two communication devices. Techniques related to composite beamforming are the subject matter of above-identified commonly assigned co-pending application.
There is room for still further enhancing this CBF technique to optimize cost and implementation issues at the expense of only slight degradation in performance. Such a solution is extremely valuable in manufacturing a cost-effective integrated circuit solution.
An equal gain composite beamforming technique is provided that adds the constraint that the power of the signal output by each of the plurality of transmit antennas is the same, and is equal to the total power of the transmit signal divided by the number N of transmit antennas from which the signal is to be transmitted. This reduces output power requirements at each antenna. In an example, the signal may be a multi-carrier signal. Further, a weight may be produced for each of the N transmit antennas. By reducing output power requirements for each power amplifier, the silicon area of the power amplifiers are reduced by as much as N times (where N is the number of transmit antennas) relative to non-equal gain CBF. Many implementation advantages are achieved by equal gain CBF, including savings in silicon, power requirements, etc.
Moreover, an amplifier may be used to amplify a signal. Also, the multi-carrier signal may be an orthogonal frequency division multiplex signal, in an example. In addition, the multi-carrier signal may have a plurality of K subcarriers. In a further example, a power may be applied to each of the K subcarriers per antenna which is equal to the total transmit power divided by KN. Further, the multi-carrier signal may be weighted for each of the N antennas per the produced weight. In yet a further example, each weight may vary with a sub-band. Additionally, the multi-carrier signal may be transmitted on an uplink channel. In still another example, the total transmit power may be scaled. The multi-carrier signal may be filtered in another example.
The above and other objects and advantages will become more readily apparent when reference is made to the following description taken in conjunction with the accompanying drawings.
FIG.6FIG. 6 is a graphical diagram illustrating convergence of the adaptive algorithm shown in
Referring first to
The transmit weight vectors wtx,1 and wtx2wtx,2 each comprises a plurality of transmit weights corresponding to each of the N and M antennas, respectively. Each transmit weight is a complex quantity. Moreover, each transmit weight vector is frequency dependent; it may vary across the bandwidth of the baseband signal s to be transmitted. For example, if the baseband signal s is a multi-carrier signal of K sub-carriers, each transmit weight for a corresponding antenna varies across the K sub-carriers. Similarly, if the baseband signal s is a single-carrier signal (that can be divided or synthesized into K frequency sub-bands), each transmit weight for a corresponding antenna varies across the bandwidth of the baseband signal. Therefore, the transmit weight vector is dependent on frequency, or varies with frequency sub-band/sub-carrier k, such that wtx wtx becomes wtx(f), or more commonly referred to as wtx(k), where k is the frequency sub-band/sub-carrier index.
While the terms frequency sub-band/sub-carrier are used herein in connection with beamforming in a frequency dependent channel, it should be understood that the term “sub-band” is meant to include a narrow bandwidth of spectrum forming a part of a baseband signal. The sub-band may be a single discrete frequency (within a suitable frequency resolution that a device can process) or a narrow bandwidth of several frequencies.
The receiving communication device also weights the signals received at its antennas with a receive antenna weight vector wrx(k). Communication device 100 uses a receive antenna weight vector wrx,1(k) when receiving a transmission from communication device 200, and communication device 200 uses a receive antenna weight vector wrx,2(k) when receiving a transmission from communication device 100. The receive antenna weights of each vector are matched to the received signals by the receiving communication device. The receive weight vector may also be frequency dependent.
Generally, transmit weight vector wtx,1 comprises a plurality of transmit antenna weights wtx,1,i=β1,i(k)eiϕ1,i,(k), where β1,i(k) is the magnitude of the antenna weight, ϕ1,i,(k) is the phase of the antenna weight, i is the antenna index, and k is the frequency sub-band or sub-carrier index (up to K frequency sub-bands/sub-carriers). The subscripts tx,1 denote that it is a vector that communication device 100 uses to transmit to communication device 2. Similarly, the subscripts tx,2 denote that it is a vector that communication device 200 uses to transmit to communication device 100.
Under the constraint of an equal gain composite beamforming (EGCBF) process, the power of the transmit signal output by each transmit antenna is the same, and is equal to the total power associated with the transmit signal (Ptx) divided by the number of transmit antennas. Thus, for communication device 100, that is Ptx/N. For communication device 200, that is Ptx/M. Consequently, each power amplifier associated with an antenna need only support 1/N of the total output power. Example: For N=4, Ptx=17 dBm, each power amplifier need only support a max linear output power of 17−10log(4)=11 dBm. Thus, whereas for non-equal gain composite beamforming each power amplifier must support to the total transmit power, such is not the case for equal gain beamforming. The equal-gain constraint makes the power amplifier design much simpler. Equal gain CBF performs very close to non-equal gain CBF (within 1-2 dB), but costs significantly less to implement in terms of power amplifier requirements and affords the opportunity to deploy the power amplifiers on the same silicon integrated circuit as the RF circuitry.
When considering a frequency dependent communication channel, the EGCBF constraint implies that for each and every antenna i, the sum of the power |wtx,i(k)|2 of the antenna signal across all of frequencies of the baseband signal (the frequency sub-bands or sub-carrier frequencies k=1 to K) is equal to Pix/N. This is the equal gain constraint applied to a frequency dependent channel represented by K sub-carrier frequencies or frequency sub-bands.
An additional constraint can be imposed on the frequency dependent equal gain constraint explained above. This additional constraint is a frequency shaping constraint which requires that at each frequency of the baseband signal to be transmitted (e.g., frequency sub-band or frequency sub-carrier k), the sum of the power of signals across all of the transmit antennas (|wtx,i(k)|2 for i=1 to N) is equal to Ptx/K. This constraint is useful to ensure that, in an iterative process between two communication devices, the transmit weights of the two devices will converge to optimal values. An additional benefit of this constraint is that the transmitting device can easily satisfy spectral mask requirements of a communication standard, such as IEEE 802.11x.
One solution to a system that combines the frequency selective equal gain constraint and the frequency shaping constraint is that |wtx,i(k)|2=Ptx/(KN). This solution says that the magnitude of each transmit antenna weight is Ptx/(KN). If the transmit weight vector is normalized to unity, i.e., divided by [Ptx/(KN)]1/2, then the vectors for all of the antennas across all of the k frequency sub-bands becomes a N×K matrix of phase values eiϕik, where i is the antenna index and k is the sub-band/sub-carrier index.
For EGCBF, the optimization problem becomes
argmax{wHtx,1(k)HH(k)H(k)wtx,1(k)}, subject to |wtx,i(k)|2=Ptx/(NK). (1)
A closed-form solution to equation (1) is difficult to obtain since it requires the solution of a non-linear system of equations. However, the following necessary conditions for the solution to (1) have been derived and are given below:
Optimal wrx satisfies wrx=kHwtx for some nonzero constant k.
Optimal wtx satisfies Im(Λ*HHHeiϕ)=0, Λ=diag(eiϕ0, eiϕ1, . . . , eiϕN-1), wtx=eiϕ=(eiϕ0, eiϕ1, . . . , eiϕN-1)T. One solution to equation (1) is an adaptive algorithm for EGCBF. Although the algorithm is not necessarily optimal in terms of solving equation (1), it is simple to implement and simulations have verified that it converges reliably at the expense of only a 1-2 dB performance penalty relative to non-equal gain CBF. This adaptive algorithm is described hereinafter in conjunction with
The communication devices at both ends of the link ,link, i.e., devices 100 and 200 may have any known suitable architecture to transmit, receive and process signals. An example of a communication device block diagram is shown in
The intelligence to execute the computations for the composite beamforming techniques described herein may be implemented in a variety of ways. For example, a processor 322 in the baseband section 320 may execute instructions encoded on a processor readable memory 324 (RAM, ROM, EEPROM, etc.) that cause the processor 322 to perform the composite beamforming steps described herein. Alternatively, an application specific integrated circuit (ASIC) may be configured with the appropriate firmware, e.g., field programmable gates that implement digital signal processing instructions to perform the composite beamforming steps. This ASIC may be part of, or the entirety of, the baseband section 320. Still another alternative is for the beamforming computations to be performed by a host processor 332 (in the host 330) by executing instructions stored in (or encoded on) a processor readable memory 334. The RF section 310 may be embodied by one integrated circuit, and the baseband section 320 may be embodied by another integrated circuit. The communication device on each end of the communication link need not have the same device architecture or implementation.
Regardless of the specific implementation chosen, the composite beamforming process is generally performed as follows. A transmit weight vector (comprising a plurality of complex transmit antenna weights corresponding to the number of transmit antennas) is applied to, i.e., scaled or multiplied by, a baseband signal to be transmitted, and each resulting weighted signal is coupled to a transmitter where it is upconverted, amplified and coupled to a corresponding one of the transmit antennas for simultaneous transmission. At the communication device on the other end of the link, the transmit signals are detected at each of the plurality of antennas and downconverted to a baseband signal. Each baseband signal is multiplied by a corresponding one of the complex receive antenna weights and combined to form a resulting receive signal. The architecture of the RF section necessary to accommodate the beamforming techniques described herein may vary with a particular RF design, and many are known in the art and thus is not described herein.
With reference to
In step 410, a baseband signal is scaled by the initial AP transmit weight vector wT,A,P,0(k), upconverted and transmitted to the STA. The transmitted signal is altered by the frequency dependent channel matrix H(k) from AP-STA. The STA receives the signal and matches its initial receive weight vector wR,STA,0(k) to the signals received at its antennas. In step 420, the STA gain normalizes the receive weight vector wR,STA,0(k) and computes the conjugate of gain-normalized receive weight vector to generate the STA's initial transmit weights for transmitting a signal back to the AP. The STA scales the signal to be transmitted to the AP by the initial transmit weight vector, upconverts that signal and transmits it to the AP. Computing the conjugate for the gain-normalized vector means essentially co-phasing the receive weight vector (i.e., phase conjugating the receive weight vector). The transmitted signal is effectively scaled by the frequency dependent channel matrix HT(k). At the AP, the receive weight vector is matched to the signals received at its antennas. The AP then computes the conjugate of the gain-normalized receive weight vector as the next transmit weight vector wT,AP,1(k) and in step 430 transmits a signal to the STA with that transmit weight vector. The STA receives the signal transmitted from the AP with this next transmit weight vector and matches to the received signals to compute a next receive weight vector wR,STA,1(k). Again, the STA computes the conjugate of the gain-normalized receive weight vector wR,STA,1(k) as its next transmit weight vector wT,STA,1(k) for transmitting a signal back to the AP. This process repeats for several iterations, ultimately converging to transmit weight vectors that achieve nearly the same performance as non-equal gain composite beamforming. This adaptive process works even if one of the devices, such as a STA, has a single antenna for transmission and reception. In addition, throughout the adaptive process, the frequency shaped constraint may be maintained so that for each antenna, the transmit antenna weight is constant across the bandwidth of the baseband signal.
Each communication device stores the transmit antenna weights determined to most effectively communicate with each of the other communication devices that it may communicate with. The transmit antenna weights may be determined according to the adaptive algorithm described above. When storing the transmit weights of a transmit weight vector, in order to conserve memory space in the communication device, the device may store, for each antenna, weights for a subset or a portion of the total number of weights that span the bandwidth of the baseband signal. For example, if there are K weights for K frequency sub-bands or sub-carrier frequencies, only a sampling of those weights are actually stored, such as weights for every other, every third, every fourth, etc., k sub-band or sub-carrier. Then, the stored subset of transmit weights are retrieved from storage when a device is to commence transmission of a signal, and the remaining weights are generated by interpolation from the stored subset of weights. Any suitable interpolation can be used, such as linear interpolation, to obtain the complete set of weights across the K sub-bands or sub-carriers for each antenna.
With reference to
When a STA associates or whenever a significant change in channel response is detected, the AP sends a special training sequence to help the STA select the best of its two antennas. The training sequence uses messages entirely supported by the applicable media access control protocol, which in the following example is IEEE 802.11x.
The sequence consists of 2 MSDUs (ideally containing data that is actually meant for the STA so as not to incur a loss in throughput). In step 900, the AP sends the first MSDU using a Tx weight vector having equal gain orthogonal transmit weights (also optionally frequency shaped). For example, if the number of AP antennas is 4, the transmit weight vector is [1 1 1 1]T. In step 910, the 2-antenna selection diversity STA responds by transmitting a message using one of its'its two antennas; the antenna that best received the signal from the AP. The AP decodes the message from the STA, and obtains one row of the H matrix (such as the first row hr1). In step 920, the AP sends the second MSDU using a frequency dependent transmit weight vector which is orthogonal to the first row of H (determined in step 610910), and equal-gain normalized such that the magnitude of the signals at each antenna is equal to P/N. In addition, the transmit weight vector may be frequency shaped across so that at each frequency of the baseband signal, the sum of the power output by the antennas of the first communication device is constant across. When the STA receives the second MSDU, it forces the STA to transmit a response message in step 630930 using the other antenna, allowing the AP to see the second row of the H matrix, such as hr2. Now the AP knows the entire H matrix. The AP then decides which row of the H matrix will provide “better” MRC at the STA by computing a norm of each row, hr1 and hr2, of the H matrix and, and selecting the row that has the greater norm as the frequency dependent transmit weight vector for further transmissions to that STA until another change is detected in the channel. The row that is selected is equal gain normalized before it is used for transmitting to that STA.
Equal gain composite beamforming provides significant advantages. By reducing output power requirements for each power amplifier, the silicon area of the power amplifiers are reduced by as much as N times (where N is equal to the number of antennas) relative to non-equal gain CBF. The silicon area savings translates into a cost savings due to (1) less silicon area, and (2) the ability to integrate the power amplifiers onto the same die (perhaps even the same die as the radio frequency transceiver itself).
The efficiency (efficiency being defined as the output power divided by DC current consumption) for each power amplifier is N times larger in the EGCBF case than in the non-equal gain CBF case. With EGCBF, the same output power as non-equal CBF is achieved with N times less DC current.
Equal gain CBF reduces power requirements for each of the power amplifiers, which in turn increases the output impedance of each of the power amplifiers (since impedance is inversely proportional to current, and supply current requirements will be reduced). When the output impedance of the power amplifier is increased, the matching networks required for the power amplifiers are greatly simplified and require less silicon area. Moreover, reducing power requirements for the individual power amplifiers provides greater flexibility for systems with low supply voltage.
To summarize, a method is provided that accomplishes applying a transmit weight vector to a baseband signal to be transmitted from the first communication device to the second communication device, the transmit weight vector comprising a complex transmit antenna weight for each of N plurality of antennas of the first communication device, wherein each complex transmit antenna weight has a magnitude and a phase whose values may vary with frequency across a bandwidth of the baseband signal, thereby generating N transmit signals each of which is weighted across the bandwidth of the baseband signal, wherein the magnitude of the complex transmit antenna weight associated with each antenna is such that the power to be output at each antenna is the same and is equal to the total power to be output by all of the N antennas divided by N; receiving at the N plurality of antennas of the first communication device a signal that was transmitted by the second communication device; determining a receive weight vector comprising a plurality of complex receive antenna weights for the N plurality of antennas of the first communication device from the signals received by the N plurality of antennas, wherein each receive antenna weight has a magnitude and a phase whose values may vary with frequency: and updating the transmit weight vector for the N plurality of antennas of the first communication device for transmitting signals to the second communication device by gain normalizing the receive weight vector and computing a conjugate thereof. This process may be performed such that at substantially all frequencies of the baseband signal, the sum of the magnitude of the complex transmit antenna weights across the plurality of antennas of the first communication device is constant. Moreover, where the bandwidth of the baseband signal comprises K plurality of frequency sub-bands, the magnitude of the complex transmit antenna weights associated with each of the N plurality of antennas is such that the power to be output by each antenna is the same and is equal to 1/(KN) of the total power to be output for all of the K frequency sub-bands. This process may be embodied by instructions encoded on a medium or in a communication device.
In addition, a method is provided that accomplishes a method for communicating signals from a first communication device to a second communication device using radio frequency (RF) communication techniques, comprising: applying to a first signal to be transmitted from the first communication device to the second communication device a transmit weight vector, the transmit weight vector comprising a complex transmit antenna weight for each of the N plurality of antennas, wherein each complex transmit antenna weight has a magnitude and a phase whose values may vary with frequency across a bandwidth of the baseband signal, thereby generating N transmit signals each of which is weighted across the bandwidth of the baseband signal, wherein the magnitude of the complex transmit antenna weights associated with each antenna is such that the power to be output at each antenna is the same and is equal to the total power to be output by all of the N antennas divided by N; from a first response signal at the plurality of antennas of the first communication device transmitted from a first of two antennas of the second communication device, deriving a first row of a channel response matrix that describes the channel response between the first communication device and the second communication device; applying to a second signal for transmission by the plurality of antennas of the first communication device a transmit weight vector that is orthogonal to the first row of the channel response matrix, and wherein the transmit weight vector comprises a plurality of complex transmit antenna weights, wherein each complex transmit antenna weight has a magnitude and a phase whose values may vary with frequency across a bandwidth of the second signal, thereby generating N transmit signals each of which is weighted across the bandwidth of the baseband signal, wherein the magnitude of the complex transmit antenna weights associated with each antenna is such that the power to be output at each antenna is the same and is equal to the total power to be output by all of the N antennas divided by N; deriving a second row of the channel response matrix from a second response signal received from a second of the two antennas of the second communication device; and selecting one of the first and second rows of the channel response matrix that provides better signal-to-noise at the second communication device as the transmit weight vector for further transmission of signals to the second communication device. This process may be performed such that at substantially all frequencies of the baseband signal, the sum of the magnitude of the complex transmit antenna weights across the plurality of antennas of the first communication device is constant. Moreover, where the bandwidth of the baseband signal comprises K plurality of frequency sub-bands, the magnitude of the complex transmit antenna weights associated with each of the N plurality of antennas is such that the power to be output by each antenna is the same and is equal to 1/(KN) of the total power to be output for all of the K frequency sub-bands. This process may be embodied by instructions encoded on a medium or in a communication device.
The above description is intended by way of example only.
Notice: More than one reissue application has been filed for the reissue of U.S. Pat. No. 7,881,674. The reissue applications are reissue application Ser. No. 13/755,945 which reissued as U.S. Pat. No. RE45,425 on Mar. 17, 2015, reissue application Ser. No. 14/563,231 which reissued as U.S. Pat. No. RE46,750 on Mar. 6, 2018, and the present application. This application is a continuation of U.S. application Ser. No. 10/800,610, filed Mar. 15, 2004, which is a continuation of U.S. application Ser. No. 10/174,689, filed Jun. 19, 2002, pending, which in turn claims priority to U.S. Provisional Application No. 60/361,055, filed Mar. 1, 2002, to U.S. Provisional Application No. 60/365,797 filed Mar. 21, 2002, and to U.S. Provisional Application No. 60/380,139, filed May 6, 2002. The entirety of each of the aforementioned applications are incorporated herein by reference. This application is a continuation reissue of U.S. patent application Ser. No. 14/563,231 filed Dec. 8, 2014, which reissued as U.S. Pat. No. RE46,750 on Mar. 6, 2018, which is a continuation reissue of U.S. patent application Ser. No. 13/755,945 filed Jan. 31, 2013, which reissued as U.S. Pat. No. RE45,425 on Mar. 17, 2015, which is a reissue application of U.S. patent application Ser. No. 11/879,156 filed Jul. 16, 2007, which issued as U.S. Pat. No. 7,881,674 on Feb. 1, 2011, which is a continuation of U.S. patent application Ser. No. 10/800,610 filed Mar. 15, 2004, which issued as U.S. Pat. No. 7,245,881 on Jul. 17, 2007, which is a continuation of U.S. patent application Ser. No. 10/174,689 filed Jun. 19, 2002, which issued as U.S. Pat. No. 6,785,520 on Aug. 31, 2004, which claims the benefit of U.S. Provisional Application Ser. No. 60/361,055 filed Mar. 1, 2002, U.S. Provisional Application Ser. No. 60/365,797 filed Mar. 21, 2002, and U.S. Provisional Application Ser. No. 60/380,139 filed May 6, 2002, the contents of which are hereby incorporated by reference herein.
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