BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a perspective view of a multiple-bay antenna according to one embodiment of the invention.
FIG. 2 is a calculated chart plotting signal strength versus azimuth for a prior art antenna.
FIG. 3 is a perspective view of a prior art panel antenna.
FIG. 4 is a calculated chart plotting signal strength versus azimuth for the panel antenna shown in FIG. 3.
FIG. 5 is a partial side view of a prior art antenna.
FIG. 6 is a partial side view of an antenna according to one embodiment of the invention.
FIG. 7 is a partial section view of the antenna of FIG. 1.
FIG. 8 is a calculated chart plotting signal strength versus azimuth according to one embodiment of the invention.
FIG. 9 is a measured chart plotting signal strength versus azimuth according to one embodiment of the invention.
FIG. 10 is a section view of an antenna according to one embodiment of the invention.
FIG. 11 is a section view of an antenna according to one embodiment of the invention.
FIG. 12 is a partial side view of an antenna according to one embodiment of the invention.
FIG. 13 is a section view of an antenna according to one embodiment of the invention.
DETAILED DESCRIPTION OF THE INVENTION
The invention will now be described with reference to the drawing figures, in which like reference numerals refer to like parts throughout. The invention provides an apparatus and method that in some embodiments provides an antenna that supports a substantially single-axis, null-free, vertically-polarized propagation pattern with high gain and moderate power handling capability.
FIG. 1 shows a multiple-bay antenna 10 according to one embodiment of the invention. The antenna 10 uses a self-supporting, vertical, straight coaxial line (coax) 12 to couple a signal between a feed line and multiple radiating elements in the form of a traveling wave. At each bay 14, a first dipole 16 and a second dipole 18 are positioned on opposite sides of the coax 12, with the second dipole 18 inverted with respect to the first dipole 16, so that the polarities of the radiated signals are opposite. It is noted that the cylindrical and effectively grounded outer conductor of the coax 12 is interposed between the dipoles 16, 18, and substantially blocks direct propagation from each toward the other. Similarly, also at each bay 14, a third dipole 20 and a fourth dipole 22 are positioned on opposite sides of the coax 12, and are oppositely oriented to each other, thus likewise radiating signals of opposite polarity. The signals from the second dipole pair 20, 22 are delayed with respect to signals from the first dipole pair 16, 18 by a prescribed portion of a wavelength proportional to the displacement of the tap locations of the second dipole pair 20, 22 with respect to the first dipole pair 16, 18 along the coax 12, discussed in greater detail below. The longitudinal axes of the dipoles 16, 18, 20, and 22 in all bays 14 lie in the half-planes 26, 28, 30, and 32 in at least some embodiments of the invention.
Vertical spacing of bays 14 as shown in FIG. 1 (i.e., distance along the longitudinal axis of the antenna 10 from a prescribe point on a first bay 14 to the corresponding point on a second bay 14) is approximately one wavelength per bay 14. This dimension can be adjusted over a comparatively broad range to achieve both beam tilt (by shortening the distance slightly from one wavelength) and null fill (by adjusting spacing to be slightly nonuniform, which broadens primary and secondary beams).
For convenience, the orientation axis 24 next to the antenna 10 in FIG. 1 shows four half-planes 26, 28, 30, and 32, each bounded by the orientation axis 24. The axis 24 represents the longitudinal axis of the coax 12, which is the vertical axis of the antenna 10. The longitudinal axes of the radial and vertical components of the conductors making up the respective dipoles 16, 18, 20, and 22 in all bays 14 lie in the half-planes 26, 28, 30, and 32, respectively.
Far-field signals from phase rotation-type antennas such as those of the '644 application are represented in the signal strength chart 34 of FIG. 2. FIG. 2 depicts the radiation pattern 34 with respect to azimuth of an antenna according to the '644 application, showing substantially omnidirectional emission. As noted, this is undesirable for some applications, if a substantial portion of the emitted energy propagates in unwanted directions. It may be observed that the approximate fourfold radial symmetry of this pattern 34 produces two pairs of principal lobes 60 and 62, respectively, one pair at roughly 40 degrees and 210 degrees, the other pair at roughly 130 degrees and 310 degrees. The nulls 64 have signal strength reduced by about 2.4 dB (20 log(0.76)) compared to the lobes 60, 62.
Signals from phase rotation-type antennas are substantially indistinguishable those emitted from antennas that emit signals simultaneously from a plurality of elements in each bay. An example of the latter antenna is a panel antenna 36, as represented in FIG. 3.
FIG. 3 illustrates a portion of a known panel antenna 36, wherein the antenna 36 has at least two radome-covered radiators 42 forming a single bay 52 as shown. The radiators 42 each radiate away from their common mounting 54 along the axis 56 shown, with propagation pattern and peak signal strength that depend on the details of radiator 42 design. A tower carrying multiple bays 52 stacked vertically can realize higher gain (less elevation spread) than a single bay 52 by using proper power splitting and phase synchronization.
FIG. 4 shows a representative signal strength chart 38 for known panel antennas; each of the two lobes 40 is emitted from one panel radiator 42 as shown in FIG. 3. FIG. 4 depicts the radiation pattern 38 with respect to azimuth of an antenna 36 with at least one bay using a back-to-back pair of panel antennas, such as those shown in FIG. 3. Between the lobes 40 are deep nulls 58, which require fill by some method to avoid loss of function, such as loss of signal for a mobile television as it passes near the antenna.
As shown in FIG. 5, the structure of antennas 44 according to the '644 application permits signal radiation with rotating phase and a high degree of azimuthal uniformity. The first two dipoles 46 in each bay are inverted with respect to each other, which causes them to radiate with opposite phase when excited from a common source, which would originate beyond the rightmost extent of FIG. 5. The second two dipoles 48 (the fourth dipole is located behind the coax) in each bay are located a distance 50 (L) that is a quarter-wavelength (90 degrees) further from the signal source than the first two dipoles 46. The second two dipoles 48 are likewise relatively inverted. As a result, the signal applied to the four dipoles 46, 48 radiates with successive 90 degree shifts around the coax 12, shown in FIG. 1. This phase rotation is repeated substantially synchronously (delayed approximately one cycle per bay by the traveling wave feed) at each bay 14. Provided that corresponding dipoles 46, 48 in all bays 14 are substantially aligned, the signals from all bays 14 reinforce to provide gain, and the output is substantially omnidirectional.
FIG. 6, by contrast with FIG. 5, shows a partial side view of an antenna 66 according to an embodiment of the invention, wherein the spacing between the first dipole pair 68 and the second dipole pair 70 (the fourth dipole is located behind the coax) is changed from the quarter-wavelength distance L 50 (90 degrees) of an antenna 44 according to the '644 application, shown in FIG. 5, to a one-third wavelength distance L 72 (120 degrees) in an antenna 66 according to an embodiment of the invention. This change within the spacing of the first and second dipole pairs 46 and 48 versus 68 and 70, respectively, shifts the emission in time and alters phase progression, so that intermediate azimuth angles no longer manifest substantially full-power signals with intermediate phase angles. Instead, the signals from the dipoles closest in phase reinforce at some intermediate angles and cancel at others. As a result, the primary lobes 74 (shown in FIG. 8, further addressed below) are skewed somewhat, with peak signal strength driven in part by phase-proximal dipoles, while the secondary lobes 76 are minor artifacts associated with signal reinforcement and cancellation, and the nulls 78 are associated with strong cancellation between substantially out-of-phase radiators.
The respective distances L in FIGS. 5 and 6, identified by respective reference numerals 50 and 72, are prescribed to establish the relative phase of the two sets of dipoles in each bay. As this distance L is changed, the overall radiation pattern with azimuth changes. The value of L shown in FIG. 6, which is increased by about a third, to 120 degrees or a third of a wavelength, compared to such embodiments of the '644 application as the one shown in FIG. 5, is desirable at least for a class of transmitting antennas for highway mobile broadcast applications in the indicated frequency range. For other applications and frequency ranges, adjustment of the prescribed phase spacing L, such as to a value different from a third of a wavelength, can be used to balance spacing of transmitter towers, power per transmitter, expected mobile radio sensitivity, signal leakage and intrusion from beyond the intended zone, and other considerations. As with other multiple-bay antennas, the number of bays per antenna, the spacing between bays, and the amount of signal applied to each bay can be prescribed in consideration of tower height, main beam width, beam tilt, null fill, and other factors.
Analysis and test demonstrate that the dipoles 16, 18, 20, and 22, shown in FIG. 1 and making up each bay 14 in antennas 10 according to some embodiments of the invention, exhibit substantial impedance cancellation, so that the input impedance of the antenna 10 is substantially independent of the number of bays in the antenna. Power level to each bay 14, and to some extent to each dipole 16, 18, 20, and 22 within a bay 14, can be varied by selecting dielectric thickness to establish a preferred extent of coupling from the coaxial line to each dipole, with manageable effect on overall impedance.
It is to be understood that an extent of impedance cancellation can be made substantially complete by a combination of equal coupling of dipoles 16, 18, 20, and 22 in a given bay 14 and quarter-wavelength spacing L between the longitudinally displaced dipole couplings within a given bay 14, as described in the '644 application. Variations from this equal-coupling, quarter-wavelength-spacing configuration tend to narrow antenna bandwidth and to increase the extent to which transmission line loading by the radiative elements appears as successive lump impedances across the characteristic transmission line impedance of the coax 12. Each such variation may manifest as resistance plus capacitive or inductive reactance, in series and/or parallel, with the spacing and coupling variation correlated to an extent of phase alteration. The plurality of possible variations, along with differences in the rate of change of emission pattern and line loading with dimension variation, permit an antenna according to the '644 application to be adapted according to the invention disclosed herein to provide non-omnidirectional propagation over a broad range of patterns. Techniques used may include longitudinal and radial shifting of the locations of corresponding dipoles in each bay 14 and adjusting coupling, as disclosed herein.
FIG. 7 shows a section view of an antenna 80 according to the invention. The dipoles 82 and 84 are seen to have respective hot elements 86 and 88 fed from the inner conductor 90 using insulating pads 92 that function as the dielectric of capacitors formed between the hot elements 86 and 88 and the inner conductor 90. The respective cold elements 94 and 96 of the dipoles 82 and 84 are pressed into interference-fit holes 98 in the outer conductor 100, forming electrical joints, in the embodiment shown. It is implicit in referring to the elements as hot and cold that the feed line to the antenna and the traveling wave coax of the antenna itself may provide kilowatt-level RF excitation to their respective inner conductors while keeping their outer conductors at substantially ground potential, for safety, reliability, and such considerations as lightning protection of the transmitter. While not mandatory, this assumption drives at least the hot and cold reference terminology. Other physical arrangements are possible, and the arrangement indicated should not be viewed as limiting.
The component dimensions of the dielectric pads 92 may be substantially uniform in some embodiments. In those embodiments, if the coupling capacitances are roughly equal for all dipoles, the remaining signal level in the center coaxial conductor decreases by logarithmic steps with successive bays, and, as a consequence, successive dipoles tend to couple decreasing amounts of power from the center coax. While desirable in many embodiments, and well known in the art for traveling wave antennas, this can be changed by adjusting coupling in successive bays (thickness of the pads 92) according to a chosen sequence. For example, thickness can be decreased as a function of position (such as the logarithm) in successive bays to yield substantially uniform emission from each bay. Alternatively, in order to increase bay power at the center of the aperture, for example, pad 92 thickness can decrease faster than the above function calls for from bottom to middle of the antenna 10, with uniform pad 92 thickness applied from middle to top. Any comparable strategy, including uniform pad 92 dimensions and log taper of power per bay, may provide a desirable combination of producibility and performance in some embodiments.
Termination of the antenna can be realized with a terminal short-circuit spaced a quarter-wavelength from the bay distal to the feed port; in some embodiments this can cause the termination to reflect as an open. In keeping with this, the dipoles of the distal bay may have thinner pads 92 to increase capacitive coupling and minimize the signal remaining to reach the terminal short-circuit. Various other termination strategies are known in the art for traveling wave antennas; in many embodiments, it is possible to provide at least a substantially nonreactive termination, with a minimally dissipative termination preferred in order to maximize radiated power and minimize losses and reflections.
As noted above, the separation dimension 102 (D) in FIG. 7 is nominally one-third wavelength for the embodiment shown. The first two hot elements 86 and 88, respectively, in each bay 104 are spaced one-third wavelength away 102 from the second two elements, of which one, 106, is shown dashed, and the other is not visible in this section view. Ninety degrees of phase shift for the latter elements can provide phase rotation and impedance cancellation. Complete impedance cancellation would prevent the multiple parallel loads of the bays from lowering antenna input impedance, so that no compensating input transformer would be needed. An input transformer may be appropriate in antennas according to the invention, but such a transformer may require a lesser transformation ratio than transformers in designs lacking impedance cancellation, and the phenomenon of narrowing the working bandwidth of the antenna due to the need for a high number of impedance steps—the coaxial-line equivalent of a coil-based transformer's turns ratio—may be diminished.
As employed in the invention disclosed herein, the one-third wavelength separation dimension 102 (D), i.e., 120 degrees rather than 90 degrees, affects impedance cancellation somewhat and strongly affects lobe balance and lobe skew. Because impedance cancellation changes only slowly with separation 102, while lobe balance and lobe skew vary relatively rapidly, varying separation 102 to affect lobe balance and lobe skew is a useful mechanism for producing antennas that vary widely in lobe shape, orientation, balance, and skew. As a corollary, it may be seen that the dimension 102 (D) is relatively critical in establishing a particular lobe shape, orientation, balance, and skew in at least some embodiments, although it can be obviated or combined with alternative methods of realization in other embodiments. This is shown further in the figures discussed below.
FIG. 8 depicts the radiation pattern 108 with respect to azimuth of an antenna according to an embodiment of the invention. Primary lobes 74 are substantially oriented as corresponding lobes in known antennas, while secondary lobes 76, skewed to lie in the vicinity of 110 degrees and 290 degrees, may be seen from the chart of FIG. 8 to be attenuated by about 6 dB (peak voltage value is about 50%; 20 log(0.5)=−6) and to show no appreciable nulls between the skewed secondary lobes 76 and the proximal primary lobes 74. Calculated signal strength in the nulls 78, located at approximately 160 degrees and 340 degrees, is about −8 dB (20 log(0.4)=−8) referred to the primary lobe 74 peaks, significantly higher than the vanishingly-small signal (below −10 dB over two 30-degree arcs) in the nulls 58 of the panel antenna configuration of FIG. 4.
FIG. 9 is a measured radiation pattern 110 of an antenna according to an embodiment of the invention. It may be readily observed that primary lobes 112, secondary lobes 114, and nulls 116 correspond closely to those of the analytical model of FIG. 8, other than orientation. The −7 dB (average) skewed secondary lobes 114 fall at approximately 130 degrees and 310 degrees, and the −8.4 dB and −10.5 dB nulls 116 are at approximately 105 degrees and 280 degrees, respectively. The invention may accept further alteration, such as for further narrowing or widening the primary lobes 112, reducing null 116 depth in exchange for increasing secondary lobe 114 magnitude or skew, and the like.
Substantial beam tilt can be established by adjusting the spacing between bays, with a bottom-feed antenna requiring decrease in spacing to depress the main beam below the horizon, and with the opposite case remaining valid—that is, the beam of a bottom-fed antenna can be directed upward by increasing interbay spacing, while a top-fed antenna requires increased interbay spacing for downward direction of the beam, and decreased interbay spacing to direct the beam upward. Null fill can be realized by providing interbay spacing that changes from bay to bay, with the variation determining the extent of null fill over a significant range.
It is to be understood that the software model and prototype test results of FIG. 9 refer to an embodiment wherein the second dipole pair has been shifted about a sixth of a wavelength from the omnidirectional configuration of the '644 application. Any shift greater or less than this amount over a significant range will likewise produce a potentially acceptable, albeit likely different, combination of impedance cancellation and alteration of lobe shape, orientation, balance, and skew. Dipole shifts in the negative direction—that is, shifting the second pair closer to the first pair than one-quarter wavelength rather than further away as in the embodiment shown—will likewise produce an effect comparable to that described, but with lobe alteration differing in detail, and with the characteristics of impedance cancellation differing as well.
FIG. 10 is a cross-sectional view applying equally to antennas 44 and 66 of FIGS. 5 and 6, respectively. By contrast, FIG. 11 is a cross-sectional view of the antenna 120 shown in part in FIG. 12, wherein dipole physical azimuth angle is used in place of traveling wave phase angle to achieve a broadly equivalent non-omnidirectional propagation pattern. The arrangement of FIG. 12 may introduce significant tradeoffs when used with certain fabrication apparatus. It can be beneficial, in view of the general desirability of increasing automation and decreasing setup, for holes and insert fittings to be parallel to or at right angles to each other, as in the elements 122 of FIG. 10. Thus, an antenna built as in FIGS. 11 and 12, with respective dipoles 124 not at right angles, may be more costly to position, drill, and assemble, although such an arrangement may be advantageous for some embodiments.
Similarly, rotation of physical azimuth angle, as in FIGS. 11 and 12, and traveling wave phase angle, as in FIG. 6, may be combined in some embodiments. For example, pronounced beam narrowing may be combined with reduced degradation of impedance cancellation by incorporating both processes to a greater or lesser extent, with the final configuration determined by analysis of computer modeling and prototype testing.
FIG. 13 illustrates an antenna wherein the overall structure 126 is asymmetrical—that is, while the dipoles 124 in the section in FIG. 11 lie in two planes, embodiments such as the antenna 26 of FIG. 13 place the dipoles 128 in four half-planes without excessively degrading impedance cancellation. Signal propagation for such embodiments may be asymmetrical, which can provide capabilities that symmetrical arrangements cannot achieve.
Likewise, longitudinal placement of dipoles with respect to the antenna feed port may be nonsymmetrical in some embodiments. As long as the four dipoles at each bay approximate the equal loading achieved with separation by one-quarter wavelength, impedance cancellation is preserved to at least some extent. Thus, each two dipoles may be above and below a nominal tap point, with a predictably asymmetrical propagation pattern, but without unacceptable degradation of loading.
Although elements in successive bays are suggested by the figures to have uniform spacing in successive bays, so that the beams produced have gain over azimuth that is a function of the number of bays, it is also possible to adjust the element arrangement and thus the beam shape of each bay independently of the other bays, so that the overall antenna nulls and secondary lobes are tailored to a desired profile. Such variations will generally widen the beam and reduce the effective gain by increasing signal cancellation, but may be used in lieu of omnidirectional radiators at freeway interchanges, for example. Development of individual antennas with tailored beam shape will in typical embodiments require recourse to antenna design software prior to fabrication of hardware, and validation by test afterward. Since this potentially adds to development, fabrication apparatus programming, touch labor, and testing costs, it is foreseeable that standard designs such as those of FIGS. 6 and 12 may be preferred for many applications.
The many features and advantages of the invention are apparent from the detailed specification, and, thus, it is intended by the appended claims to cover all such features and advantages of the invention which fall within the true spirit and scope of the invention. Further, since numerous modifications and variations will readily occur to those skilled in the art, it is not desired to limit the invention to the exact construction and operation illustrated and described, and, accordingly, all suitable modifications and equivalents may be resorted to that fall within the scope of the invention.
Patent Application
20080079647
