Patent Grant
8923436
This application is a National Stage Entry of PCT/JP2011/002452 filed Apr. 26, 2011, which claims priority from Japanese Patent Application 2010-174453 filed Aug. 3, 2010, the contents of all of which are incorporated herein by reference, in their entirety.
The present invention relates to a transmitter used for a communication device, such as a cellular phone or a wireless LAN, and a method for controlling the same, and more particularly, to a transmitter having an excellent power efficiency, and a method for controlling the same.
A transmission unit of a communication device, such as a cellular phone or a wireless LAN, is required to operate with low power consumption. Such a transmission unit of a communication device is required to operate with low power consumption regardless of the magnitude of output power, and is also required to ensure a high accuracy of transmission signals. In particular, a power amplifier disposed at a final stage of a transmission unit of a communication device occupies 50% or more of the power consumption of the entire transmitter. For this reason, the power amplifier disposed at the final stage of the transmission unit of the communication device needs to have a high power efficiency.
In recent years, a switching amplifier has been attracting attention as a power amplifier that is expected to achieve a high power efficiency. The switching amplifier receives pulse-shape signals. The switching amplifier amplifies the power of received pulse-shape signals while maintaining the pulse shape. The pulse-shape signals amplified by the switching amplifier are transmitted from an antenna after frequency components other than desired frequency components are sufficiently suppressed by a filter element.
A current mode class-D amplifier (hereinafter referred to as “CMCD”), which is a typical switching amplifier, will now be described.
Pulse signals are input to control terminals of the switching elements 63 and 64. The pulse signal input to the control terminal of the switching element 64 is a signal complementary to the pulse signal input to the control terminal of the switching element 63. Thus, when one of the switching elements 63 and 64 is turned on, the other of the switching elements is controlled to be turned off. The current source connected to the switching element in the OFF state supplies a current to the load 65 and the filter circuit 66. A current from the current source connected to the switching element flows into the switching element in the ON state. Further, a current from the current source connected to the switching element in the OFF state flows into the switching element in the ON state through the load 65 and the filter circuit 66.
A typical class-AB power amplifier and the like require a bias current. On the other hand, the CMCD requires no bias current. Accordingly, a power loss in the CMCD is equal to the sum of a switching loss generated during charging/discharging to/from a parasitic capacitance and a heat loss generated in a parasitic resistance. Accordingly, when the parasitic capacitance and the parasitic resistance are ideally zero, the power loss of the CMCD is “0”.
Next, a configuration example of a transmitter 700 incorporating the CMCD 6 will be described.
The RF signal generator 71 includes a digital baseband (hereinafter referred to as “DBB”) 711, sigma-delta modulators 712 and 713, a digital up-converter 714, and an inverter 715. In the case of W-CDMA, for example, the DBB 711 generates a radio signal which is a multi-bit signal of 10 bits or more. On the other hand, a pulse signal representing information by two states (1 bit) of high and low can be input to the CMCD 6. Accordingly, it is necessary to convert the multi-bit signal output from the DBB 711 into a 1-bit signal that is subjected to over-sampling in advance. The sigma-delta modulators 712 and 713 are used as means for converting a multi-bit signal into a 1-bit signal by over-sampling.
Each signal output from the sigma-delta modulators 712 and 713 is output as a pulse signal through the digital up-converter 714. The pulse signal output from the digital up-converter 714 is divided into two signals. One of the divided pulse signals is inverted by the inverter 715. A non-inverted pulse signal is input to the switching element 63 of the CMCD 6 through the driver amplifier 72. The inverted pulse signal is input to the switching element 64 of the CMCD 6 through the driver amplifier 72.
The sigma-delta modulators 712 and 713 described above can favorably maintain noise characteristics in the vicinity of a desired frequency band. Accordingly, in this configuration example, a multi-bit radio signal can be converted into a pulse signal and the pulse signal can be input to the CMCD 6, while maintaining satisfactory noise characteristics.
However, the inventor has found that highly efficient power amplification cannot be achieved in practice in the above-mentioned circuit configuration example.
Here, consideration is given to the case where each of the switching elements 63 and 64 is composed of an FET element. In this case, a voltage at the drain terminal of the FET element fluctuates to the minus side. The FET element is turned off when the voltage at the drain terminal is equal to or higher than 0 and a gate potential is smaller than a threshold (Vth) with respect to a source potential. However, when the voltage at the drain terminal fluctuates to the minus side and the difference between the gate potential and the drain potential is equal to or greater than the threshold (Vth), the channel is opened and comes into a conductive state.
Therefore, to maintain the OFF state even when the voltage at the drain terminal fluctuates to the minus side in the FET element, it is necessary to cause the gate potential to follow the drain potential and to fluctuate to the minus side. In other words, it is necessary to constantly set the difference between the gate potential and the drain potential to be equal to or smaller than Vth. In general, when a W-CDMA signal of 100 W class is output, the potential at the drain terminal fluctuates to the minus side by about several tens of volts. Accordingly, it is necessary to cause the voltage of the OFF signal, which is applied to the gate of the switching element, to fluctuate to the minus side by about several tens of volts. On the other hand, it is necessary to cause the ON signal to fluctuate to the plus side by several volts. Accordingly, the potential difference between the ON signal and the OFF signal to be applied to the gate of the switching element is about several tens of volts.
A power Pdrv necessary to drive the gate of the switching element is expressed by the following expression (1). Hereinafter, Cgs represents a gate capacitance; Vgs represents a potential difference between the ON signal and the OFF signal to be applied to the gate; and f represents a switching frequency.
Pdrv=Cgs·Vgs2·f (1)
As shown in the expression (1), the power Pdrv is proportional to the square of the potential difference Vgs. In a typical device, several volts are required as the potential difference Vgs. The power Pdrv at this time is about several watts. Accordingly, in the state where the potential difference Vgs is several tens of volts, the power Pdrv is several hundred watts, resulting in a considerable increase in power consumption.
The present invention has been made in view of the above-mentioned problems, and an object of the present invention is to provide a transmitter having favorable noise characteristics and excellent power efficiency, and a method for controlling the same.
A transmitter according to an exemplary aspect of the present invention includes: an RF signal generator that divides an input radio signal into an amplitude signal and a phase signal and outputs the amplitude signal and the phase signal; and a switching amplifier that amplifies the radio signal with the amplitude signal and the phase signal. The switching amplifier includes: at least one variable current source that is controlled by the amplitude signal and supplies a current to the switching amplifier; and at least one switching element that connects the variable current source to one of a ground potential and an output terminal of the switching amplifier according to the phase signal.
A method for controlling a transmitter according to another exemplary aspect of the present invention includes: dividing an input radio signal into an amplitude signal and a phase signal and outputting the amplitude signal and the phase signal; amplifying the radio signal with the amplitude signal and the phase signal; and connecting a variable current source controlled by the amplitude signal to one of a ground potential and an output terminal according to the phase signal.
According to the present invention, it is possible to provide a transmitter having favorable noise characteristics and excellent power efficiency, and a method for controlling the same.
Exemplary embodiments of the present invention will be described below with reference to the drawings. In the drawings, the same elements are denoted by the same reference numerals, and repeated explanation is omitted as needed. Hereinafter, assume that a current mode class-D amplifier (hereinafter referred to as “CMCD”) is one mode of a switching amplifier.
First, a transmitter according to a first exemplary embodiment of the present invention will be described.
The RF signal generator 1a includes a digital baseband (hereinafter referred to as “DBB”) 11, an IQ modulator 12, and an amplitude phase signal detector 13. The DBB 11 generates IQ signals. The IQ modulator 12 converts the IQ signals into an RF signal RF(t) by IQ modulation. Here, “t” is a variable representing time. The RF signal RF(t) is generally expressed by the following expression (2) using an amplitude signal r(t) and a phase signal th(t).
RF(t)=r(t)·th(t) (2)
The phase signal th(t) is expressed by the following expression (3).
th(t)=cos(ωc·t+theta(t)) (3)
where ωc is an angular velocity obtained by multiplying a carrier frequency by 2π, and theta(t) is a phase variation.
The amplitude phase signal detector 13 separates and extracts the amplitude signal r(t) and the phase signal th(t) which are included in the RF signal RF(t). The amplitude phase signal detector 13 outputs the amplitude signal r(t) and a rectangular phase signal Rth(t) which is obtained by forming the phase signal th(t) into a rectangular shape. When the phase signal th(t) is equal to or greater than 0, the value of the rectangular phase signal Rth(t) is “1”. When the phase signal th(t) is smaller than 0, the value of the rectangular phase signal Rth(t) is “0”. A relationship as expressed by the following expression (4) is established between the phase signal th(t) and the rectangular phase signal Rth(t).
Rth(t)=th(t)+dis(t) (4)
where dis(t) is composed of a harmonic component of the(t).
The CMCD 2a includes variable current sources 21 and 22, switching elements 23 and 24, a load 25, and a filter circuit 26. The variable current sources 21 and 22 are connected in parallel with a power supply VDD. The switching element 23 is connected between the variable current source 21 and a ground GND. The switching element 24 is connected between the variable current source 22 and the ground GND. The load 25 and the filter circuit 26 are connected in parallel between an output terminal of the switching element 23 and an output terminal of the switching element 24. The amplitude signal r(t) is input to the variable current sources 21 and 22 as a control signal. The rectangular phase signal Rth(t) is input to the switching element 23 as a control signal. A complementary signal of the rectangular phase signal Rth(t) is input to the switching element 24 as a control signal through an inverter INV.
A flow of a current in the CMCD 2a will now be described. First, a description is given of the case where the switching element 23 is turned off and the switching element 24 is turned on. In this case, a current flows into the switching element 24 from the variable current source 21 through the load 25 and the filter circuit 26. Further, a current flows into the switching element 24 from the variable current source 22. Subsequently, a description is given of the case where the switching element 23 is turned on and the switching element 24 is turned off. In this case, a current flows into the switching element 23 from the variable current source 22 through the load 25 and the filter circuit 26. Further, a current flows into the switching element 23 from the variable current source 21. That is, the pulse-like current, the flow direction of which is reversed so as to correspond to opening and closing operations of the switching elements 23 and 24, is input to the load 25 and the filter circuit 26.
Next, a configuration example of the filter circuit 26 will be described.
Suitable setting of a constant of each element that forms the filter circuit 26 allows the resonance point of the filter circuit 26 to match a desired bandwidth of a radio signal. As a result, a state close to the open condition is achieved for the desired bandwidth. On the other hand, a state close to the short-circuit condition is set for bandwidths other than the desired bandwidth. Accordingly, only the desired band component of the pulse current is supplied through the load 25. As a result, the radio signal from which undesired harmonic components are removed is extracted.
Assume herein that a current value output from the variable current sources 21 and 22 is represented by I(t). A control signal given to each switching element is represented by P(t). The control signal P(t) is a 1-bit signal having two states of “1” and “−1”. When the control signal P(t) is “1”, the switching element 23 is turned on. When the control signal P(t) is “−1”, the switching element 24 is turned on. Accordingly, a current signal Dout(t) output to each of the load 25 and filter circuit 26 is expressed by the following expression (5).
Dout(t)=I(t)·P(t) (5)
Further, the current signal Dout(t) is expressed by the following expression (6).
The resonance frequency of the filter circuit 26 is set to correspond to r(t)·th(t) that is included in Dout(t). This allows only r(t)·th(t) to be input to the load 25. In other words, only the RF signal can be extracted by the CMCD 2a.
A configuration example of each unit of the transmitter 100 will be described below. First, a configuration example of the IQ modulator 12 will be described.
The IQ local oscillator 121 generates two sinusoidal wave voltage signals having a phase difference of 90° therebetween. These two sinusoidal wave voltage signals have a frequency equal to the carrier frequency of the RF signal. The mixer 122 and the mixer 123 output the product of the signals input from two input terminals. The mixer 122 receives an I signal from the DBB 11 and a sinusoidal wave voltage signal generated in the IQ local oscillator 121. The mixer 123 receives a Q signal from the DBB 11 and a sinusoidal wave voltage signal generated in the local oscillator. The synthesizer 124 outputs the sum of the signals input from the two input terminals.
Assume that one of the sinusoidal wave voltage signals generated in the IQ local oscillator 121 is represented by vlo_i, and the other of the sinusoidal wave voltage signals is represented by vlo_q. The voltage signal vlo_i is input to the mixer 122. The voltage signal vlo_q is input to the mixer 123. There is a phase difference of 90 degrees between the voltage signal vlo_i and the voltage signal vlo_q. Accordingly, the voltage signal vlo_i and the voltage signal vlo_q are expressed by the following expressions (7) and (8), respectively.
vlo—i=A cos(ωct) (7)
vlo—q=A sin(ωct) (8)
where ωc is an angular frequency corresponding to the carrier frequency.
Assume herein that a voltage of a baseband signal I which is input to the mixer 122 is represented by Vbb_i and a voltage of a baseband signal Q which is input to the mixer 123 is represented by Vbb_q. The baseband signals Vbb_i and Vbb_q are expressed by the following expressions (9) and (10), respectively.
Vbb—i=B cos(ωbt+θ) (9)
Vbb—q=−B sin(ωbt+θ) (10)
where B represents amplitude information; θ represents phase information; and ωb represents an angular frequency corresponding to an intermediate frequency.
The mixer 122 outputs the product of a voltage signal Vlo_i and the voltage Vbb_i of the baseband signal I. The mixer 123 outputs the product of a voltage signal Vlo_q and the voltage Vbb_q of the baseband signal Q. An output voltage signal Vmix1 of the mixer 122 is expressed by the following expression (11). An output voltage signal Vmix2 of the mixer 123 is expressed by the following expression (12).
Vmix1=0.5×AB{cos((ωc+ωb)t+θ)+cos((ωc−ωb)t−θ)} (11)
Vmix2=0.5×AB{cos((ωc+ωb)t+θ)−cos((ωc−ωb)t−θ)} (12)
The synthesizer 124 outputs the sum of the output voltage signal Vmix1 of the mixer 122 and the output voltage signal Vmix2 of the mixer 123. An output voltage signal Vcomb of the synthesizer 124 is expressed by the following expression (13).
Vcomb=AB cos((ωc+ωb)t+θ) (13)
The output voltage signal Vcomb is an RF signal obtained by increasing the frequency by ωc from the angular frequency of the baseband signal.
Next, the amplitude phase signal detector 13 will be described.
Next, the switching elements 23 and 24 will be described.
Next, the variable current sources 21 and 22 will be described.
The amplitude signal r(t) generated in the RF signal generator 1a is input to the decoder 211. When the amplitude signal r(t) generated in the RF signal generator 1a is an N-bit digital signal, the pulse variable current source 212 is provided with the DC current sources Iv1 to IvN. A current value In of the DC current source Ivn is weighted by a power of 2. Specifically, the current value In is represented by I0×2−n. Here, I0 represents a given value. The decoder 211 sequentially allocates the bits of the amplitude signal r(t) from the highest bit as control signals for the current switches S1 to SN respectively connected to the DC current sources Iv1 to IvN. When the amplitude signal r(t) is an analog signal, this analog signal is converted into an N-bit digital signal by AD conversion. The digital signal generated by AD conversion is input to the decoder 211.
Subsequently, the current switch Sn will be described.
When the current switching element SW is in the opened state, a current input to the terminal A is output to the terminal B through the diode Ds. When the current switching element SW is in the short-circuit state, the current input to the terminal A is output to the ground potential through the current switching element SW.
Subsequently, the DC current source Ivn will be described.
Assume herein that the voltage value of the variable DC voltage source 41 is represented by Vdc. The inductance of the inductor 42 is represented by Ladd. The value of a load resistance connected to the output terminal of the DC current source Ivn is represented by Rload. The output current of the DC current source Ivn is represented by Tout. Assuming that the voltage value Vdc represents an input signal and Tout represents an output signal, a transfer function F(s) is expressed by the following expression (14).
F(s)=Rload/(s·Ladd+Rload) (14)
The above expression shows that the transfer function F(s) is a transfer function of a lowpass filter in which a 3 dB cutoff frequency is given by Rload/(2·π·Ladd). The control signal given to the variable DC voltage source 41 is updated at a speed sufficiently higher than the above-mentioned cutoff frequency, thereby minimizing the variation characteristic of the output current Tout.
The configuration of the DC current source Ivn will be described in more detail.
The current sensor 43 includes a resistor Rs and a differential input type amplifier AMP. The differential input type amplifier AMP receives voltage information at both terminal nodes of the resistor Rs. When the input current flows through the resistor Rs, a voltage difference which is equal to the product of the current and the resistance value is generated at the both ends of the resistor Rs. The differential input type amplifier AMP amplifies and outputs the voltage difference (in this exemplary embodiment, the amplification factor is a positive value). Note that the voltage difference between the both ends of the resistor Rs and the output value of the differential input type amplifier AMP have a 1:1 relation. That is, the current sensor 43 is capable of converting the current value of the input current into a voltage value and outputting the voltage value.
The comparison controller 44 includes a voltage comparator 52 and a coder 53. The voltage comparator 52 compares the voltage value output from the current sensor 43 with an internal reference value. The case where the output from the current sensor 43 is larger than the internal reference value indicates that the amount of current flowing through the current sensor 43 is larger than a desired value. In this case, the coder 53 outputs a control signal that brings the ground-side switching element Sv2, which forms the variable DC voltage source 41, into the ON state, and brings the power-supply-side switching element Sv1 into the OFF state. On the other hand, the case where the output from the current sensor 43 is smaller than the internal reference value indicates that the amount of current flowing through the current sensor 43 is smaller than the desired value. In this case, the coder 53 outputs a control signal which brings the ground-side switching element Sv2, which forms the variable DC voltage source 41, into the OFF state, and brings the power-supply-side switching element Sv1 into the ON state.
The comparison controller 44 operates in synchronization with an external clock signal from an external clock signal source CLKO. Specifically, the cycle of performing a comparison operation in the voltage comparator 52 and updating the control signal to be supplied to the variable DC voltage source 41 is equal to the cycle of the external clock signal source CLKO. Even when the load connected to the DC current source Ivn is temporally varied, the comparison controller is caused to operate using the clock signal source that generates a clock signal having a sufficiently shorter cycle than the variation cycle. This allows the control signal for the variable DC voltage source 41 to be updated in a period shorter than the variation cycle of the load. The operation described above allows the DC current source Ivn to continuously output a desired DC current.
Subsequently, configuration transformed examples of the CMCD will be described.
The inductors 32 and 33 suppress the variation of the current value. Accordingly, the inductors 32 and 33 provide a current source operation in a pseudo manner. Thus, the CMCD 2b according to this configuration example operates in a similar manner to the CMCD 2a shown in
A second configuration transformed example of the CMCD will be described.
Further, a third configuration transformed example of the CMCD will be described.
Next, a transmitter according to a second exemplary embodiment of the present invention will be described.
The RF signal generator 1b has a configuration in which a divider 14 is added to the RF signal generator 1a of the transmitter 100 and the amplitude detector 131 is replaced with an amplitude detector 133. In the transmitter 200, the DBB 11 outputs IQ signals to each of the divider 14 and the amplitude detector 133.
The divider 14 outputs signals obtained by dividing the IQ signals by the amplitude signal output from the amplitude detector 133. Radio signals Ib(t) and Qb(t) which are output from the divider 14 are expressed by the following expressions (15) and (16), respectively.
Ib(t)=I(t)/r(t) (15)
Qb(t)=Q(t)/r(t) (16)
The radio signals Ib(t) and Qb(t) output from the divider 14 are input to the IQ modulator 12 and converted into a radio signal of an RF band. The amplitude value of the output signal of the IQ modulator 12 is proportional to the amplitude value of the input IQ signals as shown in the expression (13). As shown in the expressions (15) and (16), Ib(t) and Qb(t) are specified by the amplitude signals of the IQ signals of the DBB 11. Accordingly, the amplitude value of the radio signal defined by Ib(t) and Qb(t) is “1”. That is, the amplitude value of the output signal of the IQ modulator 12 according to this exemplary embodiment is “1”. Therefore, an output signal RFb(t) of the IQ modulator 12 is expressed by the following expression (17) in which “1” is substituted in the amplitude signal r(t) on the right-hand side of the expression (2).
RFb(t)=th(t) (17)
That is, the output signal of the IQ modulator 12 is the phase signal itself. The phase detector 132 determines the positive or negative of the output signal of the IQ modulator 12, and generates the rectangular phase signal shown in the expression (4).
Subsequently, the amplitude detector 133 will be described.
As described above, the RF signal generator 1b outputs the output signal of the amplitude detector 133 as an amplitude signal. The RF signal generator 1b also outputs the output signal of the phase detector 132 as a phase signal. The amplitude signal and phase signal output from the RF signal generator 1b are input to the CMCD 2a, as in the transmitter 100 according to the first exemplary embodiment. Thus, the CMCD 2a operates in the same manner as in the first exemplary embodiment, and the RF signal is reproduced in the load connected to the CMCD 2a.
Note that the present invention is not limited to the above exemplary embodiments, but can be modified as needed without departing from the gist of the present invention. For example, the CMCD 2a of the transmitter 200 according to the second exemplary embodiment can be replaced with one of the CMCDs 2b to 2d, as needed, as in the transmitter 100 according to the first exemplary embodiment.
The exemplary embodiments described above illustrate transmitters including the current mode class-D amplifier which is one mode of the switching amplifier. However, the amplifier that is applied to the present invention is not limited thereto. That is, other switching amplifiers can be applied in place of the current mode class-D amplifier.
While the present invention has been described above with reference to exemplary embodiments, the present invention is not limited to the above exemplary embodiments. The configuration and details of the present invention can be modified in various manners which can be understood by those skilled in the art within the scope of the invention.
This application is based upon and claims the benefit of priority from Japanese patent application No. 2010-174453, filed on Aug. 3, 2010, the disclosure of which is incorporated herein in its entirety by reference.
The present invention can be applied to, for example, communication devices such as a cellular phone and a wireless LAN.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2010-174453 | Aug 2010 | JP | national |
| Filing Document | Filing Date | Country | Kind | 371c Date |
|---|---|---|---|---|
| PCT/JP2011/002452 | 4/26/2011 | WO | 00 | 2/4/2013 |
| Publishing Document | Publishing Date | Country | Kind |
|---|---|---|---|
| WO2012/017580 | 2/9/2012 | WO | A |
| Number | Name | Date | Kind |
|---|---|---|---|
| 6750719 | Toyota et al. | Jun 2004 | B2 |
| 8253487 | Hou et al. | Aug 2012 | B2 |
| 8620240 | Yamanouchi et al. | Dec 2013 | B2 |
| 8675725 | Staszewski et al. | Mar 2014 | B2 |
| 20050032488 | Pehlke et al. | Feb 2005 | A1 |
| 20050191976 | Shakeshaft et al. | Sep 2005 | A1 |
| 20050233714 | Kajiwara et al. | Oct 2005 | A1 |
| 20080074209 | Ceylan et al. | Mar 2008 | A1 |
| 20090163156 | Rofougaran et al. | Jun 2009 | A1 |
| 20100001793 | Van Zeijl et al. | Jan 2010 | A1 |
| 20100066429 | Ikedo et al. | Mar 2010 | A1 |
| 20120056676 | Frambach | Mar 2012 | A1 |
| 20120188024 | Yamanouchi | Jul 2012 | A1 |
| Number | Date | Country |
|---|---|---|
| H01254014 | Oct 1989 | JP |
| 2010141521 | Jun 2010 | JP |
| 2008093665 | Aug 2008 | WO |
| Entry |
|---|
| The international search report for PCT/JP2011/002452 mailed on Jun. 7, 2011. |
| R. Leberer, R. Reber, and M. Oppermann, An AIGaN/GaN class-S amplifier for RF-communication signals, IEEE/MTT-S International Microwave Symposium 2008, Jun. 2008, pp. 85-88. |
| Kobayashi, H. ; Hinrichs, J.M.; Asbeck, P.M.; Current-mode class-D power amplifiers for high-efficiency RF applications, Microwave Theory and techniques, Dec. 1, 2001, vol. 49, issue 12, pp. 2480-2485. |
| Number | Date | Country | |
|---|---|---|---|
| 20130142283 A1 | Jun 2013 | US |
