Patent Grant
10177713
This disclosure related to improvements in current mirrors, current sources, amplifiers, and output buffer drives for use in integrated circuits (ICs).
Operating ICs under ultra low currents and low power supplies, in complementary metal-oxide semiconductor (CMOS) technology, pose serious challenges in the design of integrated circuits. Low operating currents cause lower speeds and lower gain and higher noise in an IC. Also, rail-to-rail operations for ICs becomes a necessity given that signal-to-noise requirements at low power supplies demand input and output terminals of ICs to get as close as possible to the power supplies.
An aspect of the present disclosure is to make small and low cost current sources, current mirrors, amplifiers, buffers drivers, and buffer amplifiers that can operate with one or more of the following characteristics: (a) wide input-output voltage span, (b) low power supply voltage, (c) low power consumption, (d) low noise, (e) fast dynamic response, (f) symmetric design to minimize systematic errors, (g) use simple design that generally improves performance to specifications over operating and process variation, and/or (h) make the IC rugged for long term manufacturing using standard CMOS fabrication process that is inexpensive, and readily available at multiple fabrication factories, thus easing process node portability.
Another aspect of the disclosure includes a current mirror with high impedance, wide input-output span, and low drain voltage (VDDVDD). Another aspect of this disclosure is to make current sources and current mirrors with high output impedance, wide input-output voltage range, and operating with low power supply voltage. This is accomplished by a method of making a ‘current source’ or a ‘current mirror’ comprising regulated cascode current mirror (RGC-CM) coupled with diode connected self cascode (DCSC). This may also be accomplished by another method of making a ‘current source’ or a ‘current mirror’ comprising RGC-CM couple with inverting current mirror amplifier (ICMA). Moreover, this goal is met by another method of making a ‘current source’ or a ‘current mirror’ comprising RGC-CM coupled with a composite amplifier (CSGA) that contains common source armplifier (CSA) and common gate amplifier (CGA) with the gate of CGA connected to source of CSA.
Another aspect of the disclosure further includes an amplifier using the disclosed current mirrors discussed above to make amplifiers with high gain and wide input-output range, and operating with low power supply voltage. This is accomplished by a method of making an ‘amplifier’ comprising plurality of the disclosed RGC-CM couple with DCSC. This may also be accomplished by another method of making an ‘amplifier’ comprising plurality of the disclosed RGC-CM coupled with ICMA. Moreover, this goal is also met by method of making an ‘amplifier’ utilizing plurality of the disclosed RGC-CM coupled with a composite amplifier (CSGA).
Another aspect of the disclosure further includes a floating current source with low VDDVDD that also operates fast. Another aspect of this disclosure is to emulate the function of a floating current sources (FCS) that equalizes upper and lower current sources and is capable of operating with low power supply voltages. This may be accomplished by a method utilizing cascoded PMOSFETs (e.g., one PMOSFET on top and second PMOSFET in the middle), and cascoded NMOSFETs (e.g., one NMOSFET on top and second NMOSFET in the middle). The middle PMOSFET and the middle NMOSFET drain and source currents are crisscrossed and fed to one another. A lower regulating circuit holds the VGS of the middle NMOSFET constant by regulating the gate voltage of the lower NMOSFET. An upper regulating circuit holds the VGS of the middle PMOSFET constant by regulating the gate voltage of the upper PMOSFET. As a result the net sum of current in the upper PMOSFET and lower NMOSFET is equalized.
A further aspect of this disclosure is an amplifier using the disclosed FCS so that the amplifier's upper and lower current sources are equalized in order to improve amplifier's performance. Some of such improvements are in reducing the amplifier's offsets due to current asymmetries when the amplifier's input has a wide common mode span, while enabling the amplifier to operate with low power supply voltage.
A further aspect of this disclosure is reducing an amplifier's output noise, when the amplifier consumes low currents, operates at low VDDVDD, while the amplifier has wide input-output voltage span, and it is fast. Another aspect of this disclosure is to reduce an amplifier's noise by narrow banding it. When the amplifier's inputs are imbalanced in response to a large transient input signal, to make up for lost speed due to narrow-banding the amplifiers, the operating current of the amplifiers is dynamically and rapidly boosted to IPeak=h×IQ (in the current boost-on phase) where IQ is the steady state operating current of the amplifier set at low current levels to save power consumption. After the amplifier's inputs approach steady state and are near balance (not precisely equal, just roughly in balance), the amplifier's operating current is dynamically and slowly decreased (current boost-off phase) back to IQ levels. The amplifier dynamic response is improved by enhancing both the Slew Rate (SR) and the settling time (τS) of the amplifier. In summary, to rejovinate the (narrow banded) amplifier's SR while its' inputs are imbalanced due to receiving a large-signal transient, the amplifiers' operating current receive a very fast IPeak pulse during the current boost-on phase. Moreover, to rejovinate the (narrow banded) amplifier's τS when the amplifier's inputs approach balance and amplifiers enters the boost-off phase, instead of shutting off Ipeak very fast, the amplifier's operating current starts decaying slowly (following a time constant that tracks the amplifier's AC response) from Ipeak towards IQ. Accordingly, when the amplifier operating current is at IQ equilibrium or steady state, the amplifier's noise is reduced since it is narrow banded.
Another aspect of the disclosure herein is a buffer driver that operates with low VDDVDD, has wide input output voltage span, and is fast by operating chiefly in current mode. A further aspect of this disclosure is an amplifier using the disclosed buffer driver that can operate with low VDD, have near rail-to-rail input-outputs spans, and mostly operate in current mode which improve speed at low currents. This is accomplished by a method of making an ‘amplifier’ comprising plurality of minimum current selectors (MCS) or loser take all (LTA) which, directly or indirectly, monitor the sink-source currents of the buffer driver's output field effect transistors (FETs). Concurrently, a non-inverting current mirror amplifier (NICMA) or inverting current mirror amplifier (ICM) or inverting current feedback amplifiers (ICFA) would regulate the minimum stand-by currents for either the inactive sink output transistor or the inactive source output transistors. Also, in order to lower the current consumption associated with monitoring the sink source output transistor currents, a complementary non-inverting current mirror (CNICM) is utilized to (rectify) curb the sink-source signals before they are fed to the MCS or LTA.
Aspects of the embodiments disclosed herein further include that they can often be fabricated in standard digital CMOS; and embodiments have small size (e.g., for low cost and high volume applications); and the embodiments typically operate MOSFETs in subthreshold so they can operate at ultra low currents and low power supply voltages needed in particular to emerging wireless and battery less applications.
Aspects of the embodiments disclosed herein further include a method of operating an ultra low power Buffer Amplifier (BA), containing an Amplifier coupled with a Buffer Driver (BD), comprising: increasing gain and widening the input-output voltage span of the Amplifier utilizing a plurality of regulated cascode (RGC) current mirrors (RGC-CM) where each RGC-CM is made of at least one of the following three circuits: 1) a diode connected self cascode (DCSC) coupled with a common source amplifier (CSA); 2) a current mirror amplifier (CMA), inverting or non-inverting type, that contains a common source amplifier (CSA); and 3) a CSA coupled with one common gate amplifier (CGA) wherein the common gate terminal of the CGA is connected to the common source terminal of the CSA; lowering the minimum operating power supply and reducing offset of the Amplifier by utilizing a current equalizer circuit that emulates the function of a floating current source (FCS) having at least two complementary cascoded current sources made of field effect transistors (FETs), wherein middle cascoded FET's gate to source voltages (VGS) are held constant by regulating the VGS of the lower FETs, whose currents are equalized and mirrored into the Amplifier's bias network, where the lower FET's source terminals are connected to the power supplies; lowering output noise of the Amplifier, by narrow banding the Amplifier's high gain node, while concurrently rejuvenating the narrow banded Amplifier's speed by utilizing one of the following circuits: 1) minimum current select (MCS); and 2) loser take all (LTA) circuits to generate dynamically boosted operating current when the Amplifier's inputs are intermittently imbalanced; and lowering the minimum operating power supply, having near rail-to-rail input output voltage span, having high-speed, providing large sink-source current for output load, while regulating the operating current in the inactive sink-source transistor of the Buffer Amplifier (BA) by utilizing a Buffer Driver (BD) that contains at least one of the following circuits: 1) MCS; 2) LTA; 3) current mirror amplification (CMA); and 4) complementary current mirror that enable the Buffer Driver (BD) to chiefly operate and process signals in current mode.
Aspects of the embodiments disclosed herein further include a method of reducing output noise in an amplifier comprising: narrow banding the amplifier to keep the amplifier's static current consumption low; rejuvenating the dynamic response of the narrow banded amplifier by dynamically boosting the amplifier's operating current when the amplifier's inputs receive a large transient signal that cause an imbalance at the amplifier's inputs; and returning back to the steady state conditions when the amplifiers inputs are substantially equalized and shutting off the dynamic boosting of the amplifier's operating current when the amplifier's current consumption returns to low levels and the amplifier's output noise is reduced. The method further comprising: connecting a first capacitor to the high impedance or high gain node of the amplifier to narrow band the amplifier; and wherein the first capacitor can be an active or a passive capacitor that is intrinsic or extrinsic at the high impedance or high gain node of the amplifier. The method further comprising: rejuvenating the dynamic response of the narrow banded amplifier by speeding up the amplifier's slew rate and settling time upon detecting an imbalance at the amplifier's inputs by utilizing one of a: 1) a loser take all (LTA) circuit; and 2) minimum current selector (MCS) circuit; applying the output of the LTA circuit or MCS circuit to generate a ‘boost on’ or ‘boost off’ signal; using the ‘boost on’ signal to boost the operating current of the amplifier; and using the ‘boost off’ signal to return the amplifier's operating current back to the low static current at steady state condition and attaining lower output noise for the amplifier at steady state conditions. The method further comprising: using the ‘boost on’ signal to rapidly boost the operating current of the amplifier to increase the amplifier's slew rate (SR); using the ‘boost off’ signal to generate a slow declining current, with a slow decay to zero to speed up the settling time (τs) of the amplifier; improving the dynamic response of the amplifier by optimizing both its SR and τs; and using the ‘boost off’ signal to return the amplifier's operating current back to the low static current at steady state condition and attaining low output noise for the amplifier during steady state conditions.
Aspects of the embodiments disclosed herein further include a buffer driver circuit comprising: first output driver field effect transistors (FETs) having the function of sinking and sourcing currents for an external load; at least one of a minimum current selector (MCS) signal or at least one loser take call (LTA) signal having the function of monitoring and processing the sink-source currents of the first output driver FETs; at least one of a group consisting of a: 1) current mirror amplifier (CMA); 2) an inverting CMA (ICMA); and 3) non-inverting CMA (NICMA) having the function of receiving the MCS or LTA signals, and utilizing the MCS or LTA signals for regulating and controlling the current in the inactive sink-source FET; a first buffer driver that utilizes at least one MCS or LTA signals and at least one of the CMA, ICMA and NICMA; and wherein the buffer driver has the function of sinking and sourcing current for external loads and regulating a minimum operating current in the inactive sinking or sourcing FET. The circuit further comprising: one of the group consisting of: 1) a first complementary non-inverting current mirror (CNICM); and 2) a first complementary inverting current mirror (CICM) having a function of monitoring the sink-source output driver FETs current and generating rectified sink-source signals; and providing the rectified sink-source signals to the minimum current selector (MCS) or loser take call (LTA) circuits to process the sink-source output driver FETs signals. The circuit capable of being used in a first amplifier further comprising: wherein the first amplifier utilizes the first buffer driver circuit in order for the first amplifier to be coupled with the buffer driver to be capable of sinking and sourcing current for external loads and regulating a minimum operating current in the inactive sinking or sourcing FET.
Aspects of the embodiments disclosed herein further include a method of operating at least one regulated cascode (RGC) current mirror (RGC-CM) comprising: supplying voltage to the at least one RGC-CM from a positive supply voltage (VDD) and a negative supply voltage (VSS); increasing the output resistance of the at least one RGC-CM by a first auxiliary amplifier; and widening the voltage span of the input-output terminals of the at least one RGC-CM by generating a direct current (DC) voltage shift from at least one diode connected self cascode (DCSC) coupled to the first auxiliary amplifier. The method further comprising: utilizing a first amplifier that contains a plurality of RGC-CMs, wherein each of the plurality of RGC-CMs are utilized in the first amplifier to function as current mirrors; delivering power to the first amplifier by a positive supply voltage (VDD) and a negative supply voltage (VSS); increasing the gain of the first amplifier by utilizing the plurality of RGC-CMs; and widening the input-output span of the first amplifier by utilizing the plurality of RGC-CMs.
Aspects of the embodiments disclosed herein further include at least one regulated cascode (RGC) current mirror (RGC-CM) circuit comprising: cascoded transistors having a first transistor placed in series at a first node with a second transistor wherein the output of the RGC-CM is the drain terminal of the second transistor; a first diode connected self cascode (DCSC) having a diode connected third transistor in series with a fourth transistor, wherein the gate terminals of the third transistor and the fourth transistor are connected together and wherein the source of the third transistor is connected to the drain of the fourth transistor at the second node, wherein the source of the fourth transistor is connected to the first node; and a first auxiliary amplifier (AA) whose input is connected to the second node, and wherein the output of the first AA is connected to the gate terminal of the second transistor. The circuit utilized in a first amplifier further comprising: a plurality of the at least one regulated cascode (RGC) current mirrors (RGC-CMs); wherein each of the plurality of RGC-CMs function as current mirrors; wherein the gain of the first amplifier is increased by utilizing the plurality RGC-CMs; and wherein the input-output span of the first amplifier is widened by utilizing the plurality of RGC-CMs.
Aspects of the embodiments disclosed herein further include a method to operate a regulated cascode (RGC) current mirror (RGC-CM) comprising: providing power to the RGC-CM by a positive supply voltage (VDD) and a negative supply voltage (VSS); operating input and output signals in a current mode by utilizing a first inverting current mode amplifier (ICMA) that includes a first inverting current mirror (ICM); generating amplification through the first ICMA to perform the function of a first auxiliary amplifier (AA); increasing the output resistance of the RGC-CM by utilizing the first ICMA; and widening the input-output terminal spans of the RGC-CM by utilizing the first ICMA. The method of comprising: utilizing a first amplifier having a plurality of regulated cascode (RGC) current mirrors (RGC-CMs) wherein each of the plurality of RGC-CMs are utilized in the first amplifier function as current mirrors.
A regulated cascode (RGC) current mirror (RGC-CM) circuit comprising: cascoded transistors having a first transistor placed in series at a first node with a second transistor wherein the output of the RGC-CM is the drain terminal of the second transistor; a first current mirror (CM) with a diode connected third transistor and a fourth transistor; a first current mirror amplifier (CMA) having the first current mirror, and a third transistor whose source terminal, which is the input of CMA, is connected to the first node and its drain terminal is connected to the gate and drain terminals of the third transistor; wherein the output of CMA, which is the drain terminal of the fourth transistor, is connected to the gate terminal of second transistor; and wherein the CMA functions as the auxiliary amplifier for the RGC-CM in order to increase the output resistance of the RGC-CM. The circuit used in a first amplifier further comprising: wherein each of the plurality of RGC-CMs that are utilized in the first amplifier function as current mirrors; wherein the gain of the first amplifier is increased by utilizing the plurality RGC-CMs; and wherein the input-output span of the first amplifier is widened by utilizing the plurality of RGC-CM.
Aspects of the embodiments disclosed herein further include a regulated cascode (RGC) current mirror (RGC-CM) circuit comprising: cascoded transistors having a first transistor placed in series at a first node with a second transistor wherein the output of the RGC-CM is the drain terminal of the second transistor; a first common source amplifier (CSA); a first common gate amplifier (GGA); wherein the inputs of the first CSA are connected to the source terminal of the first transistor and the first node; wherein the output of the first CSA is connected to the input of the first CGA; wherein the common gate terminal of the first CGA is connected to the common source terminal of the first CSA; and wherein the first CGA's output terminal is connected to the gate of the second transistor. The circuit used in a first amplifier further comprising: a plurality of regulated cascode (RGC) current mirrors (RGC-CMs); and wherein each of the plurality of RGC-CMs that are utilized in the first amplifier function as current mirrors; wherein the gain of the first amplifier is increased by utilizing the plurality RGC-CMs; and wherein the input-output span of the first amplifier is widened by utilizing the plurality of RGC-CM.
Aspects of the embodiments disclosed herein further include a regulated cascode (RGC) current mirror (RGC-CM) circuit comprising: cascoded transistors having a first transistor placed in series at a first node with a second transistor wherein the output of the RGC-CM is the drain terminal of the second transistor; a first common source amplifier (CSA), with a built in offset, having a third transistor and a fourth transistor, whose source terminals are connected to the second node, wherein the gate of the third transistor which is one of the inputs of CSA is connected to the first node, wherein the other terminal of the CSA, which is the gate and drain terminals of the fourth transistor are connected together and are connected to the source terminal of the first transistor; and a first common gate amplifier (CGA), comprising a fifth transistor whose gate terminal is connected to the second node, where the input of the CGA which is the source terminal of is connected to the first node, and the output of the CGA which is the drain terminal of the fifth transistor is connected to the gate terminal of the second transistor. The circuit used in a first amplifier, further comprising: a plurality of regulated cascode (RGC) current mirrors (RGC-CMs), wherein each of the plurality of RGC-CMs that are utilized in the first amplifier function as current mirrors; wherein the gain of the first amplifier is increased by utilizing the plurality RGC-CMs; and wherein the input-output span of the first amplifier is widened by utilizing the plurality of RGC-CMs.
Aspects of the embodiments disclosed herein further include a current equalizing circuit comprising: a positive supply voltage (VDD) and a negative supply voltage (VSS); a first Positive Metal Oxide Silicon Field Effect Transistor (PMOSFET) and second PMOSFET forming a cascoded current source, wherein the drain of the first PMOSFET is connected to the source of the second PMOSFET at the first node; a first Negative Metal Oxide Silicon Field Effect Transistor (NMOSFET) and second NMOSFET forming a cascoded current source, wherein the drain of the first NMOSFET is connected to the source of the second NMOSFET at the second node; wherein the second PMOSFET drain terminal is connected to the source terminal of and second NMOSFET at the second node; wherein the second NMOSFET drain terminal is connected to the source terminal of and second PMOSFET at the first node; wherein a first regulating circuit keeps the gate-to-source voltage of the second PMOSFET substantially constant by regulating the gate-to-source voltage of the first PMOSFET; wherein a second regulating circuit keeps the gate-to-source voltage of the second NMOSFET substantially constant by regulating the gate-to-source voltage of the first NMOSFET; and wherein the current in the first PMOSFET and first NMOSFET are substantially equalized. The circuit used in a first amplifier further comprising: the first amplifier including the first current equalizing circuit of claim 22.
Aspects of the embodiments disclosed herein further include a current equalizing method comprising: delivering power to the current equalizer by a positive supply voltage (VDD) and a negative supply voltage (VSS); lowering the minimum operating power supply and reducing offset of the Amplifier (A) by utilizing a current equalizer circuit that emulates the function of a floating current source (FCS) containing two complementary cascode current sources, wherein the cascoded field effect transistors (FETs) gate to source voltages (VGS) are held constant by regulating the VGS of the lower FETs, whose currents are equalized and mirrored into the amplifier's bias network, and whose source terminals are connected to the power supplies.
The following terms, definitions, acronyms, term usages and abbreviations are explained below and used throughout this application:
Numerous embodiments are described in the present application and are presented for illustrative purposes only and is not intended to be exhaustive. The embodiments were chosen and described to explain principles of operation and their practical applications. The present disclosure is not a literal description of all embodiments of the disclosure(s). The described embodiments also are not, and are not intended to be, limiting in any sense. One of ordinary skill in the art will recognize that the disclosed embodiment(s) may be practiced with various modifications and alterations, such as structural, logical, and electrical modifications. For example, the present disclosure is not a listing of features which must necessarily be present in all embodiments. On the contrary, a variety of components are described to illustrate the wide variety of possible embodiments of the present disclosure(s). Although particular features of the disclosed embodiments may be described with reference to one or more particular embodiments and/or drawings, it should be understood that such features are not limited to usage in the one or more particular embodiments or drawings with reference to which they are described, unless expressly specified otherwise. The scope of the disclosure is to be defined by the claims.
Although process (or method) steps may be described or claimed in a particular sequential order, such processes may be configured to work in different orders. In other words, any sequence or order of steps that may be explicitly described or claimed does not necessarily indicate a requirement that the steps be performed in that order. The steps of processes described herein may be performed in any order possible. Further, some steps may be performed simultaneously despite being described or implied as occurring non-simultaneously (e.g., because one step is described after the other step). Moreover, the illustration of a process by its depiction in a drawing does not imply that the illustrated process is exclusive of other variations and modifications thereto, does not imply that the illustrated process or any of its steps are necessary to the embodiment(s). In addition, although a process may be described as including a plurality of steps, that does not imply that all or any of the steps are essential or required. Various other embodiments within the scope of the described disclosure(s) include other processes that omit some or all of the described steps. In addition, although a circuit may be described as including a plurality of components, aspects, steps, qualities, characteristics and/or features, that does not indicate that any or all of the plurality are essential or required. Various other embodiments may include other circuit elements or limitations that omit some or all of the described plurality.
Throughout this disclosure, the body terminal of PMOSFETs can be either connected to their respective PMOSFET source terminals or to the positive power supply, VDD. Similarly here, the body of NMOSFETs can be either connected to their respective NMOSFET source terminal or to the negative power supply, VSS. Moreover, the negative supply voltage, VSS, can be alternatively connected to the ground (GND) potential. Given that one of these teaching's target application are for ultra low power and portable electronics, the transistors utilized in circuits operate under the subthreshold region, but it is also possible to operate transistors, throughout this disclosures and illustrations, in the normal regions. Throughout this disclosure and its illustrations, current sources or current mirrors may be constructed with single FETs or cascodes one, depending on cost-performance considerations such as die size, output impedance, gain, speed, head room, amongst others. Illustrations in some embodiments utilizing NMOSFET (e.g., in a current mirror or RGC-CM), can be modified to utilize their complementary FET counterparts (i.e., PMOSFETs). Moreover, embodiments utilizing amplifier's with PMOSFET input stages can also be modified to utilize NMOSFET input stages, or complementary input stage (i.e., both PMOSFET and NMOSFET operating in parallel) for rail to rail input dynamic range. Moreover, embodiments can utilize amplifiers, or their variations obvious to one skilled in the art, that use double PMOSFETs coupled with NMOSFET level shifters or double NMOSFETs coupled with PMOSFET level shifters to attain rail-to-rail input operations. The U.S. Provisional Patent Application Ser. No. 62/304,373 utilizes an amplifier's input stage based in complementary input pairs (i.e., both PMOSFET and NMOSFET operating in parallel) for rail to rail input dynamic range. The U.S. Provisional Patent Application Ser. No. 62/320,512 and the U.S. Provisional Patent Application Ser. No. 62/415,496 utilize an amplifier's input stage that are based in double PMOSFETs coupled with NMOSFET level shifters to attain input voltage rail-to-rail operations. For demonstrative clarity and simplicity, the assumption is made that VA, β, η, gm, gds, rds, rO, and ID for PMOSFETs and NMOSFETs are substantially equal unless otherwise specified. In order to demonstrate achievable typical specifications, simulations are performed on some of the circuits illustrating the embodiments. These simulations are not intended to guarantee the embodiment performance to a particular range of specifications. Circuit simulations use the Top-Spice simulator, and are based on approximate device models for a typical standard 0.18 μm CMOS process fabrication. To simulate for sensitivity of design to device model variations (i.e., simulate for design margin), this disclosure provides some worst case simulations (WC) that subject the circuit embodiments to variations in device parameters (e.g., manufacturing process fluctuations in VTH and COX), which can indicate the circuit's design sensitivity to normal manufacturing (wafer lot-to-lot) variations. For example, WC circuit simulations illustrate variations on performance specifications such as slew rate of settling time as a function of varying VTH by ±10% from center value, and independently from varying COX by ±5% from center value (i.e., 16 process corner combinations).
Note that the teachings in this disclosure are applicable to high voltage or high current or high speed applications, and combinations thereof, as well. The teachings in this disclosure may be applied to other manufacturing fabrication processes that contain transistors, such as fabrication processes with deep sub-micron CMOS, BiCMOS, BJT, and JFET, amongst other device and manufacturing platforms that make devices that can function as transistors.
Current mirrors, current sources, amplifier, and output buffers are building blocks for any integrated circuits (IC), including for higher order functions such as analog to digital converters (ADC), digital to analog converters (DAC), regulators, references, fitters, data acquisition systems (DAS), and other building blocks in any analog and mixed mode ICs, and system on a chip (SOC). It is advantageous for IC building blocks such as current sources, current mirrors, amplifiers, and buffer drivers to: (1) consume low currents, (2) operate with low power supply voltages, especially for portable applications, (3) have near rail-to-rail input-output spans since there is little voltage headroom to waste, especially under low power supply voltages, (4) have high output impedance or high output gain, especially under ultra low operating currents when gain is lower, (5) have high-speeds, especially under low operating currents when speed gets slower, (6) have low noise, especially under low operating currents when noise increases, (7) have their other performance specifications, such as power supply rejection and common mode rejection, amongst others, unimpeded when utilizing circuit arrangements to improve gain, speed, and noise operating under low operating currents, and low power supply voltage conditions, (8) perform to specifications over fabrication process variations, and under different power supply and temperature conditions, (9) be small so they cost less, and be able to integrate multiple building block channels on the same die for better channel-to-channel matching, (10) use standard CMOS fabrication for lower cost, availability, and proven quality.
As the demand for portable, green electronics, and energy harvesting (e.g., self-powered ICs) continues to grow, so does the requirement for circuits with lower power supplies and ultra low current consumptions. Low power electronics also require performance to specifications at low VDD and VSS where there is less available voltage headroom to process signals. Meeting low power consumption and signal-to-noise specifications at low VDD and VSS require ICs with the widest input-output dynamic range possible. Although special fabrication processes can provide special transistors that enable a circuit to operate with lower power supplies and lower current consumption, but such special manufacturing are generally expensive, and may create a high cost barrier for the full market potentials to be realized in a timely manner.
Also, low power (ultra low current and low power supply voltage combined) may be required in some medical or defense applications where it is not safe to frequently replace an implantable or embedded battery operated IC. This factor would require ultra low power so that the ICs draw extremely low current so that the battery life is significantly extended.
Another example is next generation energy harvesting electronics that is wireless and battery less. They can function perpetually without ever needing to be connected to power source and with no need of being recharged. There are sub-categories of energy harvesting ICs that can be designated as self-powered ICs. Energy harvesting is part of green electronics that rely on harvesting or scavenging energy from the environment such as solar, mechanical, thermal, or magnetic, to name a few. These kinds of energies help generate voltage potential that can for example be stored on a super capacitor, which can power ultra low power electronics for signal processing.
Additionally, some biometrics system on a chip may require small die size for applications requiring multi-channel current mirrors or amplifiers for conditioning multiple sensor's outputs simultaneously. For example, multiple sensors implanted in tooth dentures that simultaneously read levels of sugar, salt, acidity, temperature, and other non-vitals, which may require and at a minimum will benefit from small circuit size for better matching between each circuit channels in one die.
Furthermore, to target high volume markets and reduce the risk of low yields for long term production, manufacturing organizations generally have an unfavorable view of circuits that require special processes. Optimal yield and quality generally avoid circuits that require variations to a standard process, or complex circuits whose specifications may depend on multiple device or (manufacturing) process parameters. Generally circuits requiring non-standard fabrication, or complex circuits are harder to optimize for maximal production yields, or they may compromise rugged end-product manufacturing (quality) goals. It is also of note that complex circuits may hinder transient response, in start up and turn off phases, for example. This may be a risky trait, particularly in energy harvesting applications that may subject the IC to less predictable or disorderly power supply on and off patterns in the field.
Besides ultra low power, emerging applications such as energy harvesting that was noted above and bio-metrics, require small circuits to keep the costs down in order to realize their full and highest volume market potentials.
As stated earlier, current mirrors are fundamental building blocks in electronics. Some of performance specifications for a current mirrors are to have high ROUT and wide input-output voltage spans. Also, making current mirrors with simple circuits that are low power and low cost is beneficial to cost-performance-quality tradeoffs, including for high order other building blocks such as amplifiers, where current mirrors are utilized.
Utilizing Regulated Cascode (RGC) is a way to increase the Rout of a current mirror.
Operating at low or ultra low operating currents, generally causes an amplifier's gain to be lower. Utilizing RGC-CM is a way to increase the gain of amplifiers, in part, by way of increasing the amplifier's ROUT at the amplifier's high-gain node.
Increasing an amplifier's gain, utilizing RGC-CM, without restricting its input and output voltage span, are disclosed in the literature and in the references provided, including in [S. Yan and E. Sanchez-Sinencio, “Low voltage analog circuit design: A tutorial”, IEICE Trans. Analog Integrated Circuits and Systems, vol. E83A , no. 2, pp. 179-196, 2000], and [P. E. Allen & D. R. Holberg, CMOS Analog Circuit Design, 2nd Ed, Oxford University press, 2002]. Generally, the available RGC-CMs with a wide input-output voltage span have higher transistor count (i.e., larger die size), consume high power, have unfavorable transient response considering current consumption, or are complex (e.g., performance depends on different types of FETs, or performance depends on multiple device parameters) which are generally unfavorable for rugged manufacturing in terms of cost-performance-quality traits.
As noted earlier, low voltage electronics require rail-to-rail input-output operations. A folded cascode transconductance amplifier (FCTA), is a suitable amplifier topology for rail to rail operations. From a high level perspective, FCTA contains a common source amplifier (CSA) at its inputs. Then the differential current outputs of this CSA feed the differential current input of a common gate amplifier (CGA) whose differential output currents are fed onto a differential input to single output current mirror, that sum at the FCTA high impedance output node to make an output voltage (VOUT). The regulated cascode current mirror (RGC-CM) is generally used in the CGA and current mirror sections of an amplifier to improve their ROUT, which increase's the amplifier's gain. Moreover, its beneficial for the RGC-CM to have wide input-output voltage span in order not to restrict the amplifier's input-output voltage span, in which it is utilized. Operating the inputs of a FCTA rail to rail is generally accomplished by running pairs PMOSFET and NMOSFET (complementary) CSAs in parallel as inputs. Near the rails, either PMOS CSA runs out of headroom while the NMOS CSA takes control and keeps feeding the next CGA gain stage, or vice versa.
Given the wide common mode range in an amplifier is especially beneficial for near rail-to-rail input-output voltage spans, utilizing a floating current source (FCS) or emulating its equivalent function, is generally a way to reduce the errors generated in upper and lower current source in the CGA and current mirror (summing node) gain stage. However, in order to operate the amplifier with lowest VDD (besides operating the amplifier inputs-outputs rail-to-rail), all of the amplifier's elements, including the FCS need to operate with lowest VDD.
Generally an amplifier's high gain stage is coupled with a buffer driver, which makes a buffer amplifier that would be able to drive external loads. In order to operate the buffer amplifier with ultra low currents and in low power supply environment, the buffer driver contained in it, must also consume ultra low currents and be capable of operating to specification with low power supplies. As noted earlier, operating in low or ultra low currents, slows down the speed. Hence, it would be advantageous to have a buffer driver that is fast, inherently, at low currents and can operate with low power supply voltage and have near rail-to-rail input-output voltage spans. Running fast and operating with low currents at low power supplies, still requires a buffer driver with high sink-source current drive capability to handle current requirements of different external loads. For example, some emerging portable applications use resistive sensors to measure environmental toxicity. The resistivity of such toxicity sensors can drop significantly when they are activated to make a measurement. Hence, it would be advantageous to have a buffer driver that can handle external low resistive loads when the sensor is activated, but return to low currents consumptions seamlessly, when the resistive sensor is no longer activated.
As mentioned earlier, next generation energy harvesting, and wireless and batteryless electronics are emerging applications, that require ultra low power ICs. All else equal, operating analog ICs at ultra low currents present additional challenges such as high noise besides low gain, and slow dynamic response (e.g., as noted earlier and generally, the lower the current, the higher the noise, the lower the gain, and the lower the speed). Hence, it would be advantageous to have a low noise amplifier noise that consumers low operating currents, and one that can be fast and operate with low power supply voltage.
Therefore, in summary here are a list of advantages of these teachings. One, is to make current mirrors that are simple, small, low cost, low power, have high ROUT, and wider input-output voltage span, whose embodiments are illustrated in
Note that the following papers providing additional analysis of relevance to low power and low cost amplifier designs are also hereby incorporated by reference in their entirety: (1) A. Far, “Small size class AB amplifier for energy harvesting with ultra low power, high gain, and high CMRR,” 2016 IEEE International Autumn Meeting on Power, Electronics and Computing (ROPEC), Ixtapa, Zihuatanejo, Mexico, 2016, pp. 1-5; (2) A. Far, “Amplifier for energy harvesting: Low voltage, ultra low current, rail-to-rail input-output, high speed,” 2016 IEEE International Autumn Meeting on Power, Electronics and Computing (ROPEC), Ixtapa, Zihuatanejo, Mexico, 2016, pp. 1-6; (3) A. Far, “Low Noise Rail-To-Rail Amplifier for Energy Harvesting Runs Fast at Ultra Low Currents,” 2017 IEEE Canadian Conference on Electrical and Computer Engineering (CCECE), Windsor, ON, 2017; and (4) A. Far, “Ultra Low Current and Low Voltage Class AB Buffer Amplifier,” 2017 IEEE Canadian Conference on Electrical and Computer Engineering (CCECE), Windsor, ON, 2017.
SECTION (I): Detailed Description of Regulated Cascode Current Mirror (RGC-CM) Coupled with Diode Connected Self Cascode (DCSC), as Illustrated in
The basic idea of RGC-CM100B coupled with DCSC100B in
Throughout the description of
The connections of the elements in
Describing the details of the circuit in
The current mirror of
In
In summary, the prior art illustrated in
In conclusion, the benefits of the
Section (II): Detailed Description of Regulated Cascode Current Mirror (RGC-CM) Coupled with Inverting Current Mirror Amplifier (ICMA), as Illustrated in
The connections of the elements in
Note that the ICMA100D function is performed by M115D, M113D, I107D, M111D, and I105D. For clarity in describing the RGC-CM100D coupled with ICMA100D of
During steady state conditions, I113D flows through M128D, whose VGS is the same as that of M117D, which causes v×IM128D=u×IM117D or 2IM128D=IM117D. As such, IDM111D≈IDM113D≈IDM115D≈IDM121D≈IDM119D≈½×IDM117D≈IDM128D≈i≈5 nA.
As noted earlier, the regulated cascode (RGC) is used in a current mirror (CM), chiefly, to increase its ROUT. When, there are voltage changes (ΔvX) at the output (node 103D), the goal is for the output current (IDM119D=i) variations (ΔiX) to be minimized. The ΔvX here can cause a ΔiX on M119D, because M119D has a finite impeadance between its source and drain terminals. Since IM117D=2I128D=2i is fixed, then at node 104A, the ΔiX would flow through M115D, which is the current input terminal of ICMA100D. Given that I107D=2i is also constant, as explained earlier, this change in current, ΔiX, flowing through M115D gets subtracted from the steady state current of IDM113D. The auxiliary amplification, AAUX, function inside the ICMA100D is performed by M111D and I105D whose ROUT∝VA/ID (when CMOS is operating in the subthreshold region). Consequently, the ΔiX in IDM113D is mirrored onto M111D whose drain (at node 101D is coupled to I105D=i that is a constant current source) responds to this ΔiX, with negative gained voltage change of about ΔiX×ROUT. At node 101D, this negative gained voltage change is fed-back to the gate terminal of M119D. This negative gain voltage feedback, is the mechanism that regulates the VGS of M119D so that ΔiX→near zero.
In summary, the ICMA100D regulates the gate voltage of M119D in order to increase the ROUT of at node 103D. The ROUT of RGC-CM100D coupled with ICMA100D is approximately ∝gm2×rds3∝(VA/VT)2×(VA/ID), assuming subthreshold operations and assuming that the gain from node 104D to 102D is about unity with equal currents flowing through M115D, M113D, and M111D.
The RGC-CM100D coupled with ICMA100D output voltage span is also improved substantially, mainly because of the flexibility in setting V105D fairly independently (e.g., V105D∝I109D, and W/L of M121D) which bias the VGM115D. The minimum V103D=VonM119D+VonM117D, and VonM117D=VGSM121D−VGSM115D=ΔVGS≈VT×ln(r/n). Hence, minimum V103D=VonM119E+VT×ln(r/n) above VSS, and approximately independent of CMOSFET's VTH.
In conclusion, the benefits of the proposed RGC-CM100D coupled with ICMA100D illustrated in of
Section (III): Detailed Description of Regulated Cascode Current Mirror (RGC-CM) Coupled with Composite Amplifier (CSGA), Illustrated in
Throughout the description of
The connections of the elements in
The details of various functions of the circuit in
The CSGA100F of
As indicated earlier, it is possible to connect the gate terminal of M111F (the other input of the CSA) to a secondary bias voltage instead of VSS or GND that established the VGM111F (e.g., VGM111F+VGSM111F−VGSM115F≈VDS117F VDS), which would then allow to make r=s=1. This arrangement would save area since M115F or M111F needs not be scaled, and the secondary bias voltage would facilitate the headroom needed at node 104F
Node 101F which is the summing junction of CSA (source terminals of M111F and M115F) establishes a DC voltage that biases VGM113F which is the gate terminal of the CGA. The CGAs are inherently fast and CSA output (drain terminal of M115F) is fed into the current input of CGA (source terminal of M113F). Therefore, the manner of arranging the RGC-CM100F coupled with CSGA100F in
In summary, voltage movements, ΔvOUT, at node 105F (output terminal of the current mirror) cause a voltage, ΔvX, at node 104F. The Δvx would cause a negative change in current, −ΔiX, in node 103F, which in turn cause a gained voltage change at the output of the CSGA100F, ≈−ΔiX×−ROUT aux≈−G×ΔvX at node 102F, which is the gate of M119F. As a result, the source voltage of M119F follows its gate voltage, thereby regulating the voltage at node 104F until ΔvX→near zero again and as such the ROUT of the RGC-CM can be increased in this arrangement.
The gain, ‘G’, of the RGC-CM amplifier is approximately G∝VA/VT, since FETs operate in the subthreshold region. The Rout of RGC-CM is approximately ∝gm2×rds3∝(VA/VT)2×(VA/ID). As stated earlier, the ROUT of RGC-CM on actual silicon will be lower due to second order effects such as η (subthreshold slope factor) and substrate leakage, amongst other factors.
In summary, the benefits of the
Section (IV): Detailed Description of Amplifier (AMP100A) is Illustrated in
Note that alternative amplifier embodiments are possible such as an amplifier with NMOS input stage or complementary (PMOS and NMOS) rail-to-rail input stages, other amplifier topologies that are not FCTA, amongst others. The connections of the elements of AMP100A of
Note that five RGC-CM100B coupled with DCSC100B are utilized in the amplifier, AMP100A, embodiment of
For explanation regarding RGC-CM100B coupled with DCSC100B refer to its detailed description, but below is a brief description of how they operate and benefit a FCTA. The FCTA has generally 3 parts, ‘common source amplifier’ (CSA), ‘common gate amplifier’ (CGA), and a current mirror (CM). The VIN is applied to a differential CSA, containing M120A and M121A, whose output feed the CGA, containing M109A and M119A. The differential outputs of this CGA feed the CM, made up of M106A and M116A, to make a single ended output, VOUT which is also the high impedance node (125A) of FCTA. The current I111A=i controls VGSM126A that establishes IM126A, which is mirrored and scaled onto IM107A=IM117A=3i. The CSA's input currents IM120A≈M121A≈i are fed into the source terminals of M109A and M119A, respectively, which are the differential inputs of the CGAs. The I103A=i and I105A=i flow through the DCSCs of RGC102A and RGC103A, respectively, which are passed onto M107A and M117A, and in that order. Therefore, the operating currents of CGA's (containing M109A and M119A) are IM109A≈IM107A−I103A−IM120A≈3i−2i=i and IM119A≈IM117A−I105A−IM121A≈3i−2i=i. As explained in the RGC-CM coupled with DCSC in previous section regarding RGC103A, the auxiliary amplifier (composed of M111A-I105A) regulates the VGM119A which increase the output impedance of the RGC-CM. This same discussion is applicable to RGC102A, RGC104A, and RGC105A. As a result, effectively, the output impedance and gain of CGAs and current mirrors in FCTA is increased. Note that main amplifier's ROUT∝ro×(ro×gm)2 and Av is ∝ROUT×gmi. Hence, Av∝(ro×gm)3, in the subthreshold region of operations for CMOS. To save current consumption, note that Av can be increased by raising only I108A and I111A, instead of increasing the current consumption of the whole amplifier, given that Av is ∝ROUT×gmi, and gmi is the input stage transconductance that is about ∝I108A/VT
Moreover, the DC common mode range of the main amplifier inputs (M120A and M121A) is expanded because, with the DC voltage shift generated by the DCSCs in RGC102A and RGC103A, the V108A and V109A can get closer to the rails. The DC voltage shift generated in DCSC of RGC103A is approximately, VGSM113A−VGSM115A=ΔVGS≈VDSM113A≈VT×ln(r/s). Hence, V109A≈VGSM111A−VT×ln(r) from the negative rail.
Additionally, the same type of FETs (e.g., NMOSFETs) are utilized in the CGAs function, the DC voltage shift function that expands the VOUT span, and the auxiliary amplifier function that increases ROUT of the RGC-CM. Utilizing the same type of FETs here, improves the consistency of DC, AC, and transient performance over temperature, power supply, and process variations.
In summary, besides increasing an amplifier's gain, the benefits of utilizing plurality of RGC-CM100B coupled with DCSC100B in an amplifier are as follows. First, the same type (e.g., NMOSFET) FETs are used in each of the main amplifier's CGA, and RGC-CM's auxiliary amplifiers plus DCSCs. Given that same type (e.g., NMOSFET) FET's device parameters match and track each other, therefore the FCTA's consistency of DC and AC specifications and stability is improved over fabrication process, temperature, and power supply variations. Second, each of the amplifier's RGC101A to RGC105A are self biased (i.e., no need for separate voltages to set its internal bias nodes) saving current and die space. Third, the listed benefits of RGC-CM100B coupled with DCSC100B (see previous section) carry over to the amplifier, including small size, low current, and faster dynamic response. Fourth, the DC voltage shift provided in the RGC-CM100B coupled with DCSC100B widen the amplifier's input-output span closer to the rails. As stated before, for example, a 75 mV head-room expansion at the input and output of each of the upper PMOS and the lower NMOS based RGC-CM100B coupled with DCSC100B, can expand the voltage span at the input of the amplifier as well as at the high gain (impedance) node of FCTA by 150 mV or 15%, which is beneficial especially in the sub-1V power supply environment.
Section (V): Detailed Description of Amplifier (AMP100C) in
The connections of the elements in
Note that five RGC-CM100D coupled with ICMA100D are utilized in the amplifier of
For explanation regarding RGC-CM100D coupled with ICMA100D refer to its detailed description. A general description is provided for more context in how utilizing RGC-CM100D coupled with ICMA100D can improve the amplifier's performance. In the amplifier embodiment of
Via M127C and M128C, the current I113C≈i is mirrored and scaled onto IDM107C≈IDM117C≈3i. The FCTA input currents IM122C≈IM123C≈i (part of CSA) are fed into the source terminals of M109C and M119C, respectively, which are the differential inputs of the CGAs. The I103C=2i and I107C=2i flow through ICMAs of RGC102C and RGC103C, respectively, and passed onto M107C and M117C, and in that order. Therefore, the operating currents of CGA's are IDM109C≈IDM107C−I103C−IDM103C−IDM122C≈i and IM119C≈IDM117C−I107C−IDM113C−IDM123C≈i. As explained in the RGC-CM100D coupled with ICMA100D section, the auxiliary amplifier function (performed by ICMA containing M115C, M113C, M111C, I107C, and I107C) regulates the VGM119C which increase the output impedance and gain of FCTA's CGA (containing M111C), and thereby increases the ROUT and gain (Av) of the FCTA at node 121C. Note that main amplifier's ROUT∝ro×(ro×gm)2 and Av is ∝ROUT×gmi. Hence, Av∝(ro×gm)3, with the amplifier FETs operating in the subthreshold region. To save current consumption, note that amplifier's gain can be increased by raising only I110C and I113C, instead of increasing the current consumption of the whole amplifier, given that Av is ∝ROUT×gmi, and gmi is the input stage transconductance that is roughly ∝I110C/VT.
Moreover, the DC common mode range of the main amplifier inputs (M122C and M123C) is expanded. The M105C and M115C source terminals sense the 103C and 104C signals in current mode. The DC input and output range of the amplifier is limited by V103C and V104C, which can be predetermined by the gate voltages of M105C and M115C, which are set by the scale and current of M121C. As such, VGS121C−VGS115C=VGS121C−VGS105C=ΔVGS≈VDSM117C≈VDSM107C≈VT×ln(r/n), which enable the input and output voltages of the amplifier to can get much closer to the rails. Additionally, note that the same type of FETs (e.g., NMOSFETs) are utilized in the CGAs function, the DC voltage shift function that expands the VIN and VOUT span, plus the auxiliary amplifier function (performed by ICMA) that increases ROUT of RGC103C and RGC105C (and hence raises the Av of the amplifier) which improves consistency of DC, AC, and transient performance over temperature, power supply, and process variations
In summary, besides providing the extra gain, the benefits of utilizing plurality of RGC-CM100D coupled with ICMA100D in an amplifier are as follows. First, the same type (e.g., NMOSFET) FETs are used in each of the main amplifier's common gate amplifiers (CGA), and RGC-CM's auxiliary amplifiers (AAUX) whose function is accomplished by the ICMAs. Given that for example the NMOSFET device parameters match and track each other better, therefore the consistency and stability of the main amplifier's DC, AC, and transient specifications are improved over fabrication process, temperature, and power supply variations. If for example CGA was based on NMOFETS, and AAUX was based on mix of PMOSFETs and NMSOFETs, then that would increase the risk of inconsistencies in the amplifier's performance in the long run in manufacturing.
Section (VI): Detailed Description of Amplifier (AMP100E) Utilizing Plurality of RGC-CM100F Coupled with CSGA100F, as Illustrated in
Note that alternative amplifier embodiments may be possible such as an amplifier with NMOS input stage or complementary (PMOS and NMOS) rail-to-rail input stages, other amplifier topologies that are not folded cascode transconductance, amongst others.
The connections of the elements in
Note that there are five of RGC-CM100F coupled with CSGA100F that are utilized in the amplifier embodiment of
For explanation regarding RGC-CM100F coupled with CSGA100F, refer to its detailed description. A general description is provided for more context in how utilizing RGC-CM100F coupled with CSGA100F can improve the main amplifier's performance. The VIN is applied to a differential CSA, containing M120E and M121E, whose current outputs feed a differential CGA, containing M109A and M119A. The differential current outputs of this CGA feed a current mirror function containing M106E and M116E, to make a single ended output, VOUT. For the FCTA during steady state, the current I119E≈i controls VGSM131E that establishes IDM131E, which is mirrored and scaled onto IDM107E≈IDM117E≈2i. The FCTA's CSA input currents IM120E≈M121E≈i are fed into the source terminals of M109E and M119E, respectively, which are the differential inputs of the CGAs. At node 105E, the IDM120E≈1i subtracts from IDM117E≈2i which provides for IDM119E≈1i. At node 104E, the IDM121E≈i subtracts from IDM107E≈2i which provides for IDM109E≈1i. As explained in the RGC-CM coupled with composite amplifier in previous sections, the AAUX (containing M113E-I109E) regulates the VGM119E which increases ROUT of RGC-CM in RGC103E. Similar discussion in applicable to role of RGC102E, RGC104E, and RGC105E in FCTA here. As a result, effectively, the output impedance and gain of CGAs and current mirrors in FCTA are increased, and thereby the ROUT and gain (Av) of the FTCA are increased at node 115E. Note that main amplifier's ROUT is about ∝ro×(ro×gm)2 and Av is about ∝ROUT×gmi∝(ro×gm)3, in the subthreshold region of operations. To save current consumption, note that Av can be increased by raising only I112E and I119E, instead of increasing the current consumption of the FCTA, given that Av is ∝ROUT×gmi, and gmi is the input stage transconductance that is roughly ∝I112E/VT. Moreover, the input DC common mode range of the FCTA (M120E and M121E) is expanded because, with the built-in offsets (generated by the scaled M101E-M105E and M111E-M115E) in RGC102E and RGC103E, the V104E and V105E can get closer to VSS. As noted earlier, this built-in offsets in for example RGC103E is approximately, VGSM111E−VGSM115E=ΔVGS≈VDSM117E≈VT×ln(r/s). Hence, by proper scaling and operating current in M120E-M121E , the inputs of FCTA can span to the negative rail, VSS.
The voltage span of the output of FCTA is also improved. The built in offset in PMOSFETs in RGC101E to RGC103E is chiefly a function of VT and mostly independent of PMOSFET device parameters, such as PMOSFET VTH. Similarly, the built in offset in NMOSFETs in RGC104E and RGC105E is chiefly a function of VT and mostly independent of NMOSFET device parameters, such as NMOSFET VTH. Therefore, the voltages at nodes 104E, 105E, 109E, and 114E are mostly independent of MOSFET device parameters, and track each other given that they are mostly a function of VT. Therefore, maximum VOUT is approximately ≤VDD−VT×ln(r/s)−VDSM118E-sat. Also, minimum VOUT is approximately ≤VSS+VT×ln(r/s)+VDSM119E-sat.
The amplifier is improved in consistency of DC, AC, and transient performance over temperature, power supply, and process variations, in part because the auxiliary amplifier (AAUX) function in each of RGC-CM100F coupled with CSGA100F is made of the same channel FET as the FCTA's CGA and current mirror. The fact that the built-in offset for all RGC-CM (coupled with composite amplifier) is mostly a function of VT also helps reduce systematic mismatches in the FCTA signal path and helps improve FCTA performance over temperature, power supply, and process variations.
In summary, besides providing the extra gain, the benefits of utilizing RGC with composite amplifier in the main amplifier include the following. First, the same type of FETs are used in FCTA's CGA and current mirrors and those used in RGC-CM's AAUX. The build-in offset in RGC-CM is generated by PMOSFET and NMOSFET. However, the build-in offset itself is roughly independent of either PMOSFET and NMOSFET device parameters such as CMOSFET VTH. The build-in offset is mostly a function of VT which is highly predictable and insensitive to process variations. Therefore the FCTA's DC and AC specifications and stability is improved over fabrication process, temperature, and power supply variations. Second, each of the amplifier's RGC101E to RGC105E are self biased (i.e., no need for separate voltages to set its internal bias nodes) saving current and die space. Third, the aforementioned benefits of the same RGC-CM100F coupled with CSGA100F that is utilized in the FCTA in repeated instances, carries over to the FTCA the improvements in small size, low current, higher speed, and wider output voltage span as well as consistency of specification performance over fabrication process, temperature, and power supply variations. Fourth, the DC voltage shift provided in the RGC-CM100F coupled with CSGA100F expands the amplifier's DC common mode input and output ranges closer to the rails. As noted earlier, for example, a 75 mV head-room expansion at the input-output of each of the upper PMOS and the lower NMOS of RGC-CM100F coupled with CSGA100F, can expand the voltage span at the input-output of the amplifier as well as at the high gain (impedance) node of the FCTA by 150 mV or 15%, which is beneficial especially in the sub-1V power supply environment.
Section (VII): Detailed Description of First Embodiment of an Amplifier of
The amplifier of
There are other amplifier configurations that can utilize this FCS. One such example would be a FCTA with a gm control circuit to keep the amplifier's input transconductance (gm) constant across input voltage common mode range (VINCM). Another example would be FCTA that utilizes regulated cascode current mirrors (RGCs) to improve the amplifier's performance, including increasing the gain of the amplifier. Note also that in
The connections of the elements in
One of the reasons for utilizing a FCS (or its equivalent function in an amplifier current mirror network) such as that of a FCTA, is to make the biasing at the summing junction of the FCTA output more insensitive to the common mode voltage swings in order to improve the amplifier's accuracy. The prior art FCS circuit is illustrated in
The steady state operations of the
With regards to node 204A, operating currents IDM202A=IDM204A≈i and IDM203A≈i are held constant, setting aside non-idealities. As the KCL operates on node 204A, an input voltage change (ΔvIN) at the FCTA amplifier applied across M209A-M211A generates a current change (ΔiN) in IDM209A. This in turn would cause the first regulating circuit (containing M204A and current source IM200A) to regulate the gate voltage of M200A which results in the ΔiN to flow into M200A, while IDM200A is ‘current mirrored’ with IDM212A (and scaled). Note that the dynamic response of this current mirror (M200A-M212A) is also improved. This is because the second regulating circuit containing M204A and current source IM200A is configured as a common gate amplifier (CGA), which is very fast, and whose output drives the gate terminals of M200A-M212A.
Similarly, with regards to node 205A, as explained earlier, the operating currents IDM203A=IDM205A≈i and IDM202A≈i are held constant, setting aside non-idealities. As the KCL operates on node 205A, an input voltage change (ΔvIN) at FCTA, applied across M208A-M210A, generates a current change (ΔiP) in IDM208A. This in turn would cause the second regulating circuit (containing M205A and current source IM201A) to regulate the gate voltage of MM201A which would result in the ΔiP to flow into MM201A, while IDM201A is ‘current mirrored’ with IDM213A (and scaled). Note also that the dynamic response of this current mirror (M201A-M213A) is improved. Similarly, this is because the second regulating circuit containing M205A and current source IM201A is configured as a common gate amplifier (CGA), which is inherently fast, and whose output drives the gates terminals of M201A-M213A.
The FCS200A minimum VDD≥VGS2VDS is improved using fewer transistors and less current (versus prior art of
In summary, the FCS200A block that is utilized in the amplifier of
In conclusion, some of the benefits of utilizing FCS200A in an amplifier may include one or more of the following. First, the FCS can operate at lower power supply, using fewer transistors with less current, which is beneficial for the amplifier that utilizes this FCS. Second, the FCS can provides matching between upper and low cascoded current sources which improves amplifier's performance, including lowering its offset and noise. Third, given the regulating circuit of FCS200A is based on common gate amplifier (CGA) configuration, the dynamic response of the FCS is improved which improves the dynamic response of the amplifier that utilizes it.
Section (VIII): Detailed Description of Second Embodiment of an Amplifier of
As noted in the prior sections, it would be possible that there are other amplifier configurations besides FCTA that can utilize this FCS200B. Moreover, note that in
The connections of the elements in
For clarity and consistency with the prior sections, the operations of the FCS200B is described first, independent of that of the amplifier. Thus, IDM208A, and IDM209A are assumed to be zero for the purpose of this segment's description, and also non-idealities such as device mismatches are set aside.
The FCS200B top side utilizes the first regulating circuit made up of M204B, M206B, M216B, M218B, and M220B, and current sources I200B, I202B, I206B, and I208B. In this embodiment, the I200B≈I206B≈I208B≈1i are set as equals constant current sources that bias VGSM204B≈VGSM220B≈VGSM216B. Applying the KVL to the voltage loop containing VGSM202B=VGSM220B−VGSM216B+VGSM204B≈VGSM220B. Therefore, IDM202B≈IDM220≈1i. The first regulating circuit is a current input amplifier, where VGSM206B−VGSM220B+VGSM216B establish the VGM204B. As such, M204B and I200B function like a common gate amplifier (CGAP200B), which is fast. The output of this CGAP200B regulates VGSM200B at the node 208B until the KCL at node 210B is satisfied, which is when IDM200B≈IDM202B+IDM203B+IDM204B+IDM209B.
The second regulating circuit utilized in the bottom side of FCS200B is made up of M205B, M207B, M217B, M219B, and M221B, and current sources I201B, I203B, I207B, and I209B. Similar to the top side, the I201B≈I207B≈I209B≈1i are set as equal constant current sources that bias VGSM205B≈VGSM217B≈VGSM221. Applying the KVL to the voltage loop containing VGSM203B=VGSM221B−VGSM217B+VGSM205B≈VGSM221B. Therefore, IDM203B≈IDM221≈1i. The second regulating circuit is a current input amplifier, where VGSM207B−VGSM221B+VGSM217B establish the VGM205B. As such, M205B and I201B function like a common gate amplifier (CGAN200B), which is fast. The output of this CGAN200B regulates VGSM201B at the node 209B until the KCL at node 211B is satisfied, which is when IDM201B≈IDM203B+IDM202B+IDM205B+IDM208B.
The steady state operations of the
With regards to node 210B, operating currents IDM202B=IDM204B≈i and IDM203B≈i are held constant, setting aside non-idealities. As the KCL operates on node 210B, an input voltage change (ΔvIN) across M209B-M211B generates a current change (ΔiN) in IDM209B that would cause the first regulating circuit (containing M204B and current source IM200B) to regulate the gate voltage of M200B. As a result, ΔiN would flow into M200B, while IDM200B is ‘current mirrored’ with IDM212B (and scaled). Note that the dynamic response of this current mirror (M200B-M212B) is also improved. This is because this first regulating circuit containing M204B and current source IM200B is configured as a common gate amplifier (CGA), which is inherently fast, and whose output drives the gate terminals of the current mirror containing M200B-M212B.
With regards to node 211B, the operating currents IDM203A=IDM205A≈i and IDM202A≈i are held constant, setting aside non-idealities. While the KCL operates on node 211B, the input voltage change (ΔvIN) across M208B-M210B generates a current change (ΔiP) in IDM208B which would cause the second regulating circuit (containing M205B and current source IM201B) to regulate the gate voltage of MM201B. As a result, this ΔiP would flow into MM201B, while IDM201B is ‘current mirrored’ with IDM213B (and scaled). Note also that the dynamic response of this current mirror (M201B-M213B) is improved. This is because this second regulating circuit containing M205B and current source IM201B is configured as a common gate amplifier (CGA), which is very fast, and whose output drives the gates terminals of the current mirror containing M201B-M213B. The FCS200B minimum VDD≥VGS+2VDS is improved compared to the prior art of
In summary, the FCS200B block that is utilized in the amplifier of
In conclusion, some of the benefits of utilizing FCS200B in an amplifier are the following. First, the FCS can operate at lower power supply which frees the amplifier to operate with lower power supply as well. Second, performance of FCS is improved, including its dynamic response, by separating nodes 207B from 201B and separating nodes 206B from 200B, which helps roughly shield the CGAs used in FCS from transients on nodes 200B and 200B. Hence, the transient response of the amplifier, in which the FCS is utilized, can be improved. Third, the FCS can provides some matching between upper and low cascoded current sources which improves amplifier's performance, including lowering its offset and noise of the amplifier in which the FCS is utilized. Fourth, given the regulating circuit of FCS200B is based on common gate amplifier, the dynamic response of the FCS is improved which improves the dynamic response of the amplifier in which it is utilized.
Section (IX): Detailed Description of Third Embodiment of an Amplifier of
As just noted, from a high level functional perspective, the embodiment of FCS200C here is similar to that of FCS200A and FCS200C disclosed in the previous sections. The embodiment of the FCS200C is illustrated in BLOCK: FCS200B (at the left bottom side of
As noted in the prior sections, it is possible that there are other amplifier configurations, besides a FCTA amplifier topology, that can utilize this FCS200C. Moreover, note that in
The connections of the elements in
For clarity and consistency with the prior sections, the operations of the FCS200C is described first, independent of that of the amplifier. Thus, IDM208C, and IDM209C are set to zero and non-idealities such as device mismatches are set aside. The FCS200C utilizes the first regulating circuit, on the top side, made up of M206C, M218C, and M220C, and current sources, I202C, and I208C plus the amplifier AN200C (containing M204C, M266C, M226C, M222C, and current source I200C). Here, AN200C's output regulates VGSM200C until its inputs are substantially equalized, which is when VGSM202C≈VGSM220C that occurs when IDM202C≈IDM220C. Therefore, IDM202C≈IDM220≈1i, and the KCL operating on node 210C would result in IDM200C≈IDM202C+IDM203C+IDM209C.
On the complementary or bottom side, the FCS200C utilizes the second regulating circuit made up of M207C, M219C, and M221C, and current sources, I203C, and I209C plus the amplifier AP200C (containing M205C, M217C, M225C, M223C, and current source I201C). Here, AP200C'S output regulates VGSM201C until its inputs are substantially equalized, which is when VGSM203C≈VGSM221C that occurs when IDM203C≈IDM221C. Therefore, IDM203C≈IDM221≈1i, and the KCL operating on node 211C would result in IDM201C≈IDM203C+IDM202C+IDM208C.
The steady state operations of the
Again while the KCL operates on node 210C, an input voltage change (ΔvIN) across M209C-M211C generates a current change (ΔiN) in IDM209C that would cause the first regulating circuit, containing AN200C, to regulate the gate voltage of M200C. As a result, ΔiN would flow into M200C, while IDM200C is ‘current mirrored’ with IDM212C (and scaled). Similarly, with the KCL operating on node 211C, the input voltage change (ΔvIN) across M208C-M210C generates a current change (ΔiP) in IDM208C that would cause the second regulating circuit containing AP200C, to regulate the gate voltage of MM201C. As a result, this ΔiP would flow into MM201C, while IDM201C is ‘current mirrored’ with IDM213C (and scaled).
The FCS200C minimum VDD≥VGS+2VDS is improved compared to the prior art of
In summary, the FCS200C block that is utilized in the amplifier of
Section (X): Detailed Description of Amplifier Illustrated in
To improve noise the amplifier is narrow banded. To maintain ultra low power consumption, the steady state quiescent current consumption of the amplifier is kept at ultra low levels. To re-invigorate the dynamic response of the narrow banded amplifier back-up, the intermittent (dynamic) current consumption of the amplifier is increased. One way to narrow band the amplifier is to connect a capacitor (Ce) to the high impedance (high gain) node of the amplifier. Given the low current consumption of the amplifier, the added capacitor to narrow band the amplifier, also makes the dynamic response of the amplifier slow. The disclosed method to reinvigorate the narrow banded (for lowering the noise of) amplifier, provides a ‘fast boost-on’ current and ‘slow boost-off’ current helps optimize for faster dynamic response.
The embodiment of
The scale factors for FETs and current sources (e.g., M314A, M316A, I300A, I301A, I302A, and I304A, M306A-M3108A, M304A-M310A,) can be altered depending on factors such as speed, and power consumption goal, amongst others. For example, with 0.01≥b≥1000000, 0.01≥t≥1000000, for the embodiment of
The connections of the elements in
The BLOCK300A contains M300A, M302A (configured in CSA), M304A, M306A, M308A, M310A (configured in minimum current selector, MCS300A), I302A, M312A, I304A, M314A, M316A, Ce300A, and Ce301A (configured in providing the ‘boost on’ and ‘boost off’ signal as well as shaping the ‘fast boost-on’ {e.g., slew} current and ‘slow boost-off’ {e.g., slow decay with one-time constant} current that is fed into the FCTA. The FCTA contains M301A, M303A (input stage configured in CSA similar to that of the BLOCK300A), M313A, M315A, (configured in CGA) M309A, M311A (configured in current mirror), M305A, M307A, M317A, M319A, M321A, M323A, V301A, and I301A (configured in the operating current and bias circuitry network for the FCTA).
Describing the details of the circuit in
All else equal, increasing the operating current ‘i’, speeds it up and vice versa, and increasing the effective capacitance at the high impedance node of the FCTA at node 313A narrows bands and makes its dynamic response slower and vice versa.
An amplifier's noise generally increases when the amplifier operates with low currents. This disclosure reduces the output noise of the amplifier by narrow banding the amplifier at its high impedance node 313A by increasing Ce301A, while keeping the steady state current consumption at ultra low levels. Because narrow banding the amplifier, slows its dynamic response (speed), then the operating current ‘i’ is boosted dynamically, which is triggered when the amplifier input (VIN) stop tracking each other (and go off balance) after (VIN) receive a large transient signal. In this teaching, the dynamic response of the amplifier, generally speaking, goes through two phases: the ‘slewing time’ or ‘slew rate’ phase and the ‘settling time’ phase. The slew rate (SR∝2i/Ce301A) of FCTA is largely determined by the effective capacitance (Ce301A) at the amplifier's high impedance output node 313, and the operating current of the amplifier, ‘i’, that is scaled as in IDM315A and IDM311A. The settling time of the FCTA, that is distinguished from SR for the purpose of this disclosure, is largely dominated by input stage gmM301A, Ce301A, and the amplifier's output impedance that is largely a function of rds∝VA/i. Because both the gm, ‘i’, and Ce of the noise reduction plus speed boost circuit, BLOCK300A, and that of the amplifier, FCTA, are arranged to be a function of similar device parameters on silicon, therefore the dynamic response of FCTA and BLOCK 300A track each other more consistently over process and operating condition variations.
Although the gmM301A of FCTA and gmM300A of FCTA and BLOCK300A at their input stages are similar, but an aspect of this disclosure is that FCTA and BLOCK300A are arranged to respond differently to their input voltage entering and exiting balance (i.e., in and out of steady state conditions). This aspect of the disclosure, that will be described shortly, helps a smoother and more consistent dynamic response in and out of the speed boost phase, over process and operating condition variations.
First, the steady state phase for
First, at a high level, lets describe the operations of the minimum current selector (MCS300A) composed of M304A mirrored and scaled with M310A coupled with M306A mirrored and scaled with M308A. When IDM304A≈0→VGSM304A≈0→VGM310A≈0, and thus the output current of MCS300A which is outputted via IDM310A≈0, thus the min(IDM304A, IDM306A)≈IDM304A≈0 is selected. Conversely, when IDM306A≈0→VGSM306A≈0→VGM308A≈0, and thus the output current of MCS300A which is outputted via IDM310A≈0, thus the min(IDM304A, IDM306A)≈IDM306A≈0 is selected. In the steady state phase there is current equilibrium, when IDM304A≈i→VGSM304A≈VGSi→VGM310A≈VGSi. Similarly in equilibrium, when IDM306A≈i→VGSM306A≈VGSi→VGM308A≈VGSi. Given that M310A is scaled at 2× the size of M304A and M308A is scaled at 2× the size of M306A, and that M310A is in series with M310A, and that VGM308A≈VGM310A≈VGSi, therefore the output current of MCS300A which is outputted via IDM310A≈i. Thus, in the steady state phase when IDM304A≈IDM306A≈i, the FETs in MCS300A, including M304A, M306A, M308A, and M310A are in balance and the MCS300A operates accordingly and min(IDM304A, IDM306A)≈IDM304A≈IDM306A≈i.
Therefore, in steady state given that the output of MCS300A generates i≈IDM310A>I302A≈0.5i, then V310A that controls the boost on-off signal, pulls down (V310A≈VSS) and there is the boost-off phase. As a result, VGM312A≈0→IDM312A≈0→IDM314A≈IDM316A≈0. Thus, in steady state phase, the amplifier's bias current is supplied only by I301A, and with IDM316A≈0 amplifier remains in the ‘boost off’ phase.
In this segment, the ‘boost on’ phase is described. When the boost is inactive, IDM323A≈I301A establishes the ultra low quiescent operating current of the amplifier. While the boost is active, BLOCK300A dynamically generates a boost current that is added to IDM323A to boost the FCAT dynamic response. Here is how this happens. When large differential signal transients are applied at VIN, then all or majority of the I300A current 2i would flow through, for example, M302A. Thus, IDM302A≈IDM306A≈2i and IDM300A≈IDM304A≈0<<2i. Therefore, as noted earlier, then MCS300A operates accordingly and min(IDM304, IDM306)≈IDM304A≈0<<2i. In this phase, MCS300A output (flowing through M310A) conduct near or at zero current, and hence V310A is pulled up by I302A (when V310≈VDD) and there is the boost-on phase. In the boost-on phase, all of I304A≈b×i passes through M312A (that functions like a current switch) onto M314A that is mirrored and scaled with M316A. Hence, IDM316A≈b×t×i, which is the boost-on dynamic current that is added onto the amplifier bias network via M316A to M323A. Note that the operating currents in the input stage of FCTA (M301A-M303A) is boosted via the amplifier's bias network that contains M323A-M317A-M307A-M305A. As noted earlier, boosting the operating current in the amplifier's input stage, increases its ‘i’ and gm and thus speeds up the amplifier's dynamic response. Similarly, the operating currents in the high-impedance high-gain stage of FCTA (M313A-M315A-M319A-M311A) is boosted via the amplifier's bias network that contains M323A-M319A-M321A. As noted earlier, boosting the operating current in the amplifier's high impedance stage, increases its ‘i’ and also speeds up the amplifier's dynamic response, including its slew rate.
Before the dynamics of getting in and out of the current boost phase is described, a clarification with respect to Ce301A and Ce301A is in order. From a high level point of view, this disclosure is flexible in how Ce301A and Ce301A are made. For example, Ce301A and Ce301A can be made of similar devices such as standard capacitors, FET capacitors, or relying on Ce associated with the intrinsic capacitances coupled to the 313A node of amplifier and the 314A node of BLOCK300A. The dynamic response of the FCTA and BLOCK300A are designed to match by using the same type of devices that impact their dynamic response, including using similar Ce and running the amplifier and the noise reduction (plus speed boost) circuits with similar bias currents that track each other over process and operating variations. Matching the dynamic response of FCTA and BLOCK300A helps with smooth transient response in and out of boost plus consistency of dynamic response across process and operating condition variations. The Ce301A is the effective capacitance at the high gain high impedance VOUT node 313A of the amplifier that low pass filter filters the noise. This Ce301A can be made with FETs or regular capacitors available in standard CMOS fabrication processes. Part of the Ce301A can also be contributed by the intrinsic input capacitance associated with, for example, a buffer transistor drivers that could be coupled to the main FCTA's VOUT node 313A. Similarly, the Ce300A is the effective capacitance at node 314A, in BLOCK300A, that contributes to shaping the transient profile of the ‘fast boost-on’ current and ‘slow boost-off’ current that is injected into the amplifier. This Ce300A capacitance can be made with FETs (e.g., PMOSFET capacitors), or regular capacitors available in standard CMOS fabrication processes. Part or all of the effective capacitance, Ce300A, can also be contributed by the intrinsic input capacitance associated with, for example, CeM314A and CeM316A.
Next the dynamics of getting in and out of current boost phase is described. When the boost-on signal is triggered, I304A≈b×i slews the node 314A. The ‘boost-on’ signal at node 310A turns on M312A, which acts like a current switch, which causes I304A to provide a ‘fast boost-on’ current onto M314A that is mirrored and scaled up onto M316A. Note that the speed at which node 314A slews (i.e., the dynamic profile of ‘fast boost-on’ current) is also a function of Ce314A, beside I304A.
For a better perspective about the dynamic response, note that while the ‘fast boost-on’ current is still feeding the FCTA's current bias network, the FCTA's SR remains boosted. At some point when the amplifier's inputs approach balance, BLOCK300A triggers the boost off signal (bringing down V310A to VSS) which shuts off the current switch MM312A. Here, the flow of I304A is closed off, and the amplifier enters the ‘boost-off’ phase when a decaying current continues feeding the FCTA bias network. When the boost off signal is triggered and I304A is cut off from M314A, the profile of ‘slow boost-off’ current is a function of the Ze314A and Ce300A, where Ze314A∝1/gmM314A, which will be described further shortly.
An aspect of this disclosure is that the boost-off signals is generated responding to different levels of inputs equilibrium, although both FCTA and BLOCK300A are receiving and mentoring the same input voltage. The BLOCK300A in this disclosure is arranged such that when its inputs can be coarsely equalized, it triggers the boost-off signal. BLOCK300A can be arranged such that its inputs can be coarsely equalized in different ways. Some example are for BLOCK300A to have lower gain than the main amplifier, or having the MSC300A trigger node 310A when its currents (e.g., M304A and M306A) are coarsely equalized instead of accurately, or inserting a hysteresis in the inputs stage of BLOCK300A, or combination thereof, amongst other means. Another aspect of this disclosure is that after the boost off signal is triggered in BLOCK300A, subsequently there continues to be a slow decay current (thus the term ‘slow boost-off’ current) that feeds the amplifier's operating bias current network, until the slow boost-off current dies off. This arrangement improves the FCTA settling time. The aforementioned attributes (boost off signal triggered when inputs are equalized coarsely, plus the slow-boost off current) improve the amplifier's settling time. As noted earlier, the ‘slow boost-off’ current decay rate is roughly determined by node 314A's time constant set that is set approximately by the equivalent capacitance, Ce314A, and impedance, Ze314A, at node 314A. Note that in this embodiment Ce314A is dominated by effective capacitance CeM314A and CeM316A, and Ze314A is dominated by approximately 1/gmM314A computed in the saturation region, in light of the boosted transition currents from subthreshold to saturation region during the intermittent boosted phase. As stated earlier, this embodiment enables the SR and settling time of both the amplifier and BLOCK300A (which provides and shapes the dynamic response profile of the ‘fast boost on’ and ‘slow boost-off’ currents for the FCTA) to approximately track and match each other. As such, this trait benefits the amplifier with smooth dynamics response (in and out of boost phases) and its consistency over process and operating condition variations.
Comparative simulations in
In conclusion, here is a summary of some of the improvements to an amplifiers performance that can be attained by this disclosure. First the amplifier's output noise is reduced by narrow banding the amplifier. Second, the static operating current consumption of the amplifier is kept ultra low. Third, the slowed dynamic response of the narrow-banded amplifier (both slew rate and settling time) are reinvigorated and boosted. Fourth, lower noise, and most other attributes of the amplifier including its gain, bandwidth, static power consumption, common mode range, PSRR, and CMRR are generally not affected by the boost stage, since boost is only engaged dynamically when inputs experience large differential signals. Fifth, amplifier's input structure and that of the boost stage are substantially similar, and hence the boosting function accommodates the full common mode range and power supply span. Sixth, as noted earlier, the AC, slew rate, and transient profiles of the amplifier and that of the boost circuit of BLOCK300A should approximately track each other over temperature and process variations. This is because generally, the amplifier and the boost stage's operating currents, gain (e.g., ∝VA/VT2), and input's 1/gm (e.g., ∝VT/i) roughly track each other, as do their poles in BLOCK300A and FCTA are roughly a function of similar ∝1/gm, and ∝CeFET (or standard capacitors), and ∝ro≈VA/i given that they are made of similar device parameters for both BLOCK300A and FCTA. Seventh, the switching threshold of the amplifier's input stage from small signal to large signal needs to be large enough (e.g., offset mismatch between the boost and the amplifier input stages, ΔVOFS) so that a false or premature turning on of the boost function is avoided. This is the case considering that in steady state (before boost-on phase), when the VIN imbalance is first detected, the amplifier and boost stage input's 1/gmPMOS where i≈10 s of nano-amperes. Note that the BLOCK300A operating currents remain ultra-low, while the main amplifier currents increase substantially in the boost-on phase to the micro-ampere range, which take the amplifier FETs out of the subthreshold region. This is a transitory change in the gain as well as the dynamic response of the amplifier while in the boost-on phase, compared to BLOCK300A (that stays in subthreshold region) while monitoring and receiving the same input voltage as the amplifier. Note also that, it is possible to provide some hysteresis at the input of BLOCK300A as guard-band against unwanted boost signal toggles. Eighth, to arrange for the boost-off signal to be triggered (setting in motion the boost current decay) when the BLOCK300A inputs are coarsely equalized (as compared to the amplifier's inputs whose inputs continue to converge towards finer balance and finer equalization), the amplifier's settling time is improved. Ninth, the maximum boost current, b×t×i, is fixed and is proportional to amplifier's static bias current, ‘i’, which helps control peak dynamic current consumption. This trait also facilitates the boost stage's peak speed to tracks that of the main amplifier, over process, temperature, and operating variations. Note that ‘i’ can be made independent of VTH and mostly a function of VT and μPMOS which are more tightly controlled in manufacturing, and helps with consistency of performance specification across manufacturing variations. Tenth, care is taken to minimize dependence of amplifier's specifications on multiple device parameters such as NMOSFET, VTH NMOS/PMOS, and N+/P+ resistors. Instead, amplifier's specifications mostly rely on PMOSFETs which dominate pertinent signal paths, and this can help optimize yield and help lower noise (e.g., PMOSFET 1/f noise<<NMOSFET 1/f noise) further.
Section XI: Detailed Description of Amplifier Illustrated in
Similar to the teaching in previous section, the disclosure here is a method of lowering the output noise of an amplifier by band passing it while keeping its static current consumption ultra low, and concurrently speeding up the amplifier's dynamic response, by boosting the amplifier's operating current in the face of the amplifier's inputs receiving a large transient signal. The noise reduction and speed boost circuit of BLOCK300B utilizes a LTA300B to detect when inputs are imbalanced. Similar to
The connections of the elements in
The BLOCK300B contains M300B, M302B, and M302B (configured in CSA), M304B, M304B′, M306B, M306B′, M308B, M310B (configured in Looser Take All, LTA300B), I302B, M312B, I304B, M314B, M316B, Ce300B, and Ce301B (configured in providing the ‘boost on’ and ‘boost off’ signal as well as shaping the ‘fast boost-on’ (e.g., slew) current and ‘slow boost-off’ {e.g., slow decay with one-time constant} current that is fed into the FCTA. The FCTA contains M301B, M303B (input stage configured in CSA similar to that of the BLOCK300B), M313B, M315B, (configured in CGA) M309B, M311B (configured in current mirror), M305B, M307B, M317B, M319B, M321B, M323B, V301B, and I301B (configured in the operating current and bias network circuitry for the FCTA).
Here, description of the operations Loser Take All, LTA300B (which is similar in function to that of prior section regarding MCS300A) is provided, ignoring non-idealities such as mismatches. The boundary conditions are described first. When large signals are applied to VIN and input FETs M300B-M302B-M302B′ become imbalanced. First, the case is described when all of I300B=4×i flows through M300B, and IDM302B≈IDM302B′≈0<<4×i. Therefore, IDM304B≈2×IDM304B′≈4×i. Given that IDM302B≈0 and IDM304B′≈2×i→V306B≈0→VGSM306B≈0→IDM306B≈IDM306B′≈0. Given that IDM302B′≈0 and IDM306B′≈0→IDM308B≈0≈IDM310B. Thus the output current of LTA300B is outputted via IDM310B≈0, thus the loser(IDM300B, IDM302B&B′)≈IDM302B&B′≈0 takes all. Note that IDM302B&B′ denotes IDM302B, IDM302B′ that are equal (having the same VGS) whose sum is 2i during steady state.
Next, considering the case where the large signal VIN causes an imbalance such that all of I300B=4×i flows through M302B and M302B′. Here, IDM300B≈0<<4×i and IDM302B≈IDM302B′≈2×i. Therefore, IDM304B≈2×ID′M304B≈0<<4×i. With IDM302B≈2×i and IDM304B′≈0→IDM306B≈2×i≈IDM306B′. With IDM302B′≈2×i and IDM306B′≈2×i, therefore no current is left for M308B. Thus, ID308B≈0≈IDM310B. Therefore the output current of LTA300B is outputted via IDM310B≈0, thus the loser(IDM300B, IDM302B&B′)≈IDM300B≈0 takes all.
In summary, when there is an imbalance (in either direction) as a result of a large signal VIN, the LTA300B generates a zero current through IDM310B≈0. As such I302B=1×i pulls up on node 310B (V310B≈VDD)→M312B turns on and passes I304B=b×i onto M314B whose current is mirrored and scaled up onto M316B with IDM314≈b×t×i . This IDM314B is the boost-on current that is added to the FCTA quiescent current, I304B, to feed the FCTA current bias network through M323B.
Next considering the steady state, or static, conditions when the amplifier is regulated its input FETs M300B-M302B-M302B′ are in balance. In steady state, IDM300B≈2×IDM302B≈2×IDM302B′≈2×i. Therefore, ID304B≈2×IDM304B′≈2×i and IDM302B≈i. Therefore, there is no current left for M306B→IDM306B≈IDM306B′≈0. With IDM306B′≈0, then all of IDM302B′≈i flows through M308B which is mirrored and scaled up 2× into M310B and thus 2×IDM308B≈IDM310B≈2×i. Thus the output current of LTA300B is outputted via IDM310B≈2×i since there are no loser, but equals per se here, the loser(IDM300B, IDM302B&B′)≈IDM300B≈ID′M302B&B≈2×i takes all. Therefore, when the amplifier's inputs arrive at balance in steady state conditions, the output current of LTA300B generates a IDM310B≈2×i (½ the tail current of the input stage). As such, IDM310B pulls down on I302B=1×i which takes node 310B down (V310B≈VSS)→M312B turns off and blocks I304B=b×i from flowing into M314B. Thus, IDM314B≈0≈IDM316B. As a result, the ultra low static FCTA quiescent current, I304B, feeds M323B in steady state conditions with no boost current.
In summary,
As explained earlier, while reinvigorating the SR and settling time, the price to pay for lowering the amplifier's noise, is the increased dynamic current consumption (intermittent power consumption). After the amplifier's inputs receive a large signal transient, and the boost-on signal is triggered, the ‘fast boost-on’ current kick starts speeding up the amplifier's dynamic response by boosting up its slew rate, which moves the amplifier back on the path towards balancing its inputs ahead of boost-off signal trigger. To save on the intermittent power consumption, the boost off signal can be triggered sooner which cuts off the slewing current boost sooner. However, a ‘slow boost-off’ arrangement continues injecting a dynamic decaying current into the amplifier bias network (which eventually levels off to zero) to speed up the amplifier's settling time. This is accomplished, in part, as explained previously above by arranging BLOCK300B to initiate the boost-off signal when its inputs are not equalized in the same size and fashion as that of the inputs of FCTA (e.g., the amplifier has approached but not fully regulated).
The whole amplifier operates in the subthreshold region when it is un-boosted, but its' input, transconductance gain, and (would be) output buffer stages would become faster by receiving the boosted dynamic current and transition in and out of saturation during the intermittent boosted phases. It is also of note that PMOSFETs can be used at input stages and also as active resistors to set the operating bias currents for the whole amplifier (including, input stages, gain stage, the boost circuit, and output stage), which can establish both their input stage's transconductance (gmPMOS). Moreover, PMOSFET or normal capacitors can be utilized to set the dominant poles of each of the amplifier gain stage and that of the boost stage. In other words, dynamic response of both the FCTA and BLOCK300B can be largely proportional to the same (operating current) ‘i’ and (dominant effective capacitance) ‘Ce’, and same kind of device parameters. Hence, the dynamic response in and out of boost track each other, and follow a reasonably smooth and stable passage, in and out of boost phases. Moreover, the profiles of the amplifier and the boost stage dynamic response can be more consistent from lot-to-lot in manufacturing, across operating variations.
Section (XII): Describing a Prior Art Amplifier Gain Stage Illustrated in
Generally a buffer amplifier, as depicted in
As such, the amplifier gain stage's output node 414A (vo1400A) of
In the next sections of this disclosure, AMP400B, AMP400C, or AMP400D are configured in unity gain. In unity gain configuration, the output of BUF400B which is Vo400B, or output of BUF400C which is Vo400C, or output of BUF400D which is Vo400D connect to the negative input terminal node 412A or vin− of the amplifier gain stage of
Next, considering the AMP400B, or AMP400c, or AMP400D. Note that steady state conditions may be disturbed indirectly as well. For example, a transitory current imbalance, Δi, at one of the current input node of the (inverting) buffer drivers that may be initiated by sink-source current in output FETs. In the presence of such transitory Δi, the closed loop containing the amplifier gain stage (
It would be possible that the buffer driver methodology described in this disclosure would operate with other alternative amplifiers gain stages, beside that of
Section (XIII): Detailed Description of Buffer Driver Comprised of Complementary Non-Inverting Current Mirrors (CNICM), Minimum Current Selector (MCS) and Inverting Current Mirror Amplifier (ICMA), Illustrated in
A buffer driver may be required to sink or source large amounts of current when interfacing with an external load. It would be advantageous to reduce the current consumption, in a buffer driver, that is attributed to the function of monitoring and sensing of output FET driver sink-source currents (i.e. M438B, M435B) that varies depending on the size an external load. One of the benefits of CNICM400B here is to help reduce the buffer driver's load-dependent current consumption. Also, CNICM400B helps the buffer driver with having more consistent performance to specifications that are less dependent on the external load variations.
Throughout the description of
The (inverting) buffer driver or BUF400B is comprised of BLOCK400B which is the output FETs, or FET400B (containing source PMOSFET M438B and sink NMOSFET M435B), plus BLOCK401B which is the inverting current mirror amplifier or ICMA400B (containing PMOSFETs M430B, M432B, M434B, M436B, and current sources I407B, I408B), plus BLOCK402B which is the minimum current selector or MCS400B (containing PMOSFET MM428B, and NMOSFETs M429B, M431B, and M433B), plus BLOCK403B which is the complementary non-inverting current mirror or CNICM400B (containing on PMOSFETs side M420B, M422B, M424B, M426B, and on the NMOSFET side M421B, M423B, M425B, M427B, and current sources I405B and I406B). As noted earlier I405B can be is the functional equivalent of 2 current sources, and is shown as one current source for demonstrative simplicity. There can be I405B1 and I405B2 where upper terminal of I405B1 is connected to VDD and lower terminal of I405B2 is connected to VSS. As such, lower terminal of I405B1 and upper terminal of I405B2 can be the functional equivalence of the lower and upper terminals of I405B. As stated earlier, AMP400B is comprised of the amplifier gain stage of
Note that it would be possible to utilize a ICMA that is a complementary version of ICMA400B (i.e., PMOSFET based or NMOSFET based). It would also be possible to utilize an MCS that is a complementary version of MCS400B (i.e., PMOSFET base or NMOSFETs based). It would also be possible that a plurality of MCS coupled with a plurality of MCSs can be utilized, which are complementary to one another (e.g., NMSOFET based ICMA coupled with primarily PMOSFET based MCS, utilized along with PMSOFET based ICMA coupled with primarily NMOSFET based MCS).
Moreover, it would be possible to design a buffer driver that utilize other equivalent functional implementations that are complementary non-inverting current mirrors (CNICM), minimum current selector (MCS) or inverting current mirror amplifier (ICMA). For example, a loser take all (LTA) can perform the equivalent function of MCS. Another example would be to utilize, a current mirror amplifier (CMA) instead of an non-inverting current mirror amplifier (ICMA) by arranging the proper circuit signal signs. Another example would be to utilize a current amplifier to perform the equivalent function of a ICMA.
In describing the function of CNICM400B, the non-idealities (e.g., offsets and mismatches) are set aside and the boundary conditions are discussed (when M438B is maximally on and M435B is off, and when M438B is off on and M435B is maximally on). After that, CNICM400B's steady state condition is described (when the quiescent currents in M438B and M435B is equal to i×s, where MOSFET scale factors s=k). The M438B current is scaled down (by a factor of ‘s’) and mirrored by M420B. For clarity of description here, the IDM420B is the input, and IDM426B is the output for PMOSFET side of CNICM400B. If IDM438B≈0→IDM420B≈0→all of I405B≈2i flows onto IDM422B≈2i→IDM424B≈2i. Thus, all of I406B≈2i is taken by M424B→IDM426B≈0, which in effect is the (zero scale) non-inverting current mirror of IDM438B≈0. If IDM438B≈imax→IDM420B≈imax×1/s. Assuming that imax×1/s>2i. Note here that the maximum current consumed in IDM420B is ‘2i’ that substantially lowers VDS of M420B, which substantially curbs the load-dependent source current consumption of BUF400B. Thus, all of I405B≈2i is taken by M420B→IDM422B≈0→IDM424B≈0. Thus, all of I406B≈2i flows into M426B→IDM426B≈2i, which in effect is the rectified non-inverting current mirror of IDM438B≈imax. Note that the maximum (full scale) output current possible for M426B is 2i which corresponds to imax of M438B (and its scaled down and mirrored current imax×1/s of M420B). Also note that IDM426B that is monitored by MCS400B (via M428B, M429B, M431B) has its full scale current caped at 2i enabling it to respond more consistently to imax swings imposed on FET400B for sourcing current, for example, for heavy external loads (e.g., imax>>2i×s).
For steady state, let's assume that there is no external load, and that IDM438B≈s×i. Thus, IDM420B≈i→the I405B≈2i is split in half and IDM420B≈IDM422B≈i→IDM424B≈i. Thus, I406B≈2i is also split in half→IDM424B≈IDM426B≈i→IDM426B≈i, which in effect is the scaled down (by 1/s) non-inverting current mirror of IDM438B≈s×i. Similar signal flow applies to the NMOSFET side of CNICM400B, where the M435B current is scaled down (by a factor of ‘k’) and mirrored onto M421B. If IDM435B≈0→IDM421B≈0→all of I405B≈2i flows onto IDM423B≈2i→IDM425B≈2i. Thus, all of I406B≈2i is taken by M425B→IDM427B≈0, which in effect is the (zero scale) non-inverting current mirror of IDM435B≈0. If IDM435B≈imax→IDM421B≈imax×1/k. Let's assume that imax×1/k>2i. Note here that the maximum current consumed in IDM421B is ‘2i’ that substantially lowers VDS of M421B, which substantially curbs the load-dependent sink current consumption of BUF400B. Thus, all of I405B≈2i is taken by M421B→IDM423B≈0→IDM425B≈0. Thus, all of I406B≈2i flows into M427B→IDM427B≈2i, which in effect is the rectified non-inverting current mirror of IDM435B≈imax. Note that the maximum (full scale) output current possible for M427B is 2i which corresponds to imax of M435B (and its scaled down and mirrored current imax×1/k of M421B). Also note that IDM427B that is monitored by MCS400B (via M433B) has its full scale current caped at 2i enabling it to respond more consistently to imax swings imposed on FET400B for sinking current, for example, for heavy external loads (e.g., imax>>2i×k). For steady state, again it is assumed that there is no external load, and that IDM435B≈k×i. Thus, IDM421B≈i→the I405B≈2i is split in half and IDM421B≈IDM423B≈i→IDM425B≈i. Thus, I406B≈2i is also split in half→IDM425B≈IDM427B≈i→IDM427B≈i, which in effect is the scaled down (by 1/k) non-inverting current mirror of IDM435B≈k×i.
In summary, BLOCK403B is CNICM400B which is the complementary non-inverting current mirror. One of the functions of CNICM400B is to monitor and mirror the currents in FET400B, and rectify the full scale (maximal) mirrored currents that feed the MCS400B. Given that the sink-source currents of FET400B can vary significantly, depending on the external load, the scaled down but un-rectified mirroring of FET400B raw currents can also vary a lot, and increase the current consumption of a (inverting) buffer driver. Mirroring and rectifying the sink-source currents of FET400B via CNICM400B, caps the full scale and curbs the maximal mirrored currents (e.g., to ‘2i’) before the sink-source current signals are fed to MCS400B for signal processing.
Therefore, arranging BUF400B comprising CNICM400B coupled with MCS400B coupled with ICMA400B provides the BUF400B with added benefit of lower (load-dependent) sink-source current consumption, as well as more consistent DC, AC, and transient performance, especially under varying external load conditions. As stated earlier, description of BUF400B is provided considering that it is part of AMP400B that is configured in unity gain.
Before describing the BUF400B operating in steady state and over drive (sink-source), the current equilibrium conditions are described at nodes 414A, 414B and 415A, 415B for steady state to persist. At nod 415B in
During steady-state conditions, IDM421B≈IDM427B≈i (with k≈s, and corresponding to IDM435B≈k×i), and IDM420B≈IDM426B≈i (with k≈s, and corresponding to IDM438B≈s×i). Therefore, IDM426B≈IDM428B≈IDM429B≈IDM431B≈i→VGSM427B≈VGSM429B≈VGSM431B. As explained in the previous sections above, with M429B and M427B (both with W/L=1X) operating at current ‘i’ while providing equal gate voltages for M431B and M433B that are in series (both with W/L=2X), causes the operating current of M431B and M433B to also be ‘i’. As such, the MCS400B selects current equality when min(M426B, M427B)≈min(M428B, M433B)≈M433B≈M428B≈i. The IDM433B×i is the output of MCS400B that is fed onto M430B which is the input to ICMA400B. Here, for the current mirror IDM430B≈i≈IDM432B→the current source I407B≈2i is split equally between IDM432B≈i≈IDM434B→the current mirror IDM434B≈IDM436B≈i. The IDM436B≈i is in balance with the current source I408B=i. This is how current equilibrium is held at node 415B, which sustains the controlled quiescent currents in the buffer driver output FETs, IDM438B≈IDM435B≈s×i (assume setting k≈s).
A brief summary is provided here for BUF400B (contained in AMP400B) of
In the sink mode, node 414B and node 415B are pulled up towards VDD, MN435B and MN421B are turned on→IDM421B>>i→initially before current equilibrium IDM427B≈2i. Also, MN438B and MN420B are turned off or are nearly off, initially→IDM426B≈ID428B≈IDM429B≈IDM431B≈0 which would block current flow in IDM433B→MCS400B selects the min(IDM427B, IDM426B)=IDM426B≈0<<i, which starves M433B, and initially→IDM432B≈IDM430B<<i→all of I407B=2i flows into M434B→IDM434B≈IDM436B≈2i at node 415B with I408B=i, which in a current imbalance at node 415B initially, and as such the AMP400B loop kicks to bring back the balance. In order for “conditions of current equilibrium for nodes 414B and 415B” to return, and to simplify the explanation for clarity of description, in
Conversely, in the source mode, when node 414B and node 415B of
In summary, when there is an imbalance between sink-source FETs currents, IDM438B and IDM435B, such imbalance is scaled down, mirrored, and rectified by the complementary non-inverting current mirror (CNICM400B). The outputs of CNICM400B, IDM427B and IDM426B, which are the rectified version of sink-source currents, are then fed onto the MCS400B. The MCS400B effectively selects the (scaled) minimum current between the output drive currents IDM438B and IDM435B. The selected minimum current, at the output of MCS400B, is then inputted onto ICMA400B which, together with the operations of amplifier's gain stage, regulates a minimum quiescent current in the inactive FET (e.g. for s=k, when IDM435B is sinking current for an external load→inactive source FET IDM438B=s×i, and when IDM438B is sourcing current for an external load→inactive sink FET IDM435B=k×i).
In conclusion, as noted in the previous section, the benefits of the
Section (XIV): Detailed Description of Buffer Driver Comprised of Complementary Non-Inverting Current Mirrors (CNICM), Loser Take All (LTA) and Current Mirror Amplifier (CMA), Illustrated in
Throughout the description of
The connections of the elements in
The buffer driver (e.g., inverting buffer drive) or BUF400C is comprised of BLOCK400C which is the output FETs, or FET400C (containing source PMOSFET M432C and sink NMOSFET M439C); plus BLOCK401C which is the inverting current mirror amplifier or CMA400C (containing NMOSFETs M435C, and M437C, and current sources I407C); plus BLOCK402C which is the loser take all or LTA400C (containing PMOSFET M428C, and M430C, and NMOSFETs M429C, M431C, and M433C); plus BLOCK403C which is the complementary non-inverting current mirror or CNICM400C (containing on PMOSFETs side M420C, M422C, M424C, M426C, and on the NMOSFET side M421C, M423C, M425C, M427C, and current sources I405C and I406C). As stated earlier, AMP400C includes an amplifier gain stage of
The detailed description of the CNICM400C, LTA400C, CMA400C were provided in previous sections, and their inner working as part of AMP400C loop will be be briefly discussed in this section. For clarity of description, non-idealities (e.g., offsets or mismatches) are set aside in these discussions. Before describing the BUF400C operating in steady state conditions and over drive (sink-source), the current equilibrium conditions are described at nodes 414A, 414C and 415A, 415C in order for the steady state conditions to persist. At node 415C in
As explained in the previous section, BLOCK403C is CNICM400C which is the complementary non-inverting current mirror, and one of its function is to monitor by mirroring the FET400C currents and rectify them before feeding the LTA400C for signal processing. Mirroring and rectifying the sink-source currents of FET400C via CNICM400C, curbs the maximal mirrored currents (e.g., to ‘2i’) before they are fed to LTA400C. Setting k=s, for steady state conditions (with no external load), it is assumed that IDM439C≈k×i≈IDM432C≈s×i→IDM421C≈i≈IDM420C→the I405C≈2i is split in half→IDM421C≈IDM423C≈i≈IDM425C and IDM420C≈IDM422C≈i≈IDM424C→the I406C≈2i is split in half→IDM424C≈i≈IDM426C≈IDM428C≈IDM430C, and IDM425C≈i≈IDM427C≈IDM429C. The IDM426C and IDM427C represent the inputs and IDM435C represents the output of LTA400C. As indicated before, one of the tasks of minimum current selector (MCS) or its functional equivalent LTA400C (in the embodiment of BUF400C) here is to monitor the sink-source signals, in this case through CNICM400C that provides IDM426C and IDM427C which are then fed onto LTA400C that outputs the current IDM435C. In steady state, with IDM428C≈i≈IDM429C→at node 420C, IDM428C-IDM429C≈IDM431C≈0≈IDM433C there is no current left for M431C→IDM431C≈0≈IDM433C→at node 421C, with IDM430C≈i, and ID430C-IDM433C≈IDM435C≈i. In summary, in steady state, LTA400C responds to current equality with LTA (IDM426C, IDM427C)≈IDM426C≈IDM427C≈i, which is fed onto the input of ICMA400C via M435C. The functions of the CMA400C, as contained in the AMP400C, is to help regulate the steady-state current in FET400C. Accordingly, IDM435C≈i→IDM437C≈i≈I407C. Here, AMP400C loop is in steady-state, and node 414C and 415C currents are balanced with (IDM417A≈IDM419A)≈(IDM416A≈IDM418A) and I407C≈IDM437C.
Next will be described the sink conditions. In the sink mode, node 414C and node 415C are pulled up towards VDD, MN439C and MN421C are turned on→IDM421C>>i→initially before current equilibrium IDM427C≈2i, which feeds one of LTA400C inputs. Also, M432C and M420Care turned off or are nearly off, initially →IDM426C≈0 which feeds the other input of LTA400C→IDM426C≈IDM428C≈IDM430C≈0 and IDM427C≈2i≈IDM429C, which pulls down on the voltage at node 420C towards VSS→IDM431C≈0≈IDM433C→with IDM430C≈0≈IDM433C, then IDM430C-IDC433C≈0 is the net current output of the LTA400C that is fed onto M435C, as the input of CMA400C. In summary so far, before equilibrium, the LTA (IDM426C, IDM427C)≈IDM426C≈0 that is fed onto CMA400C via M435C→IDM435C≈IDM437C≈0<I407C=i at node 415C, which is a current imbalance, initially→AMP400C loop kicks to bring back the balance. In order for “conditions of current equilibrium for nodes 414C and 415C” to return here, and to simplify the explanation for clarity of description, in
Conversely, in the source mode, when node 414C and node 415C of
In summary, when there is an imbalance between sink-source FETs currents, IDM432C and IDM439C, such imbalance is scaled down, mirrored, and rectified by the complementary non-inverting current mirror (CNICM400B). The outputs of CNICM400C, IDM427C and IDM426C, which are the rectified version of sink-source currents, are then fed onto the LTA400C. The LTA400C effectively performs the equivalent function of selecting the (scaled) minimum current between the output drive currents IDM432C and IDM439C. The selected minimum current, at the output of LTA400C, is then inputted onto CMA400C which, together with the operations of amplifier's gain stage inside the AMP400C loop, regulates a minimum quiescent current in the inactive FET (e.g. for s=k, when IDM439C is sinking current for an external load→inactive source FET IDM432C=s×i, and when IDM432C is sourcing current for an external load→inactive sink FET IDM439C=k×i).
In conclusion, as noted in the previous section, the benefits of the
Section (XV): Detailed Description of Buffer Driver (BUF) Comprised of Minimum Current Selector (MCS) and Inverting Current Mirror Amplifier (ICMA), Illustrated in
While the buffer drivers sinks or source large currents for an external load, for continued proper performance including fast dynamic response, low cross over distortion, and low output impedance, buffer drivers generally run a minimum (well controlled) quiescent current in their inactive output FET drivers. In this embodiment, the MCS400D sense the BUF400D's output FET driver's (M426D-M431D) current signals and feed a proportional signal into ICMA400D. While the output FET is sinking current, the MCS400D coupled with ICMA400D, and in concert with AMP400D loop, regulate the minimum current in the (inactive) source output FET. Conversely, while the output FET is sourcing current, the MCS400D coupled with ICMA400D, and in concert with AMP400D loop, regulate the minimum current in the (inactive) sink output FET.
Throughout the description of
The buffer driver (e.g., inverting buffer driver) is comprised of BLOCK400D which is the output FETs, or FET400D (containing source PMOSFET M426D and sink NMOSFET M431D), BLOCK401D which is the non-inverting current mirror amplifier or ICMA400D (containing NMOSFETs M423D, M425D, M427D, M429D, and current sources I407D, and I408D), and BLOCK402D which is the minimum current selector or MCS400D (containing NMOSFET M421D, and PMOSFETs M420D, M422D,and M424D). As stated earlier, AMP400D is comprised of amplifier gain stage of
Note that it would be obvious for one skilled in the art to utilize a ICMA that is complementary version of ICMA400D (i.e., PMOSFET based or NMOSFET based). It would also be obvious for one skilled in the art to utilize an ICMA that is a complementary version of MCS400D (i.e., PMOSFET base or NMOSFETs based). Additionally, it would be obvious for one skilled in the art that plurality of ICMAs coupled with plurality of MCSs can be utilized, which are complementary to one another (e.g., NMSOFET based ICMA coupled with primarily PMOSFET based MCS, utilized along with PMSOFET based ICMA coupled with primarily NMOSFET based MCS). Moreover, it would be obvious for one skilled in the art to design a buffer driver that utilizes other equivalent functional implementations that are minimum current selector (MCS) or non-inverting current mirror amplifier (ICMA). For example, a loser take all (LTA) can perform the equivalent function of MCS. Another example would be to utilize, a current mirror amplifier (CMA) instead of an inverting current mirror amplifier (ICMA) by arranging the circuit signal signs. Another example would be to utilize a current amplifier to perform the equivalent function of a ICMA.
Describing the details of the circuit in
Before describing the BUF400D operating in over drive (sink-source modes) and steady state conditions, there is described the current equilibrium conditions at nodes 414 and 415 for steady state to hold. At node 414D in
In the sink mode, when node 414D and node 415D are pulled up towards VDD, MN431D (i.e., sinking current for the external load) and MN421D are turned on hard that cause IDM421D≈IDM420D≈IDM422D>>i. Here, M426D and M424D turn off (initially) and IDM424D<<i, which would block IDM422D. Thus, the output current of MCS400D selects the min(IDM421D, IDM424D)=IDM424D<<i, which flows onto M423D, and initially→IDM423D=IDM425D<<i→all of I407D=2i flows into M427D→IDM427D≈IDM429D=2i. For “conditions of current equilibrium for nodes 414D and 415D” to hold, node 414D gets regulated by being pulled down in part by IDM429D, that is initially before equilibrium runs at ‘2i’, until I408D=i≈IDM429D. In summary, in order for this current equilibrium to persist, ICMA400D contained within AMP400D loop regulate node 414D by lowering node 414D voltage sufficiently enough, which is when I408D=i≈IDM429D→IDM424D≈IDM423D≈IDM425D≈IDM427D≈IDM429D≈i≈(1/s)×IDM426D. As such, for BUF400D in current ‘sink mode’ when IDM431D>>IDM426D, the inactive source FET, M426D, is regulated to run at a controlled current ID426D≈s×i . Note that ‘i’ is a constant current source set by the main bias network of the amplifier, and ‘s’ is FET aspect ratio that is tightly controllable.
Conversely, in the source mode, when node 414D and 415D of
Again, ignoring non-idealities, here the mechanism for the BUF400D operating in steady state is described, when there is no output load. At steady state conditions, I408D≈IDM429D at node 414D in
In summary, a near rail-to-rail input-output (i.e., inverting) buffer driver is proposed utilizing a novel current mode output stage comprising of minimum current selectors (MCS) and non-inverting current mirror amplifiers (ICMA) which are fast and operate at low VDD. The preceding gain amplifier stage high-impedance high-gain node, dominates in setting the AC and transient response, with minimal speed degradation caused by the fast buffer driver. Such is the case in part because of the (inverting) buffer driver wide bandwidth since it operates chiefly in current mode makes its fast inherently. The (inverting) buffer driver is based on the main-stream, standard, low cost, and rugged standard digital CMOS manufacturing platform.
In conclusion, the benefits of the
The definitions of the words or elements of the claims shall include not only the combination of elements which are literally set forth, but all equivalent structure, material or acts for performing substantially the same function in substantially the same way to obtain substantially the same result.
All references, including publications, patent applications, patents and website content cited herein are hereby incorporated by reference to the same extent as if each reference were individually and specifically indicated to be incorporated by reference and was set forth in its entirety herein.
The words used in this specification to describe the invention and its various embodiments are to be understood not only in the sense of their commonly defined meanings, but to include by special definition in this specification any structure, material or acts beyond the scope of the commonly defined meanings. Thus if an element can be understood in the context of this specification as including more than one meaning, then its use in a claim must be understood as being generic to all possible meanings supported by the specification and by the word itself.
Recitation of ranges of values herein are merely intended to serve as a shorthand method of referring individually to each separate value falling within the range, unless otherwise indicated herein, and each separate value is incorporated into the specification as if it were individually recited herein. Therefore, any given numerical range shall include whole and fractions of numbers within the range. For example, the range “1 to 10” shall be interpreted to specifically include whole numbers between 1 and 10 (e.g., 1, 2, 3, . . . 9) and non-whole numbers (e.g., 1.1, 1.2, . . . 1.9).
Neither the Title (set forth at the beginning of the first page of the present application) nor the Abstract (set forth at the end of the present application) is to be taken as limiting in any way as the scope of the disclosed invention(s). The title of the present application and headings of sections provided in the present application are for convenience only, and are not to be taken as limiting the disclosure in any way.
This application claims the priority benefit of U.S. Provisional Patent Application Ser. No. 62/304,373, filed Mar. 7, 2016 and entitled “Class AB amplifier low power and fast”; and U.S. Provisional Patent Application Ser. No. 62/320,512 filed Apr. 9, 2016 and entitled “Class AB amplifier high gain”; and U.S. Provisional Patent Application Ser. No. 62/415,496 filed Oct. 13, 2016 and entitled “Class AB Amplifier Low Noise”; Each of the above applications are herein specifically incorporated by reference in their entirety.
| Number | Name | Date | Kind |
|---|---|---|---|
| 4677369 | Bowers et al. | Jun 1987 | A |
| 5283579 | Tasdighi et al. | Feb 1994 | A |
| 5294927 | Levinson et al. | Mar 1994 | A |
| 5640084 | Tero et al. | Jun 1997 | A |
| 5668710 | Caliboso et al. | Sep 1997 | A |
| 5734260 | Tasdighi et al. | Mar 1998 | A |
| 5734292 | Tasdighi et al. | Mar 1998 | A |
| 5814995 | Tasdighi | Sep 1998 | A |
| 5923208 | Tasdighi et al. | Jul 1999 | A |
| 6005374 | Tasdighi | Dec 1999 | A |
| 6054823 | Collings et al. | Apr 2000 | A |
| 6122284 | Tasdighi et al. | Sep 2000 | A |
| 6281743 | Doyle | Aug 2001 | B1 |
| 6366071 | Yu | Apr 2002 | B1 |
| 6384683 | Lin | May 2002 | B1 |
| 6489831 | Matranga | Dec 2002 | B1 |
| 6657480 | Ochi | Dec 2003 | B2 |
| 6885178 | Marinca | Apr 2005 | B2 |
| 7211993 | Marinca | May 2007 | B2 |
| 7612606 | Marinca | Nov 2009 | B2 |
| 20020093325 | Ju | Jul 2002 | A1 |
| 20040124822 | Marinca | Jul 2004 | A1 |
| 20050264345 | Ker | Dec 2005 | A1 |
| 20050285666 | Garlapati | Dec 2005 | A1 |
| 20060224662 | Haiplik | Sep 2006 | A1 |
| 20070052473 | McLeod | Mar 2007 | A1 |
| 20070076483 | Colman | Apr 2007 | A1 |
| 20070132506 | Fujisawa | Jun 2007 | A1 |
| 20070200620 | Iriguchi | Aug 2007 | A1 |
| 20080018316 | Chang | Jan 2008 | A1 |
| 20080278137 | Harvey | Nov 2008 | A1 |
| 20090243711 | Marinca | Oct 2009 | A1 |
| 20100164467 | Jo | Jul 2010 | A1 |
| 20100225384 | Hirose | Sep 2010 | A1 |
| 20100237926 | Kikuta | Sep 2010 | A1 |
| 20110025666 | Baek | Feb 2011 | A1 |
| 20110068767 | Carper | Mar 2011 | A1 |
| 20110102049 | Conte | May 2011 | A1 |
| 20110205193 | Nishimura | Aug 2011 | A1 |
| 20120176186 | Chen | Jul 2012 | A1 |
| 20150160680 | Marinca | Jun 2015 | A1 |
| Entry |
|---|
| Sansen, Willy M., “A CMOS Temperature-Compensated Current Reference”, IEEE Journal of Solid-State Circuits, Jun. 1988,p. 821, vol. 23. |
| De La Plaza, Alejandro,“Nanoampere Supply Independent Low-Voltage Current Reference”, Faible Tension Faible Communication, May 2011,p. 9. |
| Fayomi, Christian, “Sub 1V CMOS bandage reference design techniques a survey”, Analog Integrated Cir Sig Process, Feb 2010, p. 141. |
| Vittoz, Eric, “A Low-Voltage CMOS Bandgap Reference”, IEEE Journal of Solid-State Circuits, Jun. 1979, vol. SC-14, No. 3, Feb. 2010, p. 141. |
| Galeano, Edgar,“A Very Low Power Area Efficient CMOS Only Bandgap Reference”, Integrated Circuits and Sys Design, Aug. 2012, p. 1. |
| Osaki, Yuji, “A 95-nA, 523 ppm/C, .6-uW CMOS Current Reference Circuit w/ Subthreshold MOS Resistor Ladder”, ASPDAC '11, Jan. 2011, p. 113. |
| Ueno, Ken, “A 1-uW 600-ppm/C Current Ref Circuit Consisting of Subthreshold CMOS Circuits”, IEEE Transaction on Circuits, Sep. 2010, p. 681. |
| De Vita, “A 109 nW, 44 ppm/C CMOS Current Reference with Low Sensitivity to Process Variations”, Circuits and Systems, Feb. 2007, p. 3804. |
| Ledesma, F., “Comparison of New and Conventional Low Voltage Current Mirrors”,Circuits and Systems, Aug. 2003, p. 49-52. |
| Hihose, Tetsuya, “A Nano-Ampere Current Reference Circuit and Its Temperature Dependence,” Proceedings of the EECIRC, Oct. 2010, p. 114. |
| Camarcho-Galeano, Edgar, “A 2-nW 1.1V Self-Biased Current Reference in CMOS Technology”, On Circuits and Systems, Feb. 2005,p. 62. |
| Camarcho-Galeano, Edgar,“Temperature Performance of Sub-1V Ultra-Low Power Current Sources”, Circuits and Systems, May 2008, p. 2230. |
| Osaki,“Nano-Ampere CMOS Current Reference with Little Temp. Dependence Using Small Offset Voltage”, Proc. at 53rd IEEE Intl, Aug. 2010, p. 668. |
| Gallegos, S.,“A 128.7nW Neural Amplifier and Gm-C Filter for EEG, Using gm/ID Methodology..”, Circuits and Systems, Aug. 2014, p. 876. |
| Cordova, David,“A Low-Voltage Current Reference with High Immunity to EMI”, Integrated Circuits and Systems Design, Sep. 2014, p. 1-6. |
| Annema, Anne-Johan,“Low-Power Bandgap Heferences featuring DTMOST's”, IEEE Journal of Solid-State Circuits, Jul. 1999, p. 949. |
| Buck, Ame, “A CMOS Bandgap Reference without Resistors”, Solid-State Circuits Conference, Dec. 2000, p. 442-443. |
| Chouhan, Shailesh, “A moditied CMOS Nano-Power Resistorless Current Reference Circuit”, Microelectronics and Electronics, Jun. 2014,p. 1. |
| Guiges, Fabrice, “Sub-1V Oguey's Current Reference Without Resistance”, Circuits and Systems, Dec. 2006, pp. 288-291. |
| Guistolisi,G.,“A Low-Voltage Low Power Voltage Reference Based on Subthreshold MOSFETS”, Journal of Solid-State Circuits, Jan. 2003, p. 151. |
| Hirose, T., “A CMOS Bandgap and Sub-Bandgap Voltage Reference Circuits for Nanowatt Power LSIs”, IEEE Asian Solid-State, Nov. 2010, p. 77. |
| Kussener, F, “50 nA, 1V Nanowatt Resistor-Free Compact CMOS Current References”, Electronics, Circuits and Systems, Dec. 2010, p. 1212. |
| Oguey, H., “CMOS Current Reference Without Reference”, IEEE Journal of Solid-State Circuits, Jul. 1997, p. 1132, vol. 32. |
| Serra-Graelis, F.,“Sub-1V CMOS Proportional-to-Absolute Temp. References”, Journal of Solid-State Circuits, Jan. 2003, p. 84. |
| Ueno,Ken,“A 300 nW,15 ppm/C,20 ppm/V CMOS Voltage Ref Circuit . . . ”, Journal of Solid-State Circuits, Jun. 2009, p. 2097. |
| VanPetegham, Peter, “On a New CMOS Resistor Implementation for Linear Analog IC Signal Processing Applications,” IEEE, Jun. 1998, p. 163-166. |
| Vittoz, Eric, “CMOS Analog ICs Based on Weak Inversion Operation”, Journal of Solid-State Circuits, Jun. 1977, p. 224. |
| Wilson,G., “Novel Voltage-Controlled Grounded Resistor”, Electronic Letters, Dec. 1989, pp. 1725-1726. |
| Huang, Zhangcai, “A CMOS Sub-1-V Nanopower Current and Voltage Ref with Leakage Compensation”, Circuits and Systems, Jul. 2010, p. 4069. |
| Columbo, Dalton, “A CMOS 25.3 ppm/C Bandgap Voltage Ref using Self-Cascode Composite Transistor”, Circuits and Systems, Apr. 2012, p. 1. |
| Colombo, Dalton, “A CMOS 25.3 ppm/C Banagap Voltage Ref Using Self-Cascade Composite Transistor”, Circuits and Systems, Dec. 2012, p. 1. |
| Number | Date | Country | |
|---|---|---|---|
| 62304373 | Mar 2016 | US | |
| 62320512 | Apr 2016 | US | |
| 62415496 | Oct 2016 | US |
