Variable resonator, variable bandwidth filter, and electric circuit device

TECHNICAL FIELD

The present invention relates to variable resonator, variable bandwidth filter and electric circuits using the same.

BACKGROUND ART

In the field of radio communications using high frequencies, signals having specific frequencies are extracted from a number of signals, so that necessary signals and unnecessary signals are separated from each other. Circuits having such a function are called filters and are installed in various radio communication devices.

Generally, filters have invariable bandwidths as design parameters. When using various frequency bandwidths in radio communication devices using such filters, it may easily occur that a plurality of filters are prepared for those bandwidths to be used and are switched by switches and so on. This method requires filters as many as required number of bandwidths and thus increases the scale of the circuit, resulting in a large device size. Further, such devices cannot be operated at frequencies other than frequencies having the frequency characteristics of prepared filters.

In order to solve this problem, in Patent literature 1, a piezoelectric element is used for a resonator composing a filter and the frequency characteristics of the piezoelectric element are changed by applying a bias voltage to the piezoelectric element from the outside, so that the bandwidth is changed.

Patent literature 1: Japanese Patent Application Laid-Open No. 2004-7352

Although the variable filter disclosed in Patent literature 1 is formed as a ladder filter to provide a certain bandwidth, a change in the center frequency is as small as under 1%, due to restrictions imposed by the characteristics of the piezoelectric element, allowing change in the bandwidth to a similar extent, so that the bandwidth cannot be largely changed.

DISCLOSURE OF THE INVENTION

In view of these circumstances, an object of the present invention is to provide a variable resonator, a variable bandwidth filter, and an electric circuit device which can largely change a bandwidth.

In order to solve this problem, a variable resonator according to a first aspect of the present invention is configured as follows: the variable resonator includes a ring-shaped conductor line provided on a dielectric substrate and having a circumferential length of one or an integral multiple of a wavelength at a resonance frequency, and two or more first circuit switches, wherein the first circuit switches have one ends electrically connected to different portions on the ring-shaped conductor line and the other ends electrically connected to a ground conductor formed on the dielectric substrate, and can switch electrical connection/disconnection between the ground conductor and ring-shaped conductor line.

With this configuration, a bandwidth around the resonance frequency can be largely changed by switching the circuit switches to be electrically connected.

The ground conductor and the other end of the circuit switch electrically connected to the ground conductor may be electrically connected to each other via a passive element.

The passive element includes, for example, a resistor, a variable resistor, a capacitor, a variable capacitor, an inductor, and a variable inductor.

In this variable resonator, the loss of a signal at the resonance frequency is mainly contributed by conductor lines composing the variable resonator, and the influence of an insertion loss caused by the circuit switch and so on is small. Thus the configuration can include the passive element.

When such a passive element is provided, a switch may be provided to switch electrical connection between the ground conductor and the ring-shaped conductor line either via the passive element or directly.

A variable resonator according to a second aspect of the present invention includes a ring-shaped conductor line provided on a dielectric substrate and having a circumferential length of one or an integral multiple of a wavelength at a resonance frequency, and two or more first circuit switches, wherein the first circuit switches have one ends electrically connected to different portions on the ring-shaped conductor line and the other ends electrically connected to a transmission line formed on the dielectric substrate, and can switch electrical connection/disconnection to the ring-shaped conductor line.

When the variable resonator according to the first or second aspect is used for, for example, a variable bandwidth filter provided mainly to allow the passage of a signal having a desired frequency, the circuit switch is not provided on the ring-shaped conductor line at the connecting portion of the transmission line or a position of a half wavelength or integral multiple thereof at the resonance frequency from the connecting portion. Even if the circuit switches are provided on these positions, a signal cannot be derived therefrom. The reason will be described later.

The ring-shaped conductor line may be closed by combining a plurality of conductor lines having different line widths. The ring-shaped conductor line enabling selection of different characteristics may be formed by providing a first conductor line, a plurality of second conductor lines having different characteristics, and a second circuit switch which electrically connects the first conductor line and selected one of the second conductor lines to form a closed path.

Further, the first variable resonator according to the first or second aspect and the second variable resonator according to the first or second aspect may be electrically connected to each other via the second circuit switch, and the second variable resonator may be disposed inside the ring-shaped conductor line of the first variable resonator.

In this configuration, the first variable resonator and the second variable resonator are connected to two different positions via the two second circuit switches. Relative to the connecting position of one of the second circuit switches, the other second circuit switch is disposed on the position of a half wavelength or integral multiple thereof at the resonance frequency of the first variable resonator on the ring-shaped conductor line of the first variable resonator, and is disposed on a position at a half wavelength or integral multiple thereof at the resonance frequency of the second variable resonator on the ring-shaped conductor line of the second variable resonator.

In order to solve the problem, the variable bandwidth filter according to a third aspect of the present invention is configured as follows: the variable bandwidth filter includes at least one variable resonator according to the first aspect and an input/output line, wherein the variable resonator and the input/output line are electrically connected to each other.

By using the variable resonator, the passband width can be largely changed.

Moreover, the at least one variable resonator may be connected in parallel to the input/output line on the connecting portion. Further, the at least two variable resonators may be connected in parallel to the input/output line on the connecting portion. The second circuit switches capable of switching electrical connection/disconnection between the input/output line and the variable resonators may be provided on the connecting portions. All or some of the variable resonators may be electrically connected to the input/output line by selecting the second circuit switches.

Alternatively, the at least one variable resonator may be connected in series with the input/output line on the two connecting portions. The two connecting portions are each disposed on the position of a half wavelength or integral multiple thereof at the resonance frequency of the variable resonator on the ring-shaped conductor line of the variable resonator, and the circuit switches may not be connected to the connecting portions.

In order to solve the problem, an electric circuit device according to a fourth aspect of the present invention is configured as follows: The electric circuit device includes the variable resonator according to the first or second aspect, a first input/output line, and a second input/output line, wherein the end of the second input/output line is connected to the connecting portion of the end of the first input/output line and the ring-shaped conductor line of the variable resonator, the first input/output line, the second input/output line, and the ring-shaped conductor line are electrically connected to one another, and on the connecting portion, the end of the first input/output line and the end of the second input/output line are disposed on different planes.

Alternatively, the electric circuit device may include a variable resonator according to the first or second aspect and an input/output line having a bent portion, and the bent portion of the input/output line and the ring-shaped conductor line of the variable resonator may be electrically connected to each other.

Further, the ring-shaped conductor line of the variable resonator may be combined with the input/output line to form an angle on and near a portion where the bent portion of the input/output line and the ring-shaped conductor line of the variable resonator are electrically connected to each other.

EFFECTS OF THE INVENTION

According to the present invention, a given circuit switch is selected from a plurality of circuit switches and is turned on (electrically connected) and thus it is possible to largely change a bandwidth while keeping a resonance frequency constant.

Further, in the variable resonator of the present invention, the loss of a signal at the resonance frequency is mainly controlled by a conductor line composing the variable resonator, thereby reducing the influence of an insertion loss caused by a circuit switch and so on. For this reason, even when a filter is configured using a circuit switch having a large loss for the variable resonator, it is possible to reduce the loss of the passband of a signal.

Further, in an electric circuit device of the present invention, by using the variable resonator of the present invention, it is possible to largely change a bandwidth around the resonance frequency and suppress an insertion loss caused by connecting the variable resonator.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a plan view showing a variable resonator according to an embodiment of the present invention;

FIG. 1B is a plan view showing a variable resonator according to another embodiment;

FIG. 1C is a sectional view showing a switch of the variable resonator; FIG. 2A is a circuit diagram for electromagnetic field simulations, showing the characteristics of the variable resonator;

FIG. 2B is a circuit diagram for electromagnetic field simulations, showing the characteristics of the variable resonator;

FIG. 3A is a graph showing the frequency characteristics of the circuit of FIG. 2A through electromagnetic field simulations;

FIG. 3B is a graph showing the frequency characteristics of the circuit of FIG. 2B through electromagnetic field simulations;

FIG. 4A shows a lossless transmission line model of the circuits shown in FIGS. 2A and 2B;

FIG. 4B is a plan view showing the variable resonator; FIG. 5A shows an embodiment of a variable bandwidth filter using two variable resonators;

FIG. 5B shows another embodiment of the variable bandwidth filter using the two variable resonators;

FIG. 6A is a graph showing the frequency characteristics of the variable bandwidth filter shown in FIG. 5A;

FIG. 6B is a graph showing the frequency characteristics of the variable bandwidth filter shown in FIG. 5B;

FIG. 7A is a graph showing the frequency characteristics of the variable bandwidth filter shown in FIG. 5A;

FIG. 7B shows a variable bandwidth filter in which resistors are disposed between switches and a ground conductor;

FIG. 7C is a graph showing the frequency characteristics of the variable bandwidth filter shown in FIG. 7B;

FIG. 7D shows a variable bandwidth filter using switches for switching over connection to a ground conductor via a resistor and direct connection to a ground conductor;

FIG. 8 shows an embodiment of a variable bandwidth filter configured by connecting two variable resonators in parallel;

FIG. 9 shows an embodiment of a variable bandwidth filter in electric field coupling;

FIG. 10 shows an embodiment of a variable bandwidth filter in magnetic field coupling;

FIG. 11A shows an embodiment of a variable bandwidth filter using variable resonators having different characteristic impedances at different resonance frequencies;

FIG. 11B shows another embodiment of a variable bandwidth filter using variable resonators having the same characteristic impedance at the same resonance frequency;

FIG. 11C shows still another embodiment of a variable bandwidth filter using variable resonators having different characteristic impedance at the same resonance frequency;

FIG. 12A shows the frequency characteristics of the variable bandwidth filter shown in FIG. 11B, where one of the switches is turned on;

FIG. 12B shows the frequency characteristics of the variable bandwidth filter shown in FIG. 11B, where both of the switches are turned on;

FIG. 12C shows the frequency characteristics of the variable bandwidth filter shown in FIG. 11B, where the characteristic impedances of the variable resonators are respectively set at twice and a half that of an input/output line, one switch is turned off, and the other switch is turned on;

FIG. 13 shows an embodiment of a variable bandwidth filter configured by inserting a variable resonator in series with an input/output line;

FIG. 14 is a graph showing the frequency characteristics of the variable bandwidth filter shown in FIG. 13;

FIG. 15 shows an embodiment of a variable bandwidth filter configured by inserting two variable resonators in series with an input/output line;

FIG. 16 shows an embodiment of a variable bandwidth filter configured by inserting one variable resonator in series with an input/output line and another variable resonator in parallel with the input/output line;

FIG. 17 shows an example of a bias circuit using a variable resonator;

FIG. 18 shows an embodiment of a variable resonator using a ring-shaped line which is formed into an ellipse;

FIG. 19 shows an embodiment of a variable resonator using a ring-shaped line which is formed into an arc;

FIG. 20A shows a connection structure of a variable resonator having a circular ring-shaped line and a transmission line;

FIG. 20B shows a connection structure of a variable resonator having an oval ring-shaped line and a transmission line;

FIG. 21A shows a connection structure of a variable resonator and transmission lines in a five-layer structure;

FIG. 21B is an explanatory drawing showing the relationship between a first layer and a second layer in the connection structure of the variable resonator and the transmission line in the case of the five-layer structure;

FIG. 21C is an explanatory drawing showing the relationship between the second layer and a third layer in the connection structure of the variable resonator and the transmission line in the case of the five-layer structure;

FIG. 22A shows a first example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 22B shows a second example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 22C shows a third example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 22D shows a fourth example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 22E shows a fifth example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 22F shows a sixth example of the cross-sectional configuration of the connection structure shown in FIG. 21A;

FIG. 23A shows a connection structure of a variable resonator and a transmission line having a bent portion;

FIG. 23B shows a connection structure of a variable resonator and a transmission line having a bent portion;

FIG. 24 shows a connection structure of a variable resonator and a transmission line having a bent portion;

FIG. 25 shows a transmission line model for explaining electric field coupling;

FIG. 26 shows an embodiment of a variable resonator using a ring-shaped conductor line made up of conductor lines having different line widths;

FIG. 27 shows an embodiment in which a variable resonator is configured by combining two variable resonators;

FIG. 28 shows an embodiment of a variable resonator capable of switching over conductor lines of two different line lengths;

FIG. 29 shows a connection structure of a variable resonator and a transmission line when using a coplanar waveguide;

FIG. 30A is a circuit diagram for explaining a problem arises when a port impedance is different from the impedance of an input/output line;

FIG. 30B is a graph showing frequency characteristics when a switch turned on;

FIG. 30C is a graph showing frequency characteristics when a switch is turned on;

FIG. 31 shows an example of the multi-level structure of a resonator causing impedance mismatch;

FIG. 32 is a graph showing an example of the frequency characteristics of the structure shown in FIG. 31;

FIG. 33A shows circuit conditions for simulations;

FIG. 33B is a graph showing the frequency characteristics for θ=90°;

FIG. 33C is a graph showing the frequency characteristics for θ=10°;

FIG. 34A shows circuit conditions for simulations when a stub length is 0;

FIG. 34B is a graph showing frequency characteristics for different θ;

FIG. 35A is a Smith chart when θ=90° is set in the circuit of FIG. 34A;

FIG. 35B is a Smith chart for θ=10°;

FIG. 36A shows circuit conditions for simulations when a stub length is 13°;

FIG. 36B is a graph showing frequency characteristics for different θ;

FIG. 37A is a Smith chart when θ=90° is set in the circuit of FIG. 36A;

FIG. 37B is a Smith chart for θ=101° ;

FIG. 38 is a perspective view showing a variable bandwidth filter having a multi-level configuration including an open-end stub;

FIG. 39 is a graph showing frequency characteristics to indicate the effect of the open-end stub;

FIG. 40A shows an example in which a circuit adjustment element is inserted between the ground and the connecting point of an input/output line and a ring-shaped line;

FIG. 40B shows an example in which the circuit adjustment element is inserted between the input/output line and the ground, on a position away from the connecting point of the input/output line and the ring-shaped line;

FIG. 40C shows an example in which the circuit adjustment element is inserted in series with the input/output line;

FIG. 40D shows an example in which the circuit adjustment element is inserted between the ring-shaped line and the ground;

FIG. 41A shows an example in which a circuit adjustment element is provided between an input/output line and a ring-shaped line;

FIG. 41B shows an example in which the circuit adjustment element is disposed inside the ring-shaped line and connected between the ring-shaped line and the ground;

FIG. 42 shows examples of various circuit adjustment elements;

FIG. 43 shows an example an input/output line has a length of 180° instead of the provision of a circuit adjustment element;

FIG. 44A shows circuit conditions for simulations when an open-end stub is provided on an input/output line;

FIG. 44B is a graph showing the frequency characteristics for θ=10°;

FIG. 44C is a graph showing the frequency characteristics for θ=90°;

FIG. 45A shows circuit conditions for simulations when a line serving as a circuit adjustment element is inserted between an input/output line and a ring-shaped line;

FIG. 45B is a graph showing the frequency characteristics for θ=10°;

FIG. 45C is a graph showing the frequency characteristics for θ=90°;

FIG. 46A shows circuit conditions for simulations when a line having different line widths is connected as a circuit adjustment element to an input/output line;

FIG. 46B is a graph showing the frequency characteristics for θ=10°;

FIG. 46C is a graph showing the frequency characteristics for θ=90°;

FIG. 47A shows circuit conditions for simulations when an individual capacitor is inserted as a circuit adjustment element between an input/output line and the ground;

FIG. 47B is a graph showing the frequency characteristics for θ=10°;

FIG. 47C is a graph showing the frequency characteristics for θ=90°;

FIG. 48A shows an embodiment of a variable bandwidth filter in which the different positions of a ring-shaped conductor line can be connected to a transmission line via switches;

FIG. 48B shows the frequency characteristics of the variable bandwidth filter;

FIG. 49A shows a modification of the variable bandwidth filter of FIG. 48;

FIG. 49B shows the frequency characteristics of the variable bandwidth filter;

FIG. 50 shows a modification of the variable bandwidth filter shown in FIG. 49A;

FIG. 51 shows another modification of the variable bandwidth filter shown in FIG. 48A;

FIG. 52 shows still another modification of the variable bandwidth filter shown in FIG. 48A;

FIG. 53 shows still another modification of the variable bandwidth filter shown in FIG. 48A;

FIG. 54 shows an embodiment of a variable resonator in which open-end transmission lines are connected to switches connected to a ring-shaped conductor line; and

FIG. 55 shows an embodiment of a variable resonator in which short-circuited end transmission lines are connected to a ring-shaped conductor line.

BEST MODES FOR CARRYING OUT THE INVENTION

FIGS. 1A and 1B show variable resonators 20 of the present invention having ring-shaped microstrip line structures of two patterns. FIG. 1C is a cross-sectional example in which the ring of the variable resonator 20 of FIG. 1A or 1B is cut on the position of one switch 3. The variable resonators 20 of FIGS. 1A and 1B are each made up of a ring-shaped conductor line 2 (hereinafter, simply will be referred to as a ring-shaped line) and the switches 3 which are at least two circuit switches. “Ring-shaped” does not always have to be a circular shape, as will be described later, as long as the line forms a closed loop. As shown in the cross-sectional view of FIG. 1C, the ring-shaped line 2 is formed of a metal on one of the surfaces of a dielectric substrate 5. The dielectric substrate 5 has a ground conductor 4 formed of a metal on the opposite surface (will be referred to as the backside) from the surface having the ring-shaped line 2. The switch 3 has one end 31 electrically connected to the ring-shaped line 2 and the other end 32 electrically connected to the ground conductor 4 on the backside of the dielectric substrate 5 via a conductor 33 and a via hole 6. Since the shape and so on of the conductor 33 are not limited at all, the conductor 33 is not shown in FIGS. 1A and 1B. The layout of the switches 3 is not limited to equal spacings and may be freely designed to obtain a desired bandwidth. In the present specification, the switches are not limited to contact type switches and thus may be so-called switching elements using, for example, diodes, transistors, MOS devices, and so on and may have a circuit switching function with no contacts provided in a network. To be specific, switching diodes and the like are available.

The ring-shaped line 2 has a length allowing a phase change of 2π, that is, 360° at a desired resonance frequency. In other words, the ring-shaped line has a length which is a wavelength at the resonance frequency or an integral multiple of the wavelength. In the variable resonators 20 of FIGS. 1A and 1B, the ring-shaped lines are circular lines.

In this case, “length” means the circumferential length of the ring-shaped line.

“Desired resonance frequency” is a factor of performance generally required for resonators and is a given design matter. The variable resonance circuit of the present invention can be used in an alternating-current circuit and the target resonance frequency is not particularly limited. For example, the variable resonance circuit is useful when the resonance frequency is a high frequency of 100 kHz or higher.

A difference between the variable resonators 20 of FIG. 1A and FIG. 1B is whether the other end 32 of the switch 3 is disposed inside or outside the ring-shaped line 2. In the variable resonator 20 of FIG. 1A, the other end 32 of the switch 3 is disposed outside the ring-shaped line 2. In the variable resonator 20 of FIG. 1B, the other end 32 of the switch 3 is disposed inside the ring-shaped line 2.

The features of the two embodiments are applicable to, for example, the configurations of FIGS. 8, 11 and 27 (will be described later).

The characteristics of the variable resonator 20 are represented by the electromagnetic field simulations of circuits 10 shown in FIGS. 2A and 2B.

In each of the circuits 10 of FIGS. 2A and 2B, the variable resonator 20 of either FIG. 1A or 1B is connected in parallel to the input/output line 7 illustrated as a transmission line between ports P1 and P2 and the circuit 10 act as a variable bandwidth filter. In the electromagnetic field simulations, the dielectric substrate 5 had a relative dielectric constant ε_rof 9.6 and a thickness of 0.635 mm, and the ring-shaped line 2 had an outside diameter of 4 mm and an inside diameter of 3.4 mm. A conductor composing the ring-shaped line 2, a conductor forming the via hole 6, and the ground conductor 4 all had a resistance of 0. Further, the port impedance of the input/output line 7 was 50Ω. The illustration of the switches 3 is omitted for the sake of simplicity and the simulations were performed while changing the position of the via hole 6 instead.

FIGS. 3A and 3B show simulation results on the frequency characteristics of the transmission coefficient of the circuit 10.

FIG. 3A shows frequency characteristics when a position X is grounded through the via hole 6 having a diameter of 0.3 mm. The position X is one of the intersecting positions of the ring-shaped line 2 and a line passing through the center of the ring-shaped line 2 and intersecting a line L at π/2, that is, 90° as shown in FIG. 2A. The position X is set at ¾ of the length of the ring-shaped line 2 from a connecting portion C, which connects to the input/output line 7, in a counterclockwise direction (¼ in a clockwise direction) and the ring-shaped line 2 is grounded on the position X. In this case, “clockwise” and “counterclockwise” indicate circumferential directions in FIG. 2A (the same is true in the following description). A line connecting the input/output line 7 and the connecting portion C indicates that the input/output line 7 and the ring-shaped line 2 are electrically connected to each other in the circuit 10 to be simulated.

FIG. 3B shows frequency characteristics when the position of the via hole 6 is set at a position Y as shown in FIG. 2B. The position Y is set at 7/12 of the length of the ring-shaped line 2 from a connecting portion C, which connects to the input/output line 7, in a counterclockwise direction ( 5/12 in a clockwise direction) and the ring-shaped line 2 is grounded on the position Y.

As is evident from the frequency characteristics shown in FIGS. 3A and 3B, in the variable resonator 20, the position of the via hole 6 is changed, that is, the position of the switch 3 to be turned on (electrically connected) is changed, so that a frequency (a frequency having the minimum transmission coefficient) β for rejecting a signal can be largely changed without changing a frequency α allowing the passage of a signal. In other words, the bandwidth of a signal to be propagated can be largely changed according to the position of the switch 3 to be turned on. Generally, the minimum point appearing on the frequency characteristics of a transmission coefficient is called a transmission zero.

These operations will be described below in accordance with a lossless transmission line model.

FIG. 4A shows a lossless transmission line model for the resonator part of the circuit 10 shown in FIGS. 2A and 2B. The operations of the circuit 10 will be described by determining an input impedance Z_inof this model. In a resonance frequency f_r=α (FIGS. 3A and 3B), a transmission line 2₁has an electric length of π and a characteristic impedance of Z₁, a transmission line 2₂has an electric length of x (radian) and a characteristic impedance of Z₂, and a transmission line 2₃has an electric length of (π−x) and a characteristic impedance of Z₃. As is evident from this model, the sum of the electric lengths of the transmission lines 2₁, 2₂and 2₃is 2π, that is, 360°.

A path P_Amade up of the transmission lines 2₁and 2₂is a counterclockwise path from the connecting portion C to the position of the via hole 6 in FIGS. 2A and 2B, that is, to the positions represented as X and Y in FIGS. 2A and 2B. A path P_Bincluding the transmission line 2₃is a clockwise path from the connecting portion C to the position of the via hole 6 in FIGS. 2A and 2B, that is, to the positions represented as X and Y in FIGS. 2A and 2B. Reference character Z_Ldenotes an impedance to the ground on the position of the via hole 6.

In this case, an input impedance Z_inis expressed by formula (1) where j represents an imaginary unit.

$\begin{matrix} Z_{i n} = \frac{y_{22} + Y_{L}}{y_{11} (y_{22} + Y_{L}) - y_{12} y_{21}} & (1) \end{matrix}$

Where

y₁₁=−jY₂cotx+jY₃cotx
y₁₂=−jY₂cscx+jY₃cscx
y₂₁=−jY₂cscx+jY₃cscx
y₂₂=−jY₂cotx+jY₃cotx
Y₂=1/Z₂,Y₃=1/Z₃,Y_L=1/Z_L

In the case of Y₂=Y₃and in all the cases other than x=nπ (n=0, 1, 2, 3, . . . ), Z_inbecomes infinite for whatever value of Z_Land exerts the same characteristics as LC parallel resonance. Thus, in FIGS. 2A and 2B, a signal inputted from the input port is propagated to the output port. In the case of Y₂=Y₃and x=nπ, Z_in=Z_Lis obtained. Thus if Z_Lis 0, the connecting portion C between the variable resonator 20 and the input/output line 7 in FIGS. 2A and 2B is short-circuited at this frequency and the signal is not propagated.

Therefore, in the case where a variable resonator and a transmission line are connected in parallel in the configuration of a variable bandwidth filter (will be described later), when allowing the passage of a signal at a frequency whose wavelength is the conductor line length of the variable resonator, it is necessary to prevent the position of a switch to be turned on from being an integral multiple of π in terms of an electric length from the connecting portion of the transmission line and the variable resonator. Conversely, when preventing the passage of a signal at the frequency whose wavelength is the conductor line length of the variable resonator, it is sufficient to set the position of a switch to be turned on at an integral multiple of π in terms of an electric length from the connecting portion of the transmission line and the variable resonator.

In the above explanation, Y₂=Y₃was set from an analytical point of view according to formula (1). However, the effect of the present invention is not strictly obtained only by Y₂=Y₃. For example, when Y₂≠Y₃but not so different from each other, that is, in the case of Y₂≈Y₃, the resonance frequency of the variable resonator may be slightly deviated and may not be constant (in short, a desired resonance frequency cannot be kept), nevertheless, a wide bandwidth can be obtained depending on a position where the switch 3 is turned on. Thus, there would be no significant difference between a bandwidth with the desired resonance frequency and a bandwidth with a slightly deviated resonance frequency, resulting in no influence in practical use.

In other words, when a somewhat wide bandwidth is made variable, design conditions strictly requiring Y₂=Y₃are not necessary from a practical point of view. Thus when a somewhat wide bandwidth is made variable, it is not always necessary to strictly set the circumferential length of the ring-shaped line 2 one wavelength or the integral multiple of the wavelength at the resonance frequency.

Therefore, the setting of the circumferential length of the ring-shaped line 2 at a wavelength or the integral multiple of the wavelength at the resonance frequency should be understood as a technical matter including the foregoing meaning.

When the variable bandwidth filter is configured not to reject a signal but mainly to allow the passage of a signal having a desired frequency, it is not originally necessary to set the switches 3 on the positions of the integral multiples of π in terms of an electric length. Thus as shown in FIG. 4B, the switches 3 are disposed on positions other than the positions of the integral multiples of π in terms of an electric length. To be more specific, in the variable resonator of FIG. 4B, no switches are disposed on a portion indicated by the input impedance Z_inwhere connection is to be made to the transmission line, and a portion which is π away in terms of an electric length from the former portion.

Further, as is evident from the lossless transmission line model of FIG. 4A, the clockwise path and the counterclockwise path from the connecting point between the ring-shaped line 2 and the input/output line 7 to the position of the electric length π are symmetrical to each other (in the case of the ring-shaped line of FIGS. 2A and 2B, the paths are symmetrical to each other with respect to the line L), so that switch 3 may not be provided on one of the symmetric positions.

In the example of the variable resonator 20 shown in FIG. 4B, all of the switches 3 on either upper side or lower side of a line H (corresponding to the line L in FIGS. 2A and 2B) in FIG. 4B may not be provided.

The following will discuss characteristics at frequencies represented by β in FIGS. 3A and 3B. A signal does not propagate at these frequencies because the input impedance Z_inis 0 on the connecting portion between the input/output line 7 and the variable resonator 20.

In FIG. 4A, when x is π/2, that is, 90° in terms of a resonance frequency f_rof the variable resonator 20, the lossless transmission line model corresponds to the circuit of FIG. 2A and exerts characteristics shown in FIG. 3A. The electric length of the path P_Ais 3π/2, that is, 270° in terms of the resonance frequency f_r. This electric length is equivalent to π, that is, 180° at a frequency ⅔ times as high as the resonance frequency f_rand the path can be regarded as a half-wavelength stub with a short-circuited end. Thus, the input impedance Z_inon a contact between the input/output line 7 and the variable resonator 20 is 0. Further, at a frequency 4/3 times as high as (that is, twice as high as ⅔ times) the resonance frequency f_r, the path P_Acan be regarded as a one-wavelength stub with a short-circuited end and thus exerts the same characteristics. Since the other path P_Bhas an electric length of π/2, that is, 90° at the resonance frequency f_r, the path can be regarded as a half-wavelength stub with a short-circuited end at a frequency twice as high as the resonance frequency f_r. Thus, the input impedance Z_inon the contact between the input/output line 7 and the variable resonator 20 is 0. However, in this case, the frequency is out of the range of the frequency axis (horizontal axis) shown in FIG. 3A and thus is not shown in FIG. 3A.

In FIG. 4A, when x is π/6, that is, 30° at the resonance frequency f_rof the variable resonator 20, the lossless transmission line model corresponds to the circuit of FIG. 2B and exerts characteristics shown in FIG. 3B. The electric length of the path P_Ais 7π/6, that is, 210° at the resonance frequency f_r. The electric length is π, that is, 180° at a frequency 6/7 times as high as the resonance frequency f_rand the path can be regarded as a half-wavelength stub with a short-circuited end. Thus the input impedance Z_inon the contact between the input/output line 7 and the variable resonator 20 is 0. Further, regarding a frequency 12/7 times as high as (that is, twice as high as 6/7 times) the resonance frequency f_r, the path P_Acan be regarded as a one-wavelength stub with a short-circuited end and thus exerts the same characteristics. Since the other path P_Bhas an electric length of 5π/6, that is, 150° at the resonance frequency f_r, the path can be regarded as a half-wavelength stub with a short-circuited end at a frequency 6/5 times as high as the resonance frequency f_r. Thus the input impedance Z_inon the contact between the input/output line 7 and the variable resonator 20 is 0.

As described above, a signal does not propagate at frequencies represented by β in FIGS. 3A and 3B.

FIGS. 5A and 5B show a variable bandwidth filter 10 configured using the two variable resonators 20 according to the present invention. The variable bandwidth filter 10 has the two variable resonators 20 electrically connected in parallel with respect to the input/output line 7. FIGS. 6A and 6B show linear circuit simulation results on the frequency characteristics of the variable bandwidth filter 10. The illustration of the switches 3 is omitted for the sake of simplicity and the position of the via hole 6 is changed for the simulations. Further, the resonance frequency of the variable resonator 20 is set at 5 GHz in the linear circuit simulations.

Moreover, in the linear circuit simulations, the variable bandwidth filters 10 shown in FIGS. 5A and 5B each have the two variable resonators 20 connected to each other via a line having a quarter wavelength (corresponding to a phase change of 90°) at 5 GHz which is the resonance frequency of the variable resonator.

In the linear circuit simulations, the variable bandwidth filters 10 were simulated as to the positioning of the via holes of the two cases shown in FIGS. 5A and 5B.

In the variable bandwidth filter 10 of FIG. 5A, the positions of the via holes 6 of the two variable resonators 20 are different from each other. To be specific, the via hole 6 of the variable resonator 20 on the left of FIG. 5A is placed at 5/12 of the length of the ring-shaped line 2 from a connecting portion D in a counterclockwise direction, and the via hole 6 of the variable resonator 20 on the right of FIG. 5A is placed at 4/9 of the length of the ring-shaped line 2 from a connecting portion E in a counterclockwise direction.

In the variable bandwidth filter 10 of FIG. 5B, the positions of the via holes 6 of the two variable resonators 20 are different from those of FIG. 5A. To be specific, the via hole 6 of the variable resonator 20 on the left of FIG. 5B is placed at 4/9 of the length of the ring-shaped line 2 from a connecting portion D in a counterclockwise direction, and the via hole 6 of the variable resonator 20 on the right of FIG. 5B is placed at 17/36 of the length of the ring-shaped line 2 from a connecting portion E in a counterclockwise direction.

As shown in FIGS. 6A and 6B, the bandwidth (in this case, a bandwidth of −3 dB around 5 GHz) of the variable bandwidth filter 10 shown in FIG. 5A is about 320 MHz and the bandwidth of the variable bandwidth filter 10 shown in FIG. 5B is about 100 MHz.

As is evident from the above description, the variable bandwidth filter 10 of the present invention makes it possible to greatly change the bandwidth while keeping the center frequency (in this case, 5 GHz) constant, by changing the position of the via hole 6, that is, the position of the switch 3.

Although the two variable resonators 20 are used in the variable bandwidth filters 10 of FIGS. 5A and 5B, the number of the variable resonators 20 is not particularly limited to two. The variable bandwidth filter 10 can be configured using at least one variable resonator 20. The variable bandwidth filter 10 using one variable resonator 20 is configured as shown in FIG. 2.

Although it is desirable to connect the variable resonators 20 by the line having a quarter wavelength at the resonance frequency of the variable resonator 20, the configuration is not particularly limited.

The variable bandwidth filter 10 of the present invention is also characterized by a small insertion loss in a passband having the center at the resonance frequency of the variable resonator 20. The influence of the switches which increase an insertion loss and are used in the variable resonator is examined in the following description.

The frequency characteristics of the variable bandwidth filter 10 were simulated in the cases where the switch 3 of the variable bandwidth filter 10 in FIG. 5A has a resistance of 0Ω and a resistance of 2Ω. FIGS. 7A and 7C show the simulation results. FIG. 7A shows the case where the switch 3 has a resistance of 0Ω as shown in FIG. 5A. FIG. 7C shows the case where the switch 3 has a resistance of 2Ω as shown in FIG. 7B. As is evident from comparisons between FIGS. 7A and 7C, even when the resistance of the switch 3 is increased, the insertion loss in a passband around the center frequency (in this case, 5 GHz) hardly changes. This finding is based on the fact that the operation of the variable resonator 20 described with FIG. 4A makes the input impedance Z_ininfinite at the resonance frequency f_rregardless of the impedance Z_L. Thus, it is understood that in the variable bandwidth filter 10 of the present invention, characteristics with a low insertion loss can be obtained even using a switch having a somewhat high resistance.

Conversely, the configuration taking the advantage of a resistance can also be used. For example, as shown in FIG. 7D, it is possible to actively use a resistance by switching the case where the ring-shaped line 2 is directly connected to the ground conductor 4 by using a switch 35 acting as a low-resistance switch and the case where the ring-shaped line 2 is connected to the ground conductor 4 via a resistor 9 having a resistance of several ohms to several tens ohms which is higher than the resistance of the switch 35. In this case, it is possible to select the case where the propagation of a signal is suppressed in a band affected by the resistor 9 having a resistance of several ohms to several tens ohms and the case where even a signal around the band which would be affected by the resistance can also be propagated by minimizing the resistance.

Although the foregoing examples show the use of a resistor, the use of an element is not limited to a resistor. It is possible to use such a passive element as variable resistor, inductor, variable inductor, capacitor, variable capacitor, or piezoelectric element. Of course, in FIGS. 1A and 1B and other embodiments, too, the switches 3 of the ring-shaped line 2 may be grounded through such a passive element, or may be made selectable by a switch 35 to ground either via such passive element or directly.

In addition to the variable bandwidth filter 10 configured by connecting the variable resonators 20 to the transmission line as shown in FIGS. 5A and 5B, the variable bandwidth filter 10 may be configured by connecting the input/output lines 7, which are electrically connected to the variable resonators 20, with each other via a variable capacitor 11 as shown in FIG. 8. A circuit element is not limited to a variable capacitor. For example, a circuit element such as a capacitor, an inductor, a variable inductor, and a transistor may be used.

Further, the variable bandwidth filter can be configured by connecting the input/output lines 7 with each other through electric field coupling or magnetic field coupling. FIG. 9 shows the variable bandwidth filter 10 configured by electric field coupling and FIG. 10 shows the variable bandwidth filter 10 configured by magnetic field coupling. In the electric field coupling of FIG. 9, two variable resonators 20 are spaced between two input/output lines 7a and 7b extended on the same straight line. In the magnetic field coupling of FIG. 10, lines 7c and 7d extended at right angles on the same side from the opposed ends of the input/output lines 7a and 7b on the same straight line of FIG. 9 are formed in parallel with each other, and the two variable resonators 20 are spaced between the parallel lines 7a and 7b.

FIGS. 11A, 11B and 11C show embodiments of the variable bandwidth filter according to the present invention. The variable bandwidth filter 10 of FIG. 11A is made up of two variable resonators 20a and 20b having different sizes and switches 3a and 3b serving as circuit switches provided between the variable resonators and an input/output line 7 acting as a transmission line. The center frequency of the variable bandwidth filter 10 can also be made variable using the two variable resonators 20a and 20b having resonance frequencies varied with different circumferential lengths of the ring-shaped lines.

As to the resonance frequencies of the variable resonators 20a and 20b, the connecting portions between the variable resonators 20a and 20b and the switches 3a and 3b have high impedances. Thus the resistances of the switches 3a and 3b between the variable resonators 20a and 20b and the input/output line 7 hardly affect the insertion loss of a passband. Thus in addition to the characteristic of the variable resonator of the present invention in which the switches between the variable resonators and the ground conductor hardly affect an insertion loss at the resonance frequency, the variable bandwidth filter of FIG. 11A is characterized in that the center frequency and the bandwidth can be changed and a passband characteristic can be obtained with a low loss regardless of the resistances of the used switches 3a and 3b.

The variable bandwidth filter 10 of FIG. 11B is made up of two variable resonators 20a and 20b having the same resonance frequency and switches 3a and 3b which are circuit switches provided between the variable resonators and an input/output line 7 acting as a transmission line. The variable bandwidth filter 10 of FIG. 11C has a configuration similar to that of the variable bandwidth filter 10 of FIG. 1B. However, the variable bandwidth filter 10 of FIG. 11C is different from that of FIG. 11B in that the variable bandwidth filter 10 of FIG. 11B uses the two variable resonators 20a and 20b having the same characteristic impedance and the variable bandwidth filter 10 of FIG. 11C uses the two variable resonators 20a and 20b having different characteristic impedances.

In the case of the variable bandwidth filter 10 of FIG. 11B, two states are selectable, that is, a state where only one of the variable resonators is connected via the switches 3a and 3b and a state where the variable resonators 20a and 20b are both connected via the switches 3a and 3b. In these states, the resonance frequency is the same but the frequency characteristics are different. When both of the variable resonators are connected, the attenuation of a signal becomes large at a frequency away from the resonance frequency as compared with the case where only one of the variable resonators is connected. This is because the two parallel-connected variable resonators equivalently have a half characteristic impedance of a single variable resonator.

FIGS. 12A, 12B and 12C show the frequency characteristics of the variable bandwidth filter for each relationship between the characteristic impedances of the variable resonator and the input/output line 7. FIG. 12A shows the frequency characteristics of the variable bandwidth filter when the characteristic impedance of the variable resonator is twice that of the input/output line 7. FIG. 12B shows the frequency characteristics of the variable bandwidth filter when the characteristic impedance of the variable resonator is the same as that of the input/output line 7. FIG. 12C shows the frequency characteristics of the variable bandwidth filter when the characteristic impedance of the variable resonator is half that of the input/output line 7.

As is evident from the frequency characteristics of FIGS. 12A to 12C, when the variable resonator is lower in characteristic impedance than the input/output line 7, the amount of attenuation of a signal increases as the frequency moves away from the resonance frequency, that is, the bandwidth decreases.

This finding will be described below with reference to the variable bandwidth filter 10 of FIG. 11B. For example, when the characteristic impedances of the variable resonators 20a and 20b are set twice as high as that of the input/output line 7, the frequency characteristics of FIG. 12A correspond to the frequency characteristics of the variable bandwidth filter 10 when one of the switches 3a and 3b of FIG. 11B is turned on, and the frequency characteristics of FIG. 12B correspond to the frequency characteristics of a variable bandwidth filter (55) when both of the switches 3a and 3b are turned on.

Further, this finding will be described below with reference to the variable bandwidth filter 10 of FIG. 11C. For example, when the characteristic impedance of the variable resonator 20a is set twice as high as that of the input/output line 7 and the characteristic impedance of the variable resonator 20b is set at half that of the input/output line 7, the frequency characteristics of FIG. 12A correspond to the frequency characteristics of the variable bandwidth filter 10 in which the switch 3a is turned on and the switch 3b is turned off. The frequency characteristics of FIG. 12C correspond to the frequency characteristics of the variable bandwidth filter 10 in which the switch 3a is turned off and the switch 3b is turned on.

Thus in the variable bandwidth filter 10 of FIG. 11B, the characteristic impedances of the variable resonators can be switched relative to the input/output line 7 by changing the on/off states of the switches 3a and 3b, and the frequency characteristics of the variable bandwidth filter 10 can be changed in response to the two states.

In the variable bandwidth filter 10 of FIG. 11C, three states are selectable, that is, a state where either one of the variable resonators is connected via the switches 3a and 3b and a state where the variable resonators are both connected via the switches 3a and 3b. In these states, the resonance frequency is the same but the frequency characteristics are different.

As in the variable bandwidth filter 10 of FIG. 11B, in the variable bandwidth filter 10 of FIG. 11C, the characteristic impedances of the variable resonators are switched by changing the on/off states of the switches 3a and 3b, and the frequency characteristics of the variable bandwidth filter 10 can be changed in response to the three states.

FIG. 13 shows another embodiment of the variable bandwidth filter according to the present invention.

Unlike the variable bandwidth filters 10 of FIGS. 5A and 5B, a variable resonator 20 is electrically connected in series to an input/output line 7. The input/output line 7 is connected to the variable resonator 20 on two portions separated from each other by a half wavelength at the resonance frequency of the variable resonator 20, that is, on portions separated by π in terms of an electric length on the variable resonator 20.

The operation of the variable resonator 20 of the present invention was explained in accordance with FIG. 4A. In the explanation, x=0 is set and the part having the impedance Z_Lis regarded as the input/output line 7. This case corresponds to the variable bandwidth filter 10 of FIG. 13. In this explanation, when x=0 is set in FIG. 4A, the impedance Z_Lis equal to the input impedance Z_inat the resonance frequency of the variable resonator 20, which means that if the impedance Z_Lis not a short circuit but the input/output line 7, a signal propagates at the resonance frequency. Thus this configuration operates as a variable bandwidth filter.

FIG. 14 shows the frequency characteristics of a variable bandwidth filter 10 of FIG. 13 as circuit simulation results. In this example, a switch 3 of θ=30 is turned on. As compared with the variable bandwidth filters 10 of FIGS. 5A and 5B having the variable resonators connected in parallel, a signal extremely attenuates at only one frequency and the number of such transmission zeros is half or less. This is because in the configuration of the variable bandwidth filter 10 of FIG. 13, a signal extremely attenuates only at a frequency set by the path P_Bof the lossless transmission line model of FIG. 4A. Although the single variable resonator is used in the variable bandwidth filter 10 of FIG. 13, a plurality of variable resonators 20 may be connected in series as shown in FIG. 15 or as shown in FIG. 16, some of the variable resonators 20 may be connected in parallel to the input/output line 7 while the other variable resonators are connected in series to the input/output line 7. In FIGS. 15 and 16, the two variable resonators are illustrated.

As usage patterns of the variable resonator of the present invention, variable bandwidth filters have been mainly described in the foregoing. Referring to FIG. 17, an example of a bias circuit will be discussed as another usage pattern. In an illustrated bias circuit 40, a bias voltage is supplied to a field-effect transistor 43. In the bias circuit 40, by taking the advantage of the input impedance on the connecting portion between the input/output line 7 and the variable resonator 20 being infinite in the variable resonator 20 as long as a switch having been turned on is disposed on a position other than positions separated by nπ from the connecting portion between the input/output line 7 and the variable resonator 20, a bias supply point B can be disposed in a wide region on the variable resonator other than positions separated by nπ from the connecting portion. On the bias supply point B, a capacitor 41 plays the same role as the switch having been turned on (not shown). Thus by using the variable resonator of the present invention, it is possible to suppress the influence of the bias circuit on high-frequency characteristics without the need for high working accuracy for the bias circuit.

The bias circuit requires a mere resonator and not necessarily requires a variable resonator. However, the above example was described as an exemplary usage pattern of the variable resonator.

As is evident from this example, it should be noted that the variable resonator of the present invention is equivalent to a mere resonator in some usage patterns. In other words, when only one specific switch 3 is used, the variable resonator of the present invention simply acts as a fixed resonator. Furthermore, instead of switching electrical connection/disconnection by the switch 3, the capacitor 41 may be provided on, for example, a point on the ring-shaped line 2 to keep only the on state. In this case, the on state is kept not only by the capacitor 41 but also by an appropriate circuit element.

From this point of view, the variable bandwidth filter can be similarly configured as a fixed filter. To put it simply, for example, in FIG. 5A, the switch 3 is provided only on a predetermined position (at 30° in FIG. 5A) on the ring-shaped line 2 of the left resonator or the capacitor 41 is provided on the position to keep only an on state, and similarly the switch 3 is provided only on a predetermined position (at 20° in FIG. 5A) on the ring-shaped line 2 of the right resonator or the capacitor 41 is provided on the position to keep only the on state, so that a fixed filter operating in a predetermined bandwidth can be configured.

Although the above variable resonators and the variable resonators used in the variable bandwidth filter are all circular, the shape of the variable resonator is not particularly limited to a circle. In FIG. 4A, when a characteristic impedance Z₂and a characteristic impedance Z₃satisfy the condition of Z₂=Z₃in the lossless transmission line model, the variable resonator may be oval as shown in FIG. 18 or may be arched as shown in FIG. 19.

FIGS. 20A and 20B show modifications of the variable resonator and the connection of the variable resonator and the transmission line, from a viewpoint of an insertion loss which occurs on the transmission line due to the connection of the variable resonator.

FIG. 20A shows that a variable resonator having a circular ring-shaped line 2 is connected to an input/output line 7. The illustration of the switches 3 is omitted for the sake of simplicity and, instead, the grounding position is shown as a position of a via hole. As a result of electromagnetic field simulations, an insertion loss of 2.92 dB was obtained. The insertion loss occurs due to reflection on a connecting portion. The occurrence of the insertion loss will be described with reference to the transmission line model of FIG. 25. An impedance on a connecting portion decreases due to magnetic field coupling (represented as reference character M) between a transmission line and a ring-shaped line and an input signal is reflected on the connecting portion, so that the loss occurs.

Thus, it is estimated that by lowering such magnetic field coupling, the insertion loss can be reduced.

As shown in FIG. 20B, when the variable resonator having an oval ring-shaped line 2 is connected to the input/output line 7, the insertion loss decreases to 0.81 dB. In other words, the insertion loss is reduced only by changing the shape of the ring-shaped line. This is because magnetic field coupling between the input/output line 7 and the ring-shaped line 2 is reduced by connecting the variable resonator to the input/output line such that the major axis of the ellipse, which is the shape of the ring-shaped line, intersects the input/output line 7.

In order to compare insertion losses under the same conditions, the same grounding portions are illustrated and the other conditions are the same (the same is true in the following description).

When a multilayer structure is acceptable as a design for a variable resonator, for example, the configuration of FIG. 21A may be used. When it is assumed that the closest layer in FIG. 21A is an upper layer and layers behind the upper layer are lower layers, an L-like input/output line 7a is disposed atop, a variable resonator is disposed under the input/output line 7a, and the end of a right-angled extended portion 7c of the input/output line 7a and the ring-shaped line 2 of the variable resonator overlap each other in an area S as shown in FIG. 21B. Further, as shown in FIG. 21C, an L-like input/output line 7b is disposed under the variable resonator and a right-angled extended portion 7d of the input/output line 7b and the ring-shaped line 2 of the variable resonator overlap each other in the area S. A via hole 66 is provided in the area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

Some modes of this multilayer structure will be further described with reference to sectional views taken along the line of sight of FIG. 21C. FIG. 21C is a plan view showing the multilayer structure. In the sectional views, an upper layer is disposed atop and lower layers are disposed under the top layer. The illustration of the switches 3 and so on is omitted to simplify the cross-sectional configurations.

In a first example of the multilayer structure, as shown in FIG. 22A, a ground conductor 4 serving as the bottom layer is formed under a laminated dielectric substrate 5 and an input/output line 7a is formed on the dielectric substrate 5. A ring-shaped line 2 and an input/output line 7b of the variable resonator are embedded and fixed in the dielectric substrate 5. The ring-shaped line 2 is disposed above the input/output line 7b. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b. For example, in order to activate the switches 3 (not shown) from the outside, via holes 67 are used to electrically connect the outside of the dielectric substrate and the switches 3 (not shown) on the ring-shaped line 2 which is embedded and fixed in the dielectric substrate 5, and the via holes 67 are electrically connected to uppermost conductors 330 formed on the top surface of the dielectric substrate 5. Such a multilayer structure can be obtained by forming the dielectric substrate 5 as a laminate structure. In FIG. 22A, it should be noted that the via hole 6, the conductor 33, and so on of FIG. 1C are not illustrated and the via hole 67 does not have the same function with the same object as the via hole 6.

In a second example, as shown in FIG. 22B, a ground conductor 4 serving as the bottom layer is formed under a dielectric substrate 5 and a ring-shaped line 2 is formed on the top surface of the dielectric substrate 5. An input/output line 7b is embedded and fixed in the dielectric substrate 5. An input/output line 7a is disposed above the ring-shaped line 2 and is supported by a support 200. In FIG. 22B, the support 200 is disposed between the input/output line 7a and the dielectric substrate 5 but the present invention is not limited to this configuration. Other configurations may be used as long as the input/output line 7a can be supported. The material of the support 200 can be freely selected according to the arrangement of the support 200. In the example of FIG. 22B, the support 200 may be made of either a metal or a dielectric. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

In a third example, as shown in FIG. 22C, a ground conductor 4 serving as a bottom layer and a dielectric substrate 5 formed thereon are in contact with each other, and the dielectric substrate 5 is in contact with an input/output line 7b and conductors 331 which are formed thereon. A ring-shaped line 2 is supported above the input/output line 7b and the conductors 331 by supports 200. An input/output line 7a is supported above the ring-shaped line 2 by a support 201 disposed between the input/output line 7a and the input/output line 7b. In the configuration of FIG. 22C, the support 201 is made of a dielectric to prevent electrical connection between the input/output lines 7a and 7b. The conductors 331 and conductor columns 67 are disposed between the ring-shaped line 2 and the dielectric substrate 5 at positions corresponding to the switches 3. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

In a fourth example, as shown in FIG. 22D, a ground conductor 4 serving as a bottom layer and a dielectric substrate 5 formed thereon are in contact with each other, and the dielectric substrate 5 is in contact with an input/output line 7b formed thereon. A ring-shaped line 2 formed on the dielectric substrate 5 is in contact with the dielectric substrate 5. As shown in FIG. 22D, since the dielectric substrate 5 has a stepped structure, the ring-shaped line 2 is disposed above the input/output line 7b while the input/output line 7b and the ring-shaped line 2 are both in contact with the dielectric substrate 5. An input/output line 7a is supported above the ring-shaped line 2 by the support 201 disposed between the input/output line 7a and the input/output line 7b. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

In a fifth example, as shown in FIG. 22E, a ground conductor 4 serving as the bottom layer and a dielectric substrate 5 formed thereon are in contact with each other, and the dielectric substrate 5 is in contact with an input/output line 7a and a ring-shaped line 2 which are formed thereon. An input/output line 7b is embedded and fixed in the dielectric substrate 5. The input/output line 7a and the ring-shaped line 2 may be integrally formed as in, for example, the configurations of FIGS. 20A and 20B, or may be formed as separate members and electrically connected to each other. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

In a sixth example, as shown in FIG. 22F, a ground conductor 4 serving as the bottom layer and a dielectric substrate 5 formed thereon are in contact with each other, and the dielectric substrate 5 is in contact with an input/output line 7b and a ring-shaped line 2 which are formed thereon. The input/output line 7b and the ring-shaped line 2 may be integrally formed as described above, or may be formed as separate members and electrically connected to each other. An input/output line 7a is supported above the ring-shaped line 2 and the input/output line 7b by the support 201 disposed between the input/output line 7a and the input/output line 7b. Further, a via hole 66 is provided in an area S to electrically connect the input/output line 7a, the ring-shaped line 2, and the input/output line 7b.

In the configuration of FIG. 21A, the insertion loss decreased to 0.12 dB according to the result of the electromagnetic field simulations.

Moreover, as shown in FIG. 23A, a V-shaped bent portion T may be provided on a part of the input/output line 7 and the bent portion T and the ring-shaped line 2 of the variable resonator may be connected to each other. In this way, the insertion loss can be reduced by increasing a distance between the input/output line 7 and the ring-shaped line 2. In this case, the insertion loss decreased to 0.53 dB according to the result of the electromagnetic field simulations.

For the convenience of a circuit configuration having a plurality of variable resonators, a variable resonator and an input/output line entirely shaped like V can be connected to each other as shown in FIG. 23B. In this case, the insertion loss decreased to 0.5 dB according to the result of the electromagnetic field simulations.

In FIGS. 23A and 23B, the ring-shaped line 2 and the input/output line 7 are electrically connected to each other in the same layer while being integrally formed or formed as separate members. However, the ring-shaped line 2 and the input/output line 7 can be configured as a multilayer structure as shown in FIG. 21A.

Further, as a modification of the connecting configuration of FIG. 23A, as shown in FIG. 24, a ring-shaped line 2 is formed to extend in tangential directions from both ends of a circular portion indicated by a broken line, and the ring-shaped line 2 is combined with the top of the V-shaped bent portion of an input/output line 7 so as to form “X”. The ring-shaped line 2 is deformed into a teardrop shape. With this configuration, the bent portion T of the input/output line 7 may be connected to a bent portion U of the ring-shaped line 2 which is shaped like a teardrop in the variable resonator.

In the configuration of FIG. 24, the insertion loss decreased to 0.04 dB according to the result of the electromagnetic field simulations.

As compared with the connecting configuration of FIG. 23A, the insertion loss is considerably reduced in the connecting configuration of FIG. 24. This is because the input/output line 7 and the line 2 of the variable resonator are further separated from each other, and in the connecting configuration of FIG. 24, the ring-shaped line 2 hardly has a portion parallel to the input/output line 7 near the connecting portion of the input/output line 7 and the ring-shaped line 2 in contrast to the connecting configuration of FIG. 23A in which the ring-shaped line 2 has a line portion parallel to the input/output line 7, so that magnetic field coupling is more unlikely to occur. According to this examination, the shape of the ring-shaped line 2 is not limited to the teardrop shape of FIG. 24 and any shape can be used as long as the connecting configuration of the input/output line 7 and the ring-shaped line 2 causes less magnetic field coupling.

Further, as shown in FIG. 26, two input/output lines 2a and 2b having different line widths Wa and Wb may be connected like a loop to form a ring-shaped line 2 of the variable resonator. Although FIG. 26 shows two line widths, the number of line widths is not limited to two and thus lines having three or more different line widths can be similarly connected like a loop to form a ring-shaped line 2 of the variable resonator. Also in this case, the characteristic impedance Z₂and the characteristic impedance Z₃satisfy the condition of Z₂=Z₃on paths relative to the electric length 7 in the lossless transmission line model of FIG. 4A. In these drawings, illustration of the switches 3 is not shown.

In a variable resonator 20 of FIG. 27, a variable resonator 20b having a different line width is provided inside a variable resonator 20a, and the variable resonators 20a and 20b are electrically connected to each other via switches 3a and 3b which are two circuit switches. The switch 3b is connected to the position of a half wavelength or integral multiple thereof at the resonance frequency of the variable resonator 20a from the position of the connected switch 3a on the ring-shaped line 2a of the variable resonator 20a and, at the same time, connected to the position of a half wavelength or integral multiple thereof at the resonance frequency of the variable resonator 20b from the position of the connected switch 3a on the ring-shaped line 2b of the variable resonator 20b. The variable resonator 20 is a modification of the variable bandwidth filter of FIG. 11C in which the two variable resonators having different characteristic impedances are used. This configuration makes it possible to reduce an area required for a circuit configuration. In this modification, the resonators having different line widths are combined. Resonators having the same line width may be combined instead.

In a variable resonator of FIG. 28, branching switches 39 acting as two circuit switches for selecting two lines having different lengths are provided on the ring-shaped line of a variable resonator 20. The synchronized switching of the branching switches 39 makes it possible to select one of line portions 2c and 2d having different lengths, achieving two kinds of variable resonators having different circumferential lengths. One of the variable resonators has a ring-shaped line closed by a common line portion 2e and the line portion 2c and the other variable resonator has a ring-shaped line closed by the common line portion 2e and the line portion 2d. The ring-shaped lines are selected thus by the branching switches 39, so that the line length of the variable resonator can be changed and the resonance frequency can be variable. Although the variable resonator of FIG. 28 has the same function as the variable resonator of FIG. 11A, the area required for the variable resonator of FIG. 28 can be smaller.

The ring-shaped line closed by the common line portion 2e and the line portion 2c and the ring-shaped line closed by the common line portion 2e and the line portion 2d have different lengths which are a wavelength at the resonance frequency or the integral multiple of the wavelength.

In this configuration, the two lines 2c and 2d are illustrated as an example. Three or more lines having different circumferential lengths can be similarly configured.

Regarding the two embodiments of the variable resonator 20 shown in FIGS. 1A and 1B, a supplementary explanation will be described below. In the variable resonator 20 of FIG. 1A, the other end 32 of each switch 3 is disposed outside the ring-shaped line 2. Thus, the provision of the switches 3 near the connecting portion between the variable resonator 20 and the input/output line 7 is limited in order to prevent contact with the input/output line 7. Meanwhile, in the variable resonator 20 of FIG. 1B, the other end 32 of each switch 3 is disposed inside the ring-shaped line 2 and thus such a limitation is not imposed. However, in the variable resonator 20 of FIG. 1B, for example, when a wire for operating each switch 3 is connected from the outside of the variable resonator 20, the wire may have to be extended to the inside of the variable resonator 20 over the ring-shaped line 2. Thus, it is difficult to realize the variable resonator 20 on a single-layer substrate. This difficulty can be easily overcome by forming a double-layer substrate in which, for example, the variable resonator 20 is disposed as a lower layer and the wires for operating the switches 3 are disposed as an upper layer. The variable resonator 20 of FIG. 1A does not cause this difficulty.

In the foregoing embodiments, microstrip line structures are used. The present invention is not limited to such a line structure, and thus line structures such as a coplanar waveguide may be used.

FIG. 29 shows the case of a coplanar waveguide. Ground conductors 4a and 4b are disposed on the same surface of a dielectric substrate, and an input/output line 7 connected to a variable resonator 20 is disposed in a gap between the ground conductors 4a and 4b. Further, a ground conductor 4c is disposed inside the ring-shaped line 2 of the variable resonator 20 without making contact with the ring-shaped line 2. The ground conductors 4b and 4c are electrically connected to each other via air bridges 95 to have an equal potential. The air bridges 95 are not necessary constituent elements when a coplanar waveguide is used. For example, the following configuration may be used: A rear ground conductor (not shown) is disposed on one of the surfaces of the substrate, the surface being opposite from the surface having the ground conductors 4a, 4b and 4c and the input/output line 7, the ground conductor 4c and the rear ground conductor are electrically connected to each other via a via hole, and the ground conductor 4b and the rear ground conductor are electrically connected to each other via a via hole, so that the ground conductors 4b and 4c have an equal potential.

In the foregoing embodiments, the impedances of the ports P1 and P2 are equal to that of the input/output line 7. In actual designs, these impedances may not be equal to each other. In this case, the resonance frequency may be deviated by changing the position of the switch to be turned on.

FIG. 30A shows a specific example in which one of the variable resonators 20 of the present invention is connected to the input/output line 7. In the variable resonator 20, a ring-shaped line (the length is a wavelength of 5 GHz) 2 having a characteristic impedance of 50Ω is formed and the ends at a plurality of switches (two switches 3₁and 3₂in FIG. 30A) are connected to a ring-shaped line 2. The other ends of the switches are connected to the ground conductors. In FIG. 30A, the switches 3₁and 3₂are provided on the angular positions of 10° and 90° from the position of 180° in terms of an electric length from the connecting portion between the ring-shaped line 2 and the input/output line 7. The impedance Z₀of the input/output ports P1 and P2 is 50Ω. The following will describe the case where the impedance Z₁of the input/output line 7 is different from the impedance Z₀of the input/output ports P1 and P2. In this example, the characteristic impedance Z₁is 70Ω.

The present invention is characterized in that by selecting one to be turned on of the switches 3₁and 3₂connected to the ring-shaped line 2, the bandwidth can be changed while keeping the resonance frequency. However, as shown in FIG. 30A, when the variable resonator 20 is connected to the input/output line 7 having the characteristic impedance Z₁different from the port impedance Z₀, the resonance frequency is changed by the switch to be turned on, as shown in FIGS. 30B and 30C indicating the frequency characteristics of a transmission coefficient (solid line) and a reflection coefficient (broken line) between the input/output ports when the switch 3₁is turned on and when the switch 3₂is turned on, respectively.

This problem arises also in the circuit of FIG. 31. FIG. 31 shows an example of a multi-layer structure including lines 7a and 7b for inputting and outputting signals to and from the variable resonator 20. FIG. 32 shows the frequency characteristics of the reflection coefficient of FIG. 31. The angular position of the switch having been turned on corresponds to an angular position θ of FIG. 31. FIG. 32 shows that the switch to be turned on is selected by changing the value of θ to 0°, 20°, 40°, 60° and 80°. However, in this example, the resonance frequency of the variable resonator is about 10 GHz. As is evident from FIG. 32, the resonance frequency changes around 10 GHz according to the value of the angular position θ. The resonance frequency is changed by a mismatch between the characteristic impedance Z₁and the port impedance Z₀. The mismatch is caused by electromagnetic field coupling occurring on a portion where the input/output lines 7a and 7b are vertically opposed to each other and on the via hole 66 connecting the upper and lower input/output lines 7a and 7b in FIG. 31. This phenomenon is similar to that of FIG. 30A. Even when the width of the input/output line 7 changes, the characteristic impedance Z₁changes.

FIG. 33A is a circuit for simulating the influence of change in the impedance of an input/output line 7 on a characteristic between input/output ports P1 and P2 caused by electromagnetic field coupling near portions connected to a variable resonator 20. For simulations, line portions near the connecting portion of this variable resonator are represented as lines 7c connecting input/output lines 7a and the variable resonator 20. Lines connecting the two lines 7c while intersecting each other represent electromagnetic field coupling on the input/output ends of the lines 7c.

FIGS. 33B and 33C show frequency characteristics of a transmission coefficient (solid line) and a reflection coefficient (broken line) between ports P1 and P2 when the even mode impedance and the odd mode impedance of the input/output line 7 are 66Ω and 26Ω with the lines 7c brought close to each other. FIG. 33B shows characteristics when θ is 90° and FIG. 33C shows characteristics when θ is 10°. In this case, as in FIGS. 31 and 32, the resonance frequency for θ=90° is 4.88 GHz and the resonance frequency for θ=10° is 5 GHz. The resonance frequency changes in response to the switch to be turned on.

In order to solve this problem, in the following embodiment, a circuit adjustment element is added to a line and/or a resonator. FIGS. 34A and 36A are circuit diagrams for explaining the function of an added circuit adjustment element 8. In the following explanation, a stub having an open end is used as an example of the circuit adjustment element 8. Input/output lines 7 connected to a variable resonator 20 have a characteristic impedance of 70Ω and ports P1 and P2 have an impedance of 50Ω. The path length of the variable resonator 20 is equal to a wavelength at 5 GHz. On a position where the variable resonator 20 is connected to the input/output lines 7, an open-end stub 8 is connected.

First, when the stub 8 is not added, the electric length of the stub is represented as 0° in FIG. 34A and the frequency characteristics of S21 and S11 are obtained as shown in FIG. 34B. FIG. 34B shows four curves. A solid line indicates S21 (transmission coefficient), a broken line indicates S11 (reflection coefficient), a thick line indicates characteristics when a switch at 90° is turned on, and a thin line indicates characteristics when a switch at 10° is turned on. Resonance occurs at 5 GHz when the switch at 10° is turned on and at 5.1 GHz when the switch at 90° is turned on. The resonance frequency changes as in the above description.

FIG. 35A is a Smith chart showing the reflection coefficient S11 of the port P1. A thick line indicates the overall characteristics of the circuit of FIG. 34A and a thin line indicates the characteristics of only the input/output line 7 in the circuit of FIG. 34A, except for the variable resonator 20. Since the variable resonator 20 of FIG. 34A resonates at 5 GHz, the variable resonator 20 has an infinite impedance on the connecting point of the input/output line 7 and the variable resonator 20. Therefore, the impedance is equivalent to the absence of the variable resonator 20 at 5 GHz, which agrees with the characteristics of only the input/output line 7. S11 is minimized at a point S on the thick line. The point S is the closest to a point (point O in FIG. 34A) having a port impedance of 50Ω. The point S has a resonance frequency of 5.18 GHz which is different from 5 GHz, the resonance frequency of the variable resonator 20.

At θ=10°, as shown in FIG. 35B, the reactance component of the impedance of the variable resonator 20 rapidly changes relative to a frequency as compared with the case of θ=90°. Thus the point S has a frequency of 5.006 GHz which is not largely deviated from 5 GHz. The resonance frequency of the overall circuit (the minimum frequency of S11) changes thus according to the angular position θ of the switch having been turned on. As shown in FIG. 34A, even when connecting the input/output line 7 having an impedance different from that of the port to the variable resonator 20, the resonance frequency of the ring-shaped variable resonator 20 is constant regardless of the position θ of the switch having been turned on. Thus the impedance at 5 GHz is not deviated even when the position θ of the switch having been turned on is changed. If the characteristic impedance Z₁of the input/output line 7 is 50Ω which is equal to the port impedance Z₀, the thin line has the point O and such a change does not occur.

The following will describe the case where the stub 8 is added. FIG. 36A shows that the open-end stub 8 having a characteristic impedance of 50Ω and an electric length of 13° is connected in parallel to a variable resonator 20. FIG. 36B shows characteristics corresponding to FIG. 34B. As is evident from FIG. 36B, the provision of stub 8 keeps the resonance frequency of the overall circuit constant at 5 GHz regardless of the angular position of the switch having been turned on. This finding will be further described with reference to FIGS. 37A and 37B. Also in FIGS. 37A and 37B, broken lines indicate the characteristics of only the input/output lines 7 in FIG. 34A, except for the variable resonator 20. In FIG. 37A where a switch on the angular position of 90° is turned on, a point P represents the reflection coefficient of 5 GHz in FIG. 35A. The point P is moved to a point S by the stub 8. Thus S11 at 5 GHz is minimized. As described above, the variable resonator 20 has an open impedance at 5 GHz and the impedance is constant regardless of the position of the switch having been turned on. Thus also in FIG. 37B where the switch at 10° is turned on, the reflection coefficient at 5 GHz does not move from a point S. Therefore, it is understood that the resonant frequency of the overall circuit can be made invariable by properly providing the stub 8 regardless of the position of the switch having been turned on.

FIG. 38 shows a model for confirming the effect of the stub through electromagnetic field simulations. The stub 8 is added to the model of FIG. 31. FIG. 39 shows the frequency characteristics of the reflection coefficient. It is found that a frequency where S11 is minimized converges as compared with the characteristics of FIG. 32, so that the effect of the stub can be confirmed. In this case, the open-end stub is used as the circuit adjustment element 8. Any element can be used as long as the element can adjust a reactance. Moreover, a location where the circuit adjustment element 8 is connected is not limited to the connecting point of the resonator and the input/output line.

FIGS. 40A to 40D show examples of the connecting point of the circuit adjustment element 8. FIG. 40A shows an example in which the circuit adjustment element 8 is connected to the connecting point of the input/output line 7 and the variable resonator 20 in parallel with the variable resonator 20. FIG. 40B shows an example in which the circuit adjustment element 8 is connected to the input/output line 7 in parallel with the variable resonator 20, between the connecting point of the input/output line 7 and the variable resonator 20 and the port P1. FIG. 40C shows an example in which the circuit adjustment element 8 is inserted in series with the input/output line 7. FIG. 40D shows an example in which the circuit adjustment element 8 is connected between the ring-shaped line 2 and the ground, on the angular position of N π on the variable resonator 20. In this case, N represents an integer of at least 1. In FIG. 41B (will be described later), N represents 0.

FIG. 41 shows another connection example of the circuit adjustment element 8. FIG. 41A shows an example in which the input/output line 7 and the variable resonator 20 are connected to each other via the circuit adjustment element 8. FIG. 41B shows an example in which the circuit adjustment element 8 is disposed inside the ring-shaped line 2 and is connected between the ground and the connecting position of the ring-shaped line 2 and the input/output line 7.

FIG. 42 shows various examples of the circuit adjustment element 8.

FIG. 42A shows a capacitor acting as an individual element. FIG. 42B shows lines which form a gap in the same plane so as to act as a capacitor. FIG. 42C shows a multi-level line structure in which lines having different heights are opposed to each other with a dielectric interposed therebetween so as to act as a capacitor. FIG. 42D shows an inductor acting as an individual element. FIG. 42E shows a bent line acting as an inductor in a plane. FIG. 42F shows a spiral coil formed on a line. FIG. 42G shows a line inserted in series. FIG. 42H shows a line acting as an open-end stub.

This effect may be obtained without adding the circuit adjustment element 8. In this case, the input/output line 7 having the characteristic impedance Z₁different from the port impedance Z₀has a phase of 180° as shown in FIG. 43 or an integral multiple of the phase. This is because an input impedance viewed from the port P1 is always equal to the impedance of the port P2 due to the 180° line.

FIGS. 44 to 47 show structural examples of the variable bandwidth filer having the circuit adjustment element and also show the simulation results of the characteristics. In all the cases, the impedances of ports P1 and P2 are 50Ω, the impedance of an input/output line 7 is 60Ω, and two switches 3₁and 3₂are disposed on the position of 10° and the position of 90°. FIGS. 44B to 47B show characteristics when the switch 3₁is turned on, and FIGS. 44C to 47C show characteristics when the switch 3₂is turned on. Of these characteristics, a solid line indicates a transmission coefficient S21, a broken line indicates a reflection coefficient S11, and a thin line indicates characteristics in the absence of the circuit adjustment element 8.

FIG. 44A shows an example in which an open-end stub 8 is formed on the input/output line 7, on the position of a 10/360 wavelength at the resonance frequency from the connecting point between the input/output line 7 and the ring-shaped line 2 of the variable resonator. Even when switching a state in which the switch 3₁is turned on to a state in which the switch 3₂is turned on, the resonance frequency remains 5 GHz as shown in FIGS. 44B and 44C. However, when the stub 8 is not provided, the resonance frequency changes to 5.1 GHz as indicated by the thin line of FIG. 44C.

FIG. 45A shows an example in which a line having a length of a 7/360 wavelength at the resonance frequency is inserted as the circuit adjustment element 8 between the input/output line 7 and the ring-shaped line 2. Also in this example, the resonance frequency does not change from 5 GHz as shown in FIGS. 45B and 45C even when the switches 3₁and 3₂are selectively turned on.

FIG. 46A shows an example in which a line having a characteristic impedance of 57Ω is connected in series with the input end of the input/output line 7 as the circuit adjustment element 8. Also in this case, as is evident from FIGS. 45B and 45C, the resonance frequency does not change even when the switches 3₁and 3₂are switched.

FIG. 47A shows an example in which a capacitor of 0.08 pF is connected as the circuit adjustment element 8 between the input/output line 7 and the ground, instead of the open-end stub 8 of FIG. 44A. Also in this case, the resonance frequency does not change as shown in FIGS. 47B and 47C even when the switches 3₁and 3₂are selectively switched.

As described above, in all the examples, the function of the circuit adjustment element 8 makes the resonance frequency invariant regardless of the position of the switch having been turned on.

The variable resonators 20 of the foregoing embodiments enable direct grounding on different positions on the ring-shaped line 2 through the switches 3 or grounding through the passive element. An adjustment transmission line having desired characteristics may be connected via the switch 3. FIG. 48A shows the structural example.

FIG. 48A shows a modification of the variable bandwidth filter 10 shown in FIG. 13. As in FIG. 48A, a variable resonator 20 is inserted in series with an input/output line 7. Instead of grounding the ring-shaped conductor line 2 on a desired position via the switch 3, the ring-shaped conductor line 2 can be connected via the switch 3 to an adjustment transmission line 21 having desired characteristics. In this example, the electric length of each adjustment transmission line 21 is 75° at the center frequency of a used frequency band and the end of the adjustment transmission line 21 is opened.

FIG. 48B shows the frequency characteristics of a transmission coefficient when the switch 3 at θ=30° is turned on in FIG. 48A. In this example, unlike FIG. 14 showing the characteristics of the variable bandwidth filter of FIG. 13, two transmission zeros appear substantially symmetrically with respect to the resonance frequency of 5 GHz. These transmission zeros appear on both sides of the resonance frequency and thus it is possible to control attenuation characteristics on the high-frequency side and the low-frequency side of the resonance frequency. Although FIG. 48A shows the case where adjustment transmission lines 21 of the same electric length are connected to all the switches 3, adjustment transmission lines of desired electric lengths may be connected to the respective switches 3 depending on required characteristics. The same is true for the following embodiments.

FIG. 49A shows an example in which the electric length of each adjustment transmission line 21 of FIG. 48A is shortened to 50° and the end of the adjustment transmission line 21 is grounded via a capacitor 22. FIG. 49B shows the frequency characteristics of a transmission coefficient in this configuration. Also in this case, the switch 3 at θ=30° is turned on. The adjustment transmission line 21 connected to the switch 3 and the capacitor 22 connected to the end of the adjustment transmission line 21 are illustrated only for one of the switches 3, and the intermediate portions and ends of the other adjustment transmission lines 21 and the capacitors 22 connected to the ends of the other transmission lines 21 are not shown. Comparisons between FIGS. 49B and 48B prove that the passband widths around 5 GHz are the same. In other words, although the same passband width is obtainable, the electric lengths can be equivalently increased by grounding the ends of the adjustment transmission lines 21 through the capacitors 22. Accordingly, the electric length of the adjustment transmission line 21 can be reduced. In FIG. 49A, the electric length of each adjustment transmission line 21 and capacitance of the capacitor 22 may be set to desired values depending on required characteristics.

In the example of FIG. 50, a variable capacitance element 22′ is used instead of the capacitor 22 of FIG. 49A. However, the electric length of the adjustment transmission line 21 is not limited to 50°. With the adjustment transmission lines 21 and the variable capacitance elements 22′, it is possible to equivalently adjust electric lengths. In other words, it is possible to adjust the positions of the transmission zeros in FIG. 49B.

In the example of FIG. 51, on the end of each adjustment transmission line 21₁which correspond to the adjustment transmission line 21 of the example of FIG. 48A and has a desired electric length, an adjustment transmission line 21₂having a desired electric length is further connected via a switch 23. The electric length of the adjustment transmission line connected to the switch 3 can be changed by turning on/off the switch 23. Thus the positions of the transmission zeros of the frequency characteristics can be adjusted.

In the example of FIG. 52, at least two switches are provided on different positions including its end position along the length of each adjustment transmission line 21 connected to the switches 3 of FIG. 48A. In this example, three switches 23₁, 23₂and 23₃are provided to enable grounding. This configuration can also adjust the positions of the transmission zeros of the frequency characteristics. The electric length of the adjustment transmission line 21 is not limited to 75°. By turning on desired one of the switches 23₁, 23₂and 23₃, it is possible to select the case where the adjustment transmission line 21 is grounded with a desired electric length and the case where all the switches are turned off and the ends of the adjustment transmission lines 21 are opened without being grounded.

In FIG. 49A, the end of the adjustment transmission line 21 can be grounded through the capacitor 22, thereby reducing the electric length of the adjustment transmission line 21. As shown in FIG. 53, the adjustment transmission line 21 may not be connected and each switch 3 having one end connected to the ring-shaped line 2 may have the other end grounded directly through the capacitor 22. Also in this case, as in FIG. 49B, it is possible to obtain frequency characteristics having two transmission zeros near both sides of the resonance frequency.

FIGS. 48A, 49A, 50, 51, 52 and 53 show examples in which the variable resonator 20 is used to configure the variable bandwidth filter 10. These variable resonators 20 may be used in any of the variable resonators shown in FIGS. 5A, 5B, 7B, 7D, 8, 9, 10, 11A, 11B, 11C, 15, 16, 18, 19, 20A, 20B, 21A, 23A, 23B, 24, 26 to 29, 40A to 40D, 41A, 41B, 44A, 45A, 46A and 47A.

In FIGS. 49 to 53, the variable bandwidth filter has the variable resonator inserted in series with the input/output line as in the example of FIG. 13. Also in the variable bandwidth filter having the variable resonator connected in parallel with the input/output line, adjustment transmission lines may be connected to the switches 3 of the ring-shaped conductor line composing the variable resonator.

FIG. 54 shows an example in which instead of grounding one end of a switch 3 having the other end connected to a ring-shaped line 2, an adjustment transmission line 21 having an open end is connected to the one end of each switch 3, in the example in which the variable resonator 20 of FIG. 1A or 1B is connected in parallel with the input/output line 7. In this configuration, an electric length from the connecting point between the switch 3 and the ring-shaped conductor line 2 to the open end of the adjustment transmission line 21 is selected 90° (λ/4) at the used frequency. In FIG. 54, the adjustment transmission line 21 is shown only for one of the switches 3 and the illustration for the other switches 3 is omitted. With this configuration, a connecting point of desired one of the switches 3 and the ring-shaped conductor line 2 is equivalently grounded when the switch 3 is turned on, thereby avoiding the influence of a phase change caused by the structure of the switch 3 (for example, the length of the switch in the signal transmission direction). In contrast, in FIGS. 1A and 1B, a signal phase change occurs due to the structure from the connecting point of the ring-shaped conductor line 2 and the switch 3 having been turned on to a ground point. Hence, the configuration of FIG. 54 is effective for avoiding the influence of such a phase change.

FIG. 55 shows a modification of FIG. 54. An adjustment transmission line 21 whose end is short-circuited to the ground is connected to each switch 3. In this configuration, an electric length from the connecting point of the switch 3 and the ring-shaped conductor line to the short-circuited point on the end of the transmission line 21 is selected 180° (λ/2) at the frequency to be used. In this case, as in the case of FIG. 54, a connecting point of desired one of the switches 3 having been turned on and the ring-shaped conductor line 2 is equivalently grounded, thereby avoiding a signal phase change caused by the structure of the switch 3.

The open-end adjustment transmission line 21 or the adjustment transmission line 21 having the short-circuited end in FIGS. 54 and 55 can be used in the embodiments of FIGS. 5A, 5B, 8 to 11, 13, 15 to 21, 23, 24, 26 to 31, 38, 40, 41, and 43 to 47 as well as the examples of FIGS. 1A and 1B.

Number	Date	Country	Kind
2006-244707	Sep 2006	JP	national
2007-166362	Jun 2007	JP	national
2007-219967	Aug 2007	JP	national

Number	Name	Date	Kind
5162759	Yajima	Nov 1992	A
5400002	Takahashi et al.	Mar 1995	A
5659274	Takahashi et al.	Aug 1997	A
6452465	Brown et al.	Sep 2002	B1
6778023	Christensen	Aug 2004	B2
20030025563	Christensen	Feb 2003	A1
20050190018	Kawai et al.	Sep 2005	A1
20060028301	Kamata et al.	Feb 2006	A1
20060087388	Kawai et al.	Apr 2006	A1

Number	Date	Country
8-56104	Feb 1996	JP
2001-230602	Aug 2001	JP
2004-7352	Jan 2004	JP
2005-217852	Aug 2005	JP

Variable resonator, variable bandwidth filter, and electric circuit device

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