1. Field of the Invention
The present invention relates to a voltage controlled oscillator used in wireless communication equipment and PLL circuits. More specifically, the present invention relates to a voltage controlled oscillator in which phase noise can be suppressed.
2. Description of the Background Art
Voltage controlled oscillators are widely used as means for generating a local oscillating signal of wireless communication equipment.
In
The voltage controlled oscillator 900 oscillates in the vicinity of the resonance frequency of the parallel resonant circuit. Therefore, the oscillating frequency of the voltage controlled oscillator 900 can be set to a desired frequency by controlling the control voltage. The oscillating transistors 915 and 916 generate a negative resistance and cancel the loss due to a parasitic resistance component of the resonant circuit so as to satisfy the oscillating conditions.
It is known that in general, 1/f noise that is generated in the oscillating transistors 915 and 916 and the current source transistor 930 is converted to a frequency in the vicinity of the oscillating frequency by interaction with the voltage fluctuation due to the frequency 2fo at the source terminal P0. Therefore, the 1/f noise converted to a frequency in the vicinity of the oscillating frequency is superimposed on oscillating signals and becomes phase noise.
As shown in
The phase noise in the region A is caused by the 1/f noise generated in the oscillating transistors 915 and 916 and the current source transistor 930 being converted to a frequency in the vicinity of the oscillating frequency and then superimposed on oscillating signals. The transistor used for the oscillating transistor or the current source transistor is generally a field-effect transistor. The field-effect transistors have large 1/f noise. Therefore, in the conventional voltage control oscillators, occurrence of phase noise in the region A causes a large problem.
As a method to solve this problem, a voltage controlled oscillator disclosed in “CMOS Differential LC Oscillator with Suppressed Up-Converted Flicker Noise” written by Aly Ismail, Asad A. Abidi, 2003 IEEE International Solid-State Circuits Conference, pp. 98-99 (non-patent document 1) has been proposed.
In
In
The circuit made up of the coupling capacitor 920 and the inductors 918 and 919 serves to suppress the influence of the 1/f noise generated in the oscillating transistors 915 and 916. The inductors 918 and 919 have a sufficiently large impedance with respect to the oscillating frequency fo. The coupling capacitor 920 has a capacitance value that provides a high impedance with respect to the 1/f noise in the range that satisfies the oscillating conditions. With such impedance and capacitance value, the temporal change of the voltage shown in
In the circuit of
By suppressing the 2 fo component, conversion of the 1/f noise to a frequency in the vicinity of the oscillating frequency due to the interaction between the 1/f noise and the 2 fo component can be reduced, so that the phase noise in the vicinity of the oscillating frequency can be suppressed.
However, in the above configuration, two inductors having a large impedance (e.g., 23 nH in the non-patent document 1) with respect to the oscillating frequency are required. These inductors can be realized in a semiconductor integrated circuit by using spiral inductors, which occupy a large area. Therefore, the chip size of the voltage controlled oscillator is increased, resulting in an increase in the cost.
Furthermore, in the conventional configuration, a resistor is used in place of a current source transistor, so that, for example, when a power voltage is changed, for example, upon operating a transmitting circuit, the current value swings and the oscillating frequency is fluctuated.
Therefore, an object of the present invention is to provide a voltage controlled oscillator that has a low cost and has reduced fluctuations in the oscillating frequency.
The present invention has the following features to attain the object mentioned above. A first aspect of the present invention is directed to a voltage controlled oscillator for changing an oscillating frequency in accordance an input control voltage, including a parallel resonant circuit, a negative resistance circuit connected in parallel to the parallel resonant circuit, a first capacitor connected in parallel to the negative resistance circuit, a first current source connected between one terminal of the first capacitor and the negative resistance circuit, and a second current source connected between the other terminal of the first capacitor and the negative resistance circuit. The capacitance value of the first capacitor is a value that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the first capacitor. Each of the first and the second current sources is constituted by a bipolar transistor.
Thus, a signal with a frequency twice the oscillating frequency can be suppressed, so that the 1/f noise is prevented from being converted to a frequency in the vicinity of the oscillating frequency by the interaction between the second harmonic and the 1/f noise. Thus, the phase noise can be suppressed. Furthermore, the impedance as seen from both the terminals of the first capacitor to the ground is high, using the current sources, so that the circuit can be smaller than a conventional example using an inductor, which can increase the numbers of ICs that can be configured. Thus, low-cost voltage controlled oscillators can be provided. Moreover, the current source is constituted by a bipolar transistor, so that the current value can be stabilized, compared with the conventional technique in which the current source is constituted by a resistor and thus a voltage controlled oscillator having reduced fluctuations of the oscillating frequency can be provided.
For example, the negative resistance circuit includes a first oscillating transistor constituted by an field-effect transistor, a second oscillating transistor constituted by an field-effect transistor, and the parallel resonant circuit is connected between the drains of the first and the second oscillating transistors. The first capacitor is connected between the sources of the first and the second oscillating transistors. The gate of the first oscillating transistor is connected to the drain of the second oscillating transistor. The gate of the second oscillating transistor is connected to the drain of the first oscillating transistor. The collector of the bipolar transistor as the first current source is connected between the first capacitor and the source of the first oscillating transistor. The collector of the bipolar transistor as the second current source is connected between the first capacitor and the source of the second oscillating transistor.
Preferably, the voltage controlled oscillator further includes a first impedance element connected between the collector of the bipolar transistor as the first current source and the first capacitor, and a second impedance element connected between the collector of the bipolar transistor as the second current source and the first capacitor. The first and the second impedance elements have an impedance that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the first capacitor.
Thus, even if the oscillating frequency is high, the 1/f noise is prevented from being converted to a frequency in the vicinity of the oscillating frequency by the interaction between the second harmonic and the 1/f noise. Thus, a voltage controlled oscillator in which the phase noise can be suppressed can be provided.
For example, the first and the second impedance elements are inductors.
Furthermore, for example, the first and the second impedance elements are a LC parallel resonant circuit including an inductor and a capacitor, and a resonant frequency of the LC parallel resonant circuit is a frequency twice the oscillating frequency.
Thus, the size of the inductor can be further reduced.
Preferably, furthermore, the voltage controlled oscillator further includes a first impedance element connected between the emitter of the bipolar transistor as the first current source and the ground, and a second impedance element connected between the emitter of the bipolar transistor as the second current source and the ground. The first and the second impedance elements have an impedance that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the first capacitor.
For example, the first and the second impedance elements are inductors.
Preferably, the inductor is a bonding wire.
For example, the first and the second impedance elements are a LC parallel resonant circuit including an inductor and a capacitor. The resonant frequency of the LC parallel resonant circuit is a frequency twice the oscillating frequency. When the first and the second impedance elements are a LC parallel resonant circuit, the size of the inductor can be further reduced.
Preferably, the voltage controlled oscillator further includes a second capacitor, one end of which is connected a base of the bipolar transistor as the first current source and a base of the bipolar transistor as the second current source, and the other end of which is grounded.
Thus, the thermal noise can be short-circuited, so that a voltage controlled oscillator having better noise characteristics can be provided.
Preferably, the bipolar transistors constituting the first and the second current sources have a collector made of a n-type semiconductor, a base made of a p-type semiconductor and an emitter made of an n-type semiconductor, and the collector is formed inside the emitter.
Thus, the parasitic capacitance between the collector and the p-type substrate can be reduced, so that even if the oscillating frequency is high, the current source transistor side becomes a high impedance as desired.
Preferably, the first and the second oscillating transistors are field-effect transistors in which a drain electrode encloses a source electrode.
Thus, the parasitic capacitance between the source and the p-type substrate can be reduced, so that even if the oscillating frequency is high, a desired negative resistance occurs.
A second aspect of the present invention is directed to a phase locked loop (PLL) circuit for outputting a local oscillating signal, including a voltage controlled oscillator for changing an oscillating frequency in accordance a control signal and outputting the local oscillating signal, a feedback portion for feeding back the local oscillating frequency signal that is output from the voltage controlled oscillator, a phase comparator for generating the control signal in accordance with a phase difference between an input reference signal and a feedback signal that is output from the feedback portion; and a loop filter for extracting a low frequency component of the control signal generated by the phase comparator and inputting the component to the voltage controlled oscillator. The voltage controlled oscillator comprises a parallel resonant circuit, a negative resistance circuit connected in parallel to the parallel resonant circuit, a first capacitor connected in parallel to the negative resistance circuit, a first current source connected between one terminal of the first capacitor and the negative resistance circuit, and a second current source connected between the other terminal of the first capacitor and the negative resistance circuit. The capacitance value of the first capacitor is a value that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the first capacitor, and each of the first and the second current sources is constituted by a bipolar transistor.
Thus, a PLL circuit that outputs a local oscillating signal with a stable oscillating frequency can be provided.
A third aspect of the present invention is directed to wireless communication equipment for transmitting and receiving wireless signals, including a phase locked loop (PLL) circuit for outputting a local oscillating signal and a transmitting and receiving circuit for transmitting and receiving the wireless signals, using the local oscillating signal that is output from the PLL circuit. The PLL circuit comprises a voltage controlled oscillator for changing an oscillating frequency in accordance a control signal and outputting the local oscillating signal, a feedback portion for feeding back the local oscillating frequency signal that is output from the voltage controlled oscillator, a phase comparator for generating the control signal in accordance with a phase difference between an input reference signal and a feedback signal that is output from the feedback portion and a loop filter for extracting a low frequency component of the control signal generated by the phase comparator and inputting the component to the voltage controlled oscillator. The voltage controlled oscillator includes a parallel resonant circuit, a negative resistance circuit connected in parallel to the parallel resonant circuit, a first capacitor connected in parallel to the negative resistance circuit, a first current source connected between one terminal of the first capacitor and the negative resistance circuit, and a second current source connected between the other terminal of the first capacitor and the negative resistance circuit. The capacitance value of the first capacitor is a value that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the first capacitor, and each of the first and the second current sources is constituted by a bipolar transistor.
Thus, wireless communication equipment that can transmit and receive wireless signals based on a local oscillating signal having a stable oscillating frequency can be provided.
According to the present invention, a voltage controlled oscillator with small phase noise due to 1/f noise, low cost and reduced fluctuations of the oscillating frequency can be provided.
These and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings.
The variable capacitive elements 103 and 104 utilize the gate capacitance of variable capacitive diodes or field-effect transistors (FET). The oscillating transistors 105 and 106 are field-effect transistors (hereinafter, referred to as “FET”). The coupling capacitor 109 is a capacitor having a predetermined capacitance value. The current source transistor 110 and 111 are bipolar transistors.
The inductors 101 and 102 are connected in series. The power terminal 100 is connected between the inductors 101 and 102. The variable capacitive elements 103 and 104 are connected in series. The frequency control terminal 107 is connected between the variable capacitive elements 103 and 104. The series circuit including the inductors 101 and 102 and the series circuit including the variable capacitive elements 103 and 104 are connected in parallel so as to constitute a parallel resonant circuit.
The gate of the oscillating transistor 105 and the drain of the oscillating transistor 106 are connected to one end of the variable capacitive element 104. The gate of the oscillating transistor 106 and the drain of the oscillating transistor 105 are connected to one end of the variable capacitive element 103. The coupling capacitor 109 is connected between the source of the oscillating transistor 105 and the source of the oscillating transistor 106.
The collector of the current source transistor 110 is connected between the source of the oscillating transistor 105 and the coupling capacitor 109. The collector of the current source transistor 111 is connected between the oscillating transistor 106 and the coupling capacitor 109.
The base of the current source transistor 110 is connected to the base of the current source transistor 111. The emitter of the current source transistor 110 is grounded via the resistor 112. The emitter of the current source transistor 111 is grounded via the resistor 113.
A circuit for constituting a current mirror circuit is connected to the base of the current source transistor 110 and the base of the current source transistor 111.
In
The collector of the bipolar transistor 203 is connected to the power terminal 201 via the constant current source 202. The collector and the base of the bipolar transistor 203 are connected to each other. The base of the bipolar transistor 203 is connected to the base of the current source transistors 110 and 111. The emitter of the bipolar transistor is grounded via the resistor 204.
The constant current source 202 generates a reference current Iref. A current having a constant current value constantly flows from the collectors to the emitters of the current source transistors 110 and 111 in accordance with the current value of the reference current Iref, the size ratio of the bipolar transistor 203 and the current source transistors 110 and 111 and the size ratio of the resistor 204 and the resistors 112 and 113. Therefore, the current source transistors 110 and 111 are constant current sources.
The capacitance values of the variable capacitive elements 103 and 104 are adjusted in accordance with the control voltage applied to the frequency control terminal 107. Thus, the resonant frequency of a parallel resonant circuit is changed. Therefore, the oscillating frequency can be controlled by controlling the control voltage. The oscillating transistors 105 and 106 are negative resistance circuits. A negative resistance is generated by the oscillating transistors 105 and 106, and a loss due to a parasitic resistance component of a resonant circuit is cancelled, and the oscillation conditions are satisfied.
The impedance (denoted by Z0 in
Z0=ΔV/ΔI,
where ΔV is a voltage fluctuation of the source terminal P1 or P2, and ΔI is a current fluctuation thereof.
The current source transistors 110 and 111 operate so as to keep a constant current value with respect to the voltage fluctuation ΔV. Therefore, the current fluctuation ΔI is 0 with respect to the voltage fluctuation ΔV. Therefore, the impedance Z0 is high with respect to the voltage fluctuation of oscillating signals, that is, the oscillating frequency.
If the capacitance of the coupling capacitor 109 is set as appropriate, the terminals P1 and P2 are separated, and in the voltage fluctuations generated in the terminals P1 and P2, the 2 fo component can be reduced, as shown in
The capacitance value of the coupling capacitor 109 can be obtained in the following manner. First, the capacitance of the coupling capacitor 109 is set to an appropriate value that provides a high impedance with respect to the oscillating frequency. Next, the capacitance value of the coupling capacitor 109 is decreased, and the capacitance value that does not satisfy the oscillation conditions is obtained. Next, the capacitance value of the coupling capacitor 109 is a little increased, and the smallest capacitance value that satisfies the oscillation conditions is obtained. This obtained capacitance value is an optimal capacitance value of the coupling capacitor 109. The temporal waveforms of the voltages at the terminals P1 and P2 when this capacitive value is selected are those shown in
Therefore, the signal with a frequency of 2 fo is suppressed, so that the 1/f noise is suppressed from being converted to a frequency in the vicinity of the oscillating frequency by the interaction between the 1/f noise generated in the oscillating transistors 105 and 106 and the second harmonic. Thus, the phase noise can be suppressed.
In the present invention, since bipolar transistors in which the 1/f noise is small are used for the current source transistors 110 and 111, deterioration of the characteristics of phase noise caused by the current source transistors can be reduced.
Thus, in the first embodiment, the capacitance value of the coupling capacitor (first capacitor) is set to an appropriate value that provides a low impedance with respect to the oscillating frequency and a high impedance with respect to the 1/f noise so that a signal with a frequency of 2fo is suppressed at both the terminals of the coupling capacitor. Furthermore, current source transistors constituted by bipolar transistors are connected so that the ground side has a high impedance with respect to the oscillating frequency. Thus, the second harmonic can be suppressed, so that the 1/f noise is suppressed from being converted to a frequency in the vicinity of the oscillating frequency by the interaction between the 1/f noise generated in the oscillating transistors and the second harmonic. Thus, a voltage controlled oscillator with reduced phase noise can be provided.
Furthermore, the configuration in which the current sources using bipolar transistors allow the ground side to have a high impedance with respect to the oscillating frequency is adopted. Therefore, compared with a conventional case where a high impedance with respect to the oscillating frequency is generated by using two inductors occupying large areas, the chip area can be reduced significantly, and thus the cost of a voltage controlled oscillator can be reduced.
Furthermore, bipolar transistors are used as the constant current sources, so that the 1/f noise can be suppressed, compared with the case where FETs are used as the current sources, so that the phase noise can be further suppressed.
Furthermore, transistors are used as the constant current sources, so that fluctuations of the frequency due to the variation of the power voltage can be suppressed, compared with a conventional case where the constant current source is constituted by resistors.
A bypass capacitor may be connected in parallel between the base terminal of the current source transistor 110 and the base terminal of the current source transistor 111.
In this embodiment, the first and the second current sources are constituted by the current source transistors 110 and 111, respectively. However, each of the first and the second current sources may be constituted by two or more bipolar transistors. Also in the case where each of the first and the second current sources is constituted by two or more bipolar transistors, one bipolar transistor likewise serves as the current source.
In the first embodiment, the parasitic capacitance between the emitters and the collectors of the current source transistors 110 and 111 reduces the impedance Z0 as seen from the terminals P1 and P2 to the current source side with respect to higher oscillating frequencies. For example, in the case where the oscillating frequency is 5 GHz, this parasitic capacitance causes the impedance Z0 in the first embodiment to be a low impedance with respect to 10 GHz. Therefore, when the oscillating frequency is increased, in the first embodiment, a signal with a frequency twice the oscillating frequency cannot be suppressed at both the terminals of the coupling capacitor 109. Therefore, the effect of suppressing the phase noise cannot be obtained.
In the second embodiment, by inserting the inductors 120 and 121 having an impedance that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the coupling capacitor 109, the effect of suppressing the phase noise also can be obtained with respect to higher frequencies.
Such an impedance is obtained in the following manner. First, an appropriate inductor that provides a high impedance with respect to a frequency twice the oscillating frequency is connected, and the temporal change of the voltage at the terminals P1 and P2 is simulated. Then, the impedance values of the inductors 120 and 121 are changed so that an optimal impedance with which the signal with a frequency twice the oscillating frequency is suppressed is obtained. Thus, the impedances of the inductors 120 and 121 are obtained.
Thus, by connecting the inductors 120 and 121 having a large impedance at 2 fo, good phase noise characteristics can be realized even if the oscillating frequency is high.
Furthermore, the impedance ZL of an inductor is expressed by ZL=2 πfL, where L is an inductance, and f is a frequency, and is proportional to the frequency. Therefore, as the frequency is increased, the inductance can be reduced. In this embodiment, since it is sufficient that the inductors 120 and 121 provide a high impedance with respect to the frequency twice the oscillating frequency, the inductors 120 and 121 can be smaller than the inductors 918 and 919 used in the conventional configuration shown in
Thus, in the second embodiment, an inductor having a large impedance with respect to the second harmonic is provided, so that even if the oscillating frequency is high, the impedance Z0 can be kept high. Therefore, even if the oscillating frequency is high, good noise characteristics can be realized.
The inductors 120 and 121 can be realized with spiral inductors.
In place of the inductors 120 and 121, a LC parallel resonant circuit including an inductor and a capacitor can be used.
The impedance of the entire LC resonant circuit is a value that suppresses the signal with a frequency twice the oscillating frequency at both the terminals of the coupling capacitor 109, as described above.
Herein, when the inductance of the inductor is taken as L, and the capacitance of the capacitor is taken as C,
Therefore, according to the configuration in which the LC parallel resonant circuit is used, when the oscillating frequency is high, the inductance L is further reduced. Thus, even if the oscillating frequency is high, the chip area can be reduced significantly.
In the second embodiment, an inductor or a LC resonant circuit is used as an element connected between the current source transistors 110 and 111 and the coupling capacitor. However, the element is not limited thereto, as long as it is a first and second impedance element having a desired impedance.
Also in the second embodiment, as shown in
The inductors 132 and 133 are connected between the emitters of the current source transistors 110 and 111 and the ground. The inductors 132 and 133 have an impedance that suppresses a signal with a frequency twice the oscillating frequency at both the terminals of the coupling capacitor 109 similarly to the inductors 120 and 121 in the second embodiment.
Similarly to the second embodiment, by connecting the inductors 132 and 133 having a large impedance at 2 fo, even if the oscillating frequency is high, good phase noise characteristics can be realized.
Furthermore, similarly to the second embodiment, the impedance of the inductor is proportional to the frequency. Therefore, as the frequency is increased, the inductance can be reduced. In this embodiment, since it is sufficient that the inductors 132 and 133 provide a high impedance with respect to the frequency twice the oscillating frequency, the inductors 132 and 133 can be smaller than the inductors 918 and 919 used in the conventional configuration shown in
Thus, in the third embodiment, an inductor having a large impedance with respect to the second harmonic is provided, so that even if the oscillating frequency is high, the impedance in a current source portion can be kept high. Therefore, even if the oscillating frequency is high, good noise characteristics can be realized.
The inductors 132 and 133 can be realized with the spiral inductors or conductors shown in
Furthermore, the inductors 132 and 133 can be realized with bonding wires.
Thus, by realizing the inductors with wires, the circuit shown in
In the case where inductance is not sufficient with bonding wires, chip inductors may be connected to the lead pins 142 and 143 so as to supplement insufficient inductance.
In place of the inductors 132 and 133, as shown in
In the third embodiment, an inductor or a LC resonant circuit is used as the element connected between the current source transistors 110 and 111 and the ground. However, the element is not limited thereto, as long as it is a first and second impedance element having a desired impedance.
Also in the third embodiment, as shown in
In a fourth embodiment of the present invention, a voltage controlled oscillator that can provide a desired effect even with a high oscillating frequency will be described. In general, when configuring the circuit shown in
A significant influence of a parasitic capacitance 312 and a parasitic resistance 313 present between the n-type semiconductor 309 and the p-type substrate 311 on the source terminal 305 side appears, as the oscillating frequency is increased. The influence of the parasitic capacitance 312 and the parasitic resistance 313 prevents a desired high impedance from being obtained on the source side with high oscillating frequencies.
Furthermore, a significant influence of a parasitic capacitance 327 and a parasitic resistance 328 present between the n-type semiconductor 325 and the p-type substrate 326 on the collector terminal 322 side appears, as the oscillating frequency is increased. The influence of the parasitic capacitance 327 and the parasitic resistance 328 prevents a desired high impedance from being obtained on the current source transistors 110 and 111 side with high oscillating frequencies.
Therefore, in the fourth embodiment, a MOS transistor having a ring structure is used as the oscillating transistors 105 and 106, and an npn bipolar transistor that has a collector made of an n-type semiconductor, a base made of a p-type semiconductor and an emitter made of an n-type semiconductor and in which the collector is formed inside the emitter is used as the current source transistors 110 and 111. It should be noted that only the oscillating transistors 105 and 106 may be constituted by the MOS transistors having a ring structure, or that only the current source transistors 110 and 111 may be constituted by the npn bipolar transistor in which the collector is formed inside the emitter.
The voltage controlled oscillators that have been described in the first to the fourth embodiment may be used as a local oscillator in a PLL (phase locked loop) circuit.
The voltage controlled oscillator 340 changes the oscillating frequency in accordance with the input control signal and outputs it as a local oscillating signal. The frequency-divider 341 is a feedback portion that divides the local oscillating signal that is output from the voltage controlled oscillator 340, and feeds it back to the phase comparator 342 as a feedback signal. A reference signal is input to the reference signal input terminal 344. The reference signal input to the reference signal input terminal 344 is input to the phase comparator 342. The phase comparator 342 generates a control signal in accordance with the phase difference between the reference signal and the feedback signal and outputs the control signal. The loop filter 343 extracts a low frequency component of the control signal that is output from the phase comparator 342, and inputs it to the voltage controlled oscillator 340. Thus, the PLL circuit 5 can output stable local oscillating signals by synthesizing the local oscillating signals with the phase of the reference signal.
The frequency-divider 341 may be a feedback portion that feeds the local oscillating signal that is output from the voltage controlled oscillator 340 as a feedback signal back to the phase comparator 342 directly without dividing the local oscillating signal.
The voltage controlled oscillator that have been described in the first to the fourth embodiment may be used as a local oscillator in wireless communication equipment.
The PLL circuit 350 has the same configuration as that of the PLL circuit 5 shown in the fifth embodiment and outputs local oscillating signals. The transmitting circuit 351 generates an up-converted transmitting signal based on the local oscillating signal that is output from the PLL circuit 350 and outputs it as a wireless signal via a duplexer 354 and an antenna 353. The receiving circuit 352 down-converts the wireless signal that is received via the antenna 353 and the duplexer 354, based on the local oscillating signal that is output from the PLL circuit 350 for signal-receiving processing. Thus, the wireless communication equipment 6 can transmit and receive wireless signals based on stable local oscillating signals.
If a transmitting and receiving circuit that can transmit and receive wireless signals is provided, the configuration of the wireless communication circuit 6 is not limited to the configuration shown in
In the example shown in
The voltage controlled oscillator of the present invention has small phase noise, a low cost and reduced fluctuations of the oscillating frequency, and can be applied to a semiconductor integrated circuit, a PLL circuit, wireless communication equipment or the like.
While the invention has been described in detail, the foregoing description is in all aspects illustrative and not restrictive. It is understood that numerous other modifications and variations can be devised without departing from the scope of the invention.
Number | Date | Country | Kind |
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2004-011936 | Jan 2004 | JP | national |