BACKGROUND
The present invention relates to a comparator design, and more particularly, to a wide input range and low noise comparator with trigger timing control and/or gain boosting.
Analog-to-digital converters (ADCs) are widely used in many applications. In these ADCs, comparators play important roles. The comparator is a circuit which is used to compare two analog input signals and generates an analog output signal indicative of a comparison result of two analog input signals. The comparator may be regarded as a one-bit ADC. In the application of a power management integrated circuit (PMIC), the input voltage range is much wider than that in a legacy ADC. The typical two-stage comparators and single-stage comparators are not suitable due to higher offset variation and worse noise performance under a wider input voltage range.
Taking a typical two-stage comparator for example, as an input voltage level is high and near the supply voltage, at the moment when the output of the first stage has a maximum gain, the second stage does not trigger due to its input voltage level being too high. Once the second stage starts to trigger, the large gain from the first stage vanishes, resulting in higher offset and bad noise performance from the typical two-stage comparator.
Taking a typical single-stage comparator for example, as an input voltage level is high and near the supply voltage, at the moment when the output of the first sub-stage has a maximum gain, the second sub-stage does not trigger due to its input voltage level being too low. Once the second sub-stage starts to trigger, the input voltage level of the second sub-stage starts to drop and the current gain of the first sub-stage is already reduced, resulting in higher offset and bad noise performance from the typical single-stage comparator.
Thus, there is a need for an innovative comparator design which is capable of achieving a suitable offset level and a suitable noise level across the whole wide input range.
SUMMARY
One of the objectives of the claimed invention is to provide a wide input range and low noise comparator with trigger timing control and/or gain boosting.
According to a first aspect of the present invention, an exemplary multi-stage comparator is disclosed. The exemplary multi-stage comparator includes a first stage circuit, a second stage circuit, and a control circuit. The first stage circuit is arranged to receive an input signal of the multi-stage comparator, generate a first-stage output signal according to the input signal, and output the first-stage output signal at an output port of the first stage circuit. The second stage circuit is arranged to receive a second-stage input signal at an input port of the second stage circuit, and perform a second-stage operation to generate an output signal of the multi-stage comparator. The control circuit is coupled between the output port of the first stage circuit and the input port of the second stage circuit, and arranged to control a start time of the second-stage operation.
According to a second aspect of the present invention, an exemplary single-stage comparator is disclosed. The exemplary single-stage comparator includes a first sub-stage circuit, a second sub-stage circuit, and a control circuit. The first sub-stage circuit is arranged to receive an input signal of the single-stage comparator, generate a first-sub-stage output signal according to the input signal, and output the first-sub-stage output signal at an output port of the first sub-stage circuit. The second sub-stage circuit is arranged to receive a second-sub-stage input signal at an input port of the second sub-stage circuit, and perform a second-sub-stage operation upon the second-sub-stage input signal to generate an output signal of the single-stage comparator. The control circuit is coupled between the output port of the first sub-stage circuit and the input port of the second sub-stage circuit, and arranged to control a start time of the second-sub-stage operation.
These and other objectives of the present invention will no doubt become obvious to those of ordinary skill in the art after reading the following detailed description of the preferred embodiment that is illustrated in the various figures and drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagram illustrating a multi-stage comparator according to an embodiment of the present invention.
FIG. 2 is a diagram illustrating a first control circuit design according to an embodiment of the present invention.
FIG. 3 is a diagram illustrating a second control circuit design according to an embodiment of the present invention.
FIG. 4 is a diagram illustrating a second stage circuit with gate input according to an embodiment of the present invention.
FIG. 5 is a diagram illustrating a second stage circuit with drain input according to an embodiment of the present invention.
FIG. 6 is a diagram illustrating a single-stage comparator according to an embodiment of the present invention.
FIG. 7 is a diagram illustrating a third control circuit design according to an embodiment of the present invention.
FIG. 8 is a diagram illustrating a fourth control circuit design according to an embodiment of the present invention.
DETAILED DESCRIPTION
Certain terms are used throughout the following description and claims, which refer to particular components. As one skilled in the art will appreciate, electronic equipment manufacturers may refer to a component by different names. This document does not intend to distinguish between components that differ in name but not in function. In the following description and in the claims, the terms “include” and “comprise” are used in an open-ended fashion, and thus should be interpreted to mean “include, but not limited to . . . ”. Also, the term “couple” is intended to mean either an indirect or direct electrical connection. Accordingly, if one device is coupled to another device, that connection may be through a direct electrical connection, or through an indirect electrical connection via other devices and connections.
FIG. 1 is a diagram illustrating a multi-stage comparator according to an embodiment of the present invention. For better compensation of technical features of the present invention, the following uses a two-stage comparator 100 as an example of the multi-stage comparator. In this embodiment, the two-stage comparator 100 includes, but is not limited to, a first stage circuit 102, a second stage circuit 104, and a control circuit 106. The first stage circuit 102 is arranged to receive an input signal (which is a differential signal consisting of VIN and VIP) of the two-stage comparator 100, generate a first-stage output signal (which is a differential signal consisting of S1N and S1P) according to the input signal (VIN, VIP), and output the first-stage output signal (S1N, S1P) at an output port (which includes nodes n3 and n4) of the first stage circuit 102. The second stage circuit 104 is arranged to receive a second-stage input signal (which is a differential signal consisting of S2P and S2N) at an input port (which includes nodes n5 and n6) of the second stage circuit 104, and perform a second-stage operation (e.g., a latch operation) to generate an output signal (which is a differential signal consisting of VOUTN and VOUTP) of the two-stage comparator 100. For example, the first stage circuit 102 may be a preamplifier, and the second stage circuit 104 may be a latch circuit.
The control circuit 106 is added to enhance offset/noise performance of the two-stage comparator 100. Specifically, the control circuit 106 is coupled between the output port (n3, n4) of the first stage circuit 102 and the input port (n5, n6) of the second stage circuit 104, and is arranged to control a start time (i.e., trigger timing) of the second-stage operation of the second stage circuit 104. For example, the control circuit 106 may trigger the second-stage operation of the second stage circuit 104 at the time when the first stage circuit 102 has the maximum gain (e.g., an input transistor of the first stage circuit 102 operates in a saturation region) under a condition that VIN/VIP is high and near the supply voltage VDD, thereby minimizing the noise/offset contribution of the second stage circuit 104. In this way, the two-stage comparator 100 is capable of achieving a suitable offset level and a suitable noise level across the whole wide input range of the input signal VIN/VIP. Furthermore, since the two-stage comparator 100 has a wide input range when used by a particular application such as a PMIC application, it is possible that an input transistor may leave a saturation region and enter a triode region to have a smaller gain during a period in which VIN/VIP is high and near the supply voltage VDD. To address this issue, the control circuit 106 may further employ a gain boosting technique for enhancing the gain of the first stage circuit 102 to accommodate a large input range. The noise performance of the two-stage comparator 100 can be further enhanced due to the boosted gain of the first stage circuit 102. It should be noted that the use of the gain boosting technique may be optional. For example, when the two-stage comparator 100 has a narrow input range when used by a particular application, the gain boosting technique is not necessarily needed by the control circuit 106.
The two-stage comparator 100 may be a dynamic comparator (i.e., a clocked comparator) that operates under control of a clock signal RSTB. For example, the clock signal RSTB may be an internal clock of the two-stage comparator 100. For another example, the clock signal RSTB may be an external clock of the two-stage comparator 100. Specifically, the two-stage comparator 100 operates in one of a normal mode (RSTB=1) and a reset mode (RSTB=0) under control of the clock signal RSTB. In this embodiment, the control circuit 106 may further receive the clock signal RSTB, and refer to at least the clock signal RSTB to control the start time of the second-stage operation of the second stage circuit 104. In some embodiments of the present invention, the control circuit 106 may employ a sensing circuit and/or a delay circuit to control the start time of the second-stage operation of the second stage circuit 104. Further details are described as below with reference to the accompanying drawings.
FIG. 2 is a diagram illustrating a first control circuit design according to an embodiment of the present invention. The control circuit 106 shown in FIG. 1 may be implemented using the control circuit 200 shown in FIG. 2. The control circuit 200 includes a trigger signal generator circuit 202, a timing control circuit 204, and an optional gain boosting circuit 206. The trigger signal generator circuit 202 is arranged to generate a trigger signal RST according to the clock signal RSTB. In this embodiment, the trigger signal generator circuit 202 includes a sensing circuit 203 that is arranged to sense magnitude of the first-stage output signal (S1N, S1P) at the output port (n3, n4) of the first stage circuit 102. As shown in FIG. 1, when the two-stage comparator 100 operates in the reset mode (RST=1 & RSTB=0), the P-type transistors controlled by RSTB=0 are turned on, the N-type transistor controlled by RSTB=0 is turned off, and the N-type transistors controlled by RST=1 are turned on, such that both of S1N and S1P are set by the logic high level “1” due to the supply voltage VDD, and both of VOUTN and VOUTP are set by the logic low level “0” due to the ground voltage GND. When the clock signal RSTB has a level transition from the logic low level “0” to the logic high level “1”, the two-stage comparator 100 leaves the reset mode and enters the normal mode. In this embodiment, when the two-stage comparator 100 operates in the normal mode (RSTB=1), the trigger signal RST (RST=0) is derived from an output signal of the sensing circuit 203. When the clock signal RSTB has the logic high level “1”, an output of the inverter INV1 has a logic low level “0”. Hence, the N-type transistor M3 is turned off. Since the inverter INV2 is disconnected from the ground voltage GND, the input voltage of the inverter INV2 is controlled by the output signal of the sensing circuit 203. As shown in FIG. 2, the sensing circuit 203 includes two P-type transistors M1 and M2, where the gate terminal of the P-type transistor M1 is coupled to the node n4, and the gate terminal of the P-type transistor M2 is coupled to the node n3. The output signal of the sensing circuit 203 is determined by the drain voltage of a turned-on P-type transistor M1/M2. When the two-stage comparator 100 operates in the normal mode (RSTB=1), the P-type transistors controlled by RSTB=1 are turned off and the N-type transistor controlled by RSTB=1 is turned on, such that the first stage circuit 102 operates in the amplifier mode for setting the first-stage output signal S1N/S1P according to the input signal VIP/VIN. Hence, one of S1N and S1P is pulled low by the ground voltage GND through a turned-on N-type transistor controlled by one of VIN and VIP, which in turn turns on one of the P-type transistors M1 and M2 of the sensing circuit 203. The input voltage of the inverter INV2 increases correspondingly. When the input voltage of the inverter INV2 reaches a threshold voltage of the inverter INV2, an output voltage of the inverter INV2 has a level transition from the logic high level “1” to the logic low level “0”, thus resulting in RST=0 that may occur before the turned-on N-type transistor controlled by one of VIN and VIP enters the triode region. That is, RST=0 may be generated at the time when the first stage circuit 102 has the maximum gain.
In this embodiment, the timing control circuit 204 is coupled to the output port (n3, n4) of the first stage circuit 102 and the input port (n5, n6) of the second stage circuit 104, and is arranged to receive the trigger signal RST and trigger the second-stage operation of the second stage circuit 104 according to the trigger signal RST. When the trigger signal RST is set by the logic low level “0”, the P-type transistor M4 is turned on, thus allowing the supply voltage VDD to be supplied to one of the nodes n5 and n6 through one of the P-type transistors M5 and M6, where the gate terminal of the P-type transistor M5 is coupled to the node n3, and the gate terminal of the P-type transistor M6 is coupled to the node n4. When the two-stage comparator 100 operates in the normal mode (RSTB=1), the N-type transistors controlled by RST=0 are turned off, such that the second stage circuit 104 operates in the latch mode for setting the output signal (VOUTN, VOUTP) according to the supply voltage VDD that is supplied to one of nodes n5 and n6 under control of S1N at node n3 and S1P at node n4.
In this embodiment, the inverter INV2 may also be regarded as a delay circuit arranged to receive the output signal of the sensing circuit 203, and apply a predetermined delay to the output signal of the sensing circuit 203, wherein the trigger signal RST is derived from an output signal of the delay circuit.
With the help of the sensing circuit 203 and/or the inverter INV2, the second-stage operation of the second stage circuit 104 can be triggered at a proper moment (e.g., a moment when the first stage circuit 102 operates in the saturation region and has the maximum gain) to minimize the noise/offset contribution of the second stage circuit 104, thereby achieving better offset/noise performance for the two-stage comparator 100.
The gain boosting circuit 206 is coupled to the first stage circuit 102, and is arranged to enhance the gain of the first stage circuit 102. In this embodiment, the gain boosting circuit 206 is implemented by a cross-coupled load with two P-type transistors M7 and M8, where the gate terminal of the P-type transistor M7 is coupled to the node n3, and the gate terminal of the P-type transistor M8 is coupled to the node n4. With the help of the gain boosting circuit 206, the noise performance of the two-stage comparator 100 that operates under a wide input range can be further enhanced. It should be note that any gain boosting technique can be employed by the gain boosting circuit 206. That is, the present invention has no limitations on the design of the gain boosting circuit 206.
It should be noted that the circuit design of the trigger signal generator circuit 202 shown in FIG. 2 is for illustrative purposes only, and is not meant to be a limitation of the present invention. Alternatively, a trigger signal generator circuit may achieve the same objective of generating the trigger signal RST without using any sensing circuit.
FIG. 3 is a diagram illustrating a second control circuit design according to an embodiment of the present invention. The control circuit 106 shown in FIG. 1 may be implemented using the control circuit 300 shown in FIG. 3. The major difference between the control circuits 200 and 300 is the trigger signal generator circuit 302 used by the control circuit 300. As shown in FIG. 3, the trigger signal generator circuit 302 does not use any sensing circuit, and includes a delay circuit 304 that is arranged to receive the clock signal RSTB, and apply a predetermined delay to the clock signal RSTB. When the two-stage comparator 100 operates in the normal mode (RSTB=1), the trigger signal RST (RST=0) is derived from an output signal of the delay circuit 304. As mentioned above, when the two-stage comparator 100 operates in the reset mode (RST=1 & RSTB=0), the P-type transistors controlled by RSTB=0 are turned on, the N-type transistor controlled by RSTB=0 is turned off, and the N-type transistors controlled by RST=1 are turned on, such that both of S1N and S1P are set by the logic high level “1” due to the supply voltage VDD, and both of VOUTN and VOUTP are set by the logic low level “0” due to the ground voltage GND. When the clock signal RSTB has a level transition from the logic low level “0” to the logic high level “1”, the two-stage comparator 100 leaves the reset mode and enters the normal mode. When the clock signal RSTB has the logic high level “1”, the input voltage of the inverter INV2 is pulled high by a delayed version of the clock signal RSTB (RSTB=1). When the input voltage of the inverter INV2 reaches a threshold voltage of the inverter INV2, an output voltage of the inverter INV2 has a level transition from the logic high level “1” to the logic low level “0”, thus resulting in RST=0. With a proper setting of the predetermined delay possessed by the delay circuit 304, RST=0 may occur before the turned-on N-type transistor controlled by one of VIN and VIP enters the triode region. That is, RST=0 may be generated at the time when the first stage circuit 102 has the maximum gain. The timing control circuit 204 triggers the second-stage operation of the second stage circuit 104 according to the trigger signal RST (RST=0).
To put it simply, with a proper setting of the predetermined delay possessed by the delay circuit 304, the second-stage operation of the second stage circuit 104 can be triggered at a proper moment (e.g., a moment when the first stage circuit 102 operates in the saturation region and has the maximum gain) to minimize the noise/offset contribution of the second stage circuit 104, thereby achieving better offset/noise performance for the two-stage comparator 100.
As shown in FIGS. 1-3, the timing control circuit 204 is coupled to the input port (which includes nodes n5 and n6) of the second stage circuit 104, where nodes n5 and n6 are non-gate terminals (e.g., source terminals) of an input transistor pair of the second stage circuit 104. Hence, the timing control circuit 204 triggers the second-stage operation of the second stage circuit 104 through any of the non-gate terminals (e.g., source terminals) of the input transistor pair. However, this is for illustrative purposes only, and is not meant to be a limitation of the present invention. In some embodiments of the present invention, the input port of the second stage circuit 104 may be at drain terminals of an input transistor pair or gate terminals of an input transistor pair.
FIG. 4 is a diagram illustrating a second stage circuit with gate input according to an embodiment of the present invention. The second stage circuit 104 shown in FIG. 1 may be replaced with the second stage circuit 400 shown in FIG. 4. In this embodiment, the input port of the second stage circuit 400 includes nodes n5 and n6 that are gate terminals of an input transistor pair, such that the timing control circuit 204 triggers the second-stage operation of the second stage circuit 400 through any of the gate terminals of the input transistor pair.
FIG. 5 is a diagram illustrating a second stage circuit with drain input according to an embodiment of the present invention. The second stage circuit 104 shown in FIG. 1 may be replaced with the second stage circuit 500 shown in FIG. 5. In this embodiment, the input port of the second stage circuit 500 include nodes n5 and n6 that are drain terminals of an input transistor pair, such that the timing control circuit 204 triggers the second-stage operation of the second stage circuit 500 through any of the drain terminals of the input transistor pair.
The proposed control circuit (which may use a sensing circuit and/or a delay circuit) can be added to the two-stage comparator to control a start time of the second-stage operation (e.g., latch operation) of the second stage circuit, thereby enabling the two-stage comparator to achieve a suitable offset level and a suitable noise level across the whole wide input range. It should be noted that the same concept may be applied to a single-stage comparator.
FIG. 6 is a diagram illustrating a single-stage comparator according to an embodiment of the present invention. In this embodiment, the single-stage comparator 600 includes, but is not limited to, a first sub-stage circuit 602, a second sub-stage circuit 604, and a control circuit 606. The first sub-stage circuit 602 is arranged to receive an input signal (which is a differential signal consisting of VIN and VIP) of the single-stage comparator 600, generate a first-sub-stage output signal (which is a differential signal consisting of S1N and S1P) according to the input signal (VIN, VIP), and output the first-sub-stage output signal (S1N, S1P) at an output port (which includes nodes n3 and n4) of the first sub-stage circuit 602. The second sub-stage circuit 604 is arranged to receive a second-sub-stage input signal (which is a differential signal consisting of S2P and S2N) at an input port (which includes nodes n5 and n6) of the second sub-stage circuit 604, and perform a second-sub-stage operation (e.g., a latch operation) to generate an output signal (which is a differential signal consisting of VOUTN and VOUTP) of the single-stage comparator 600. For example, the first sub-stage circuit 602 may be a preamplifier, and the second sub-stage circuit 604 may be a latch circuit.
The control circuit 606 is added to enhance offset/noise performance of the single-stage comparator 600. Specifically, the control circuit 606 is coupled between the output port (n3, n4) of the first sub-stage circuit 602 and the input port (n5, n6) of the second sub-stage circuit 604, and is arranged to control a start time (i.e., trigger timing) of the second-sub-stage operation of the second sub-stage circuit 604. For example, the control circuit 606 may trigger the second-sub-stage operation of the second sub-stage circuit 604 at the time the first sub-stage circuit 602 has the maximum gain (e.g., an input transistor of the first sub-stage circuit 602 operates in a saturation region) under a condition that VIN/VIP is high and near the supply voltage VDD, thereby minimizing the noise/offset contribution of the second sub-stage circuit 604. In this way, the single-stage comparator 600 is capable of achieving a suitable offset level and a suitable noise level across the whole wide input range of the input signal (VIN, VIP). Furthermore, since the single-stage comparator 600 has a wide input range when used by a particular application such as a PMIC application, it is possible that an input transistor may leave a saturation region and enter a triode region to have a smaller gain during a period in which VIN/VIP is high and near the supply voltage VDD. To address this issue, the control circuit 606 may further employ a gain boosting technique for enhancing the gain of the first sub-stage circuit 602 to accommodate a large input range. The noise performance of the single-stage comparator 600 can be further enhanced due to the boosted gain of the first sub-stage circuit 602. It should be noted that the use of the gain boosting technique may be optional. For example, when the single-stage comparator 600 has a narrow input range when used by a particular application, the gain boosting technique is not necessarily needed by the control circuit 606.
The single-stage comparator 600 may be a dynamic comparator (i.e., a clocked comparator) that operates under control of a clock signal RSTB. For example, the clock signal RSTB may be an internal clock of the single-stage comparator 600. For another example, the clock signal RSTB may be an external clock of the single-stage comparator 600. Specifically, the single-stage comparator 600 operates in one of a normal mode (RSTB=1) and a reset mode (RSTB=0) under control of the clock signal RSTB. In this embodiment, the control circuit 606 may further receive the clock signal RSTB, and refer to at least the clock signal RSTB to control the start time of the second-sub-stage operation of the second sub-stage circuit 604. In some embodiments of the present invention, the control circuit 606 may employ a sensing circuit and/or a delay circuit to control the start time of the second-sub-stage operation of the second sub-stage circuit 604. Further details are described as below with reference to the accompanying drawings.
FIG. 7 is a diagram illustrating a third control circuit design according to an embodiment of the present invention. The control circuit 606 shown in FIG. 6 may be implemented using the control circuit 700 shown in FIG. 7. The control circuit 700 includes a trigger signal generator circuit 702, a timing control circuit 704, and an optional gain boosting circuit 706. The trigger signal generator circuit 702 is arranged to generate a trigger signal RSTB_dly according to the clock signal RSTB. In this embodiment, the trigger signal generator circuit 702 includes a sensing circuit 703 that is arranged to sense magnitude of the first-sub-stage output signal (S1P, S1N) at the output port (n3, n4) of the first sub-stage circuit 602. As shown in FIG. 6, when the single-stage comparator 600 operates in the reset mode (RSTB=0), the P-type transistors controlled by RSTB=0 are turned on, and the N-type transistor controlled by RSTB=0 is turned off, such that both of S1N and S1P are set by the logic high level “1” due to the supply voltage VDD, and both of VOUTN and VOUTP are set by the logic high level “1” due to the supply voltage VDD. When the clock signal RSTB has a level transition from the logic low level “0” to the logic high level “1”, the single-stage comparator 600 leaves the reset mode and enters the normal mode. In this embodiment, when the single-stage comparator 600 operates in the normal mode (RSTB=1), the trigger signal RSTB_dly (RSTB_dly=1) is derived from an output signal of the sensing circuit 703. When the clock signal RSTB has the logic high level “1”, an output of the inverter INV1 has a logic low level “0”. Hence, the N-type transistor M3 is turned off. Since the inverter INV2 is disconnected from the ground voltage GND, the input voltage of the inverter INV2 is controlled by the output signal of the sensing circuit 703. As shown in FIG. 7, the sensing circuit 703 includes two P-type transistors M1 and M2, where the gate terminal of the P-type transistor M1 is coupled to the node n4, and the gate terminal of the P-type transistor M2 is coupled to the node n3. The output signal of the sensing circuit 703 is determined by the drain voltage of a turned-on P-type transistor M1/M2. When the single-stage comparator 600 operates in the normal mode (RSTB=1), the N-type transistor controlled by RSTB=1 is turned on, such that the first sub-stage circuit 602 operates in the amplifier mode for setting the first-sub-stage output signal S1N/S1P according to the input signal VIP/VIN. Hence, one of S1N and S1P is pulled low by the ground voltage GND through a turned-on N-type transistor controlled by one of VIN and VIP, which in turn turns on one of the P-type transistors M1 and M2 of the sensing circuit 703. The input voltage of the inverter INV2 increases correspondingly. When the input voltage of the inverter INV2 reaches a threshold voltage of the inverter INV2, an output voltage of the inverter INV2 has a level transition from the logic high level “1” to the logic low level “0”, thus resulting in RST=0. Since RST=0 is provided to the inverter INV3, an output voltage of the inverter INV3 has a level transition from the logic high level “0” to the logic low level “1”, thus resulting in RSTB_dly=0 that may occur before the turned-on N-type transistor controlled by one of VIN and VIP enters the triode region. That is, RSTB_dly=1 may be generated at the time when the first sub-stage circuit 602 has the maximum gain.
In this embodiment, the timing control circuit 704 is coupled to the output port (n3, n4) of the first sub-stage circuit 602 and the input port (n5, n6) of the second sub-stage circuit 604, and is arranged to receive the trigger signal RSTB_dly and trigger the second-sub-stage operation of the second sub-stage circuit 604 according to the trigger signal RSTB_dly. When the trigger signal RSTB_dly is set by the logic high level “1”, the ground voltage GND is supplied to one of the nodes n5 and n6 through a turned-on N-type transistor controlled by one of VIN and VIP. When the single-stage comparator 600 operates in the normal mode (RSTB=1), the P-type transistors controlled by RSTB=1 are turned off, such that the second sub-stage circuit 604 operates in the latch mode for setting the output signal (VOUTN, VOUTP) according to the ground voltage GND that is supplied to one of nodes n5 and n6 through the timing control circuit 704.
In this embodiment, the inverters INV2 and INV3 may also be collectively regarded as a delay circuit arranged to receive the output signal of the sensing circuit 703, and apply a predetermined delay to the output signal of the sensing circuit 703, wherein the trigger signal RSTB_dly is derived from an output signal of the delay circuit.
With the help of the sensing circuit 703 and/or the inverters INV2 and INV3, the second-sub-stage operation of the second sub-stage circuit 604 can be triggered at a proper moment (e.g., a moment when the first sub-stage circuit 602 operates in the saturation region and has the maximum gain) to minimize the noise/offset contribution of the second sub-stage circuit 604, thereby achieving better offset/noise performance for the single-stage comparator 600.
The gain boosting circuit 706 is coupled to the first sub-stage circuit 602, and is arranged to enhance the gain of the first sub-stage circuit 602. In this embodiment, the gain boosting circuit 706 is implemented by a cross-coupled load with two P-type transistors M7 and M8, where the gate terminal of the P-type transistor M7 is coupled to the node n3, and the gate terminal of the P-type transistor M8 is coupled to the node n4. With the help of the gain boosting circuit 706, the noise performance of the single-stage comparator 600 that operates under a wide input range can be further enhanced. It should be note that any gain boosting technique can be employed by the gain boosting circuit 706. That is, the present invention has no limitations on the design of the gain boosting circuit 706.
It should be noted that the circuit design of the trigger signal generator circuit 702 shown in FIG. 7 is for illustrative purposes only, and is not meant to be a limitation of the present invention. Alternatively, a trigger signal generator circuit may achieve the same objective of generating the trigger signal RSTB_dly without using any sensing circuit.
FIG. 8 is a diagram illustrating a fourth control circuit design according to an embodiment of the present invention. The control circuit 606 shown in FIG. 6 may be implemented using the control circuit 800 shown in FIG. 8. The major difference between the control circuits 700 and 800 is the trigger signal generator circuit 802 used by the control circuit 800. As shown in FIG. 8, the trigger signal generator circuit 802 does not use any sensing circuit, and includes a delay circuit 804 that is arranged to receive the clock signal RSTB, and apply a predetermined delay to the clock signal RSTB. When the single-stage comparator 600 operates in the normal mode (RSTB=1), the trigger signal RSTB_dly (RSTB_dly=0) is derived from an output signal of the delay circuit 804. When the clock signal RSTB has a level transition from the logic low level “0” to the logic high level “1”, the single-stage comparator 600 leaves the reset mode and enters the normal mode. When the clock signal RSTB has the logic high level “1”, the input voltage of the inverter INV2 is pulled high by a delayed version of the clock signal RSTB (RSTB=1). When the input voltage of the inverter INV2 reaches a threshold voltage of the inverter INV2, an output voltage of the inverter INV2 has a level transition from the logic high level “1” to the logic low level “0”, thus resulting in RST=0. Since RST=0 is provided to the inverter INV3, an output voltage of the inverter INV3 has a level transition from the logic high level “0” to the logic low level “1”, thus resulting in RSTB_dly=0. With a proper setting of the predetermined delay possessed by the delay circuit 804, RSTB_dly=1 may occur before the turned-on N-type transistor controlled by one of VIN and VIP enters the triode region. That is, RSTB_dly=1 may be generated at the time when the first sub-stage circuit 602 has the maximum gain. The timing control circuit 204 triggers the second-stage operation of the second sub-stage circuit 604 according to the trigger signal RSTB_dly (RSTB_dly=1).
To put it simply, with a proper setting of the predetermined delay possessed by the delay circuit 804, the second-sub-stage operation of the second sub-stage circuit 604 can be triggered at a proper moment (e.g., a moment when the first sub-stage circuit 602 operates in the saturation region and has the maximum gain) to minimize the noise/offset contribution of the second sub-stage circuit 604, thereby achieving better offset/noise performance for the single-stage comparator 600.
As shown in FIGS. 6-8, the timing control circuit 604 is coupled to the input port (which includes nodes n5 and n6) of the second sub-stage circuit 604, where nodes n5 and n6 are non-gate terminals (e.g., source terminals) of an input transistor pair of the second sub-stage circuit 604. Hence, the timing control circuit 704 triggers the second-sub-stage operation of the second sub-stage circuit 604 through any of the non-gate terminals (e.g., source terminals) of the input transistor pair. However, this is for illustrative purposes only, and is not meant to be a limitation of the present invention. In some embodiments of the present invention, the input port of the second sub-stage circuit 604 may be at drain terminals of an input transistor pair or gate terminals of an input transistor pair. As a person skilled in the art can readily understand details of such alternative designs after reading above paragraphs directed to FIGS. 4-5, further description is omitted here for brevity.
It should be noted that the circuit designs shown in FIGS. 1-7 are for illustrative purposes only, and are not meant to be limitations of the present invention. That is, the circuit designs shown in FIGS. 1-7 may be modified without departing from the spirit of the present invention. For example, at least a portion (i.e., part or all) of the P-type transistors shown in FIGS. 1-7 may be replaced with N-type transistors, and at least a portion (i.e., part or all) of the N-type transistors shown in FIGS. 1-7 may be replaced with P-type transistors. For example, the first stage circuit 102/602 may be modified to use an input transistor pair consisting of P-type transistors, and the trigger signal generator circuit 202/702 (particularly, sensing circuit 203/703 of trigger signal generator circuit 202/702) may be modified to use an input transistor pair consisting of N-type transistors. For another example, the second stage circuit 104/604 may use an input transistor pair consisting of P-type transistors or an input transistor pair consisting of N-type transistors, depending upon actual design considerations. For yet another example, the second stage circuit 104/604 may employ a NOR latch structure or a NAND latch structure, depending upon actual design considerations.
Furthermore, FIGS. 1-7 only show the components pertinent to the present invention. In practice, the two-stage comparator 100/single-stage comparator 600 may include additional components to achieve other designed functions. For example, the control circuit 106/606 may include additional components to achieve more reset, more clamping, more speed up cross couple, etc.
Those skilled in the art will readily observe that numerous modifications and alterations of the device and method may be made while retaining the teachings of the invention. Accordingly, the above disclosure should be construed as limited only by the metes and bounds of the appended claims.
Patent Application
20240137039
