Apparatus and method for minimizing reception nulls in heterodyned ultrasonic signals

BACKGROUND OF THE INVENTION

[0002] 1. Field of the Invention

[0003] The invention generally relates to the field of ultrasonic generators and, more particularly, to an apparatus and method for minimizing reception nulls in heterodyned ultrasonic signals.

[0004] 2. Description of the Related Art

[0005] It is well known that ultrasonic generators and detectors can be used to locate leaks or defects, e.g., in pipes. Such a system is shown in U.S. Pat. No. 3,978,915 to Harris. In that arrangement, ultrasonic generators are positioned in a chamber through which the pipes pass. At the ends of these pipes, exterior to the chamber, ultrasonic detectors are located. At the point where a leak occurs in the pipe or the pipe wall is thin, the ultrasonic energy will enter the pipe from the chamber and travel to the end of the pipe where the detector is located. The detector will receive an ultrasonic signal at the end of the pipe indicating the existence of the leak or weak spot in the pipe.

[0006] By locating an ultrasonic generator in a closed chamber, a standing wave pattern with peaks and nodes is established. If a node occurs at the position of a leak or weak spot, no ultrasonic energy will escape and the defect will not be detected.

[0007] Ultrasonic sensors have also been used to detect ultrasonic energy generated by friction within mechanical devices as disclosed in U.S. Pat. No. Re. 33,977 to Goodman, et al., the details of which are hereby incorporated herein, in their entirety, by reference. The greater the amount of friction, the greater the intensity of the generated ultrasonic energy. Applying a lubricant to the device reduces friction and consequently the intensity of the generated ultrasound drops. Measuring ultrasonic energy thus provides a way to determine when lubrication has reached the friction generating surfaces. Additionally, faulty devices, such as bearings, generate a higher level of ultrasonic energy than do good bearings and thus, this condition can also be detected. However, conventional means require two people to perform this procedure-one person to apply the lubricant to the device, and one person to operate the ultrasonic detector.

[0008] In certain instances, e.g., when detecting the malfunction of bearings, an ultrasonic detector is mechanically coupled to the casing of the bearings so that the vibrations caused by the malfunction can be mechanically transmitted to it. With such an arrangement, the frequency is not set by an ultrasonic generator, but is created by the mechanical vibration itself. Here, an ultrasonic detector circuit must be capable of sweeping over a band of frequencies to locate the one frequency that is characteristic of the malfunction. This is usually accomplished by a heterodyning circuit which can be tuned to various frequencies, much in the manner of a radio receiver.

[0009] Since ultrasonic energy used for these purposes is generally in the range of 40 kHz, it is too high in frequency to be heard by a human being. Thus, means are typically provided for heterodyning, or frequency shifting, the detected signal into the audio range, and various schemes are available for doing this.

[0010] Ultrasonic transducers generally produce a low voltage output in response to received ultrasonic energy. Thus, it is necessary to amplify the detected signal using a high-gain preamplifier before it can be accurately processed. However, if low cost heterodyning and display circuitry are to be used, means must be made available to attenuate the amplified signal to prevent saturating these circuits when high input signals are present. This attenuation also adjusts the sensitivity of the device. For a hand-held unit, the degree of attenuation should be selectable by the user. For example, U.S. Pat. No. 4,785,659 to Rose et al. discloses an ultrasonic leak detector with a variable resistor attenuator used to adjust the output level of an LED bar graph display. However, this attenuation method does not provide a way to establish fixed reference points to allow for repeatable measurements.

[0011] U.S. Pat. No. 5,089,997 to Pecukonis discloses an ultrasonic energy detector with an attenuation network positioned after an initial pre-amplifier and before the signal processing circuitry, which creates an audible output and an LED bar graph display. The resistors in the Pecukonis attenuation network are designed to provide an exponential relationship between the different levels of attenuation. However, Pecukonis does not heterodyne the detected signals to produce an audible output, but rather teaches the benefits of a more complex set of circuits which compress a broad range of ultrasonic frequencies into a narrower audible range. For many applications, the cost and complexity of this type of circuitry are not necessary.

[0012] When using ultrasonic energy to detect leaks, it is useful to have a portable ultrasonic sensor which indicates the presence and intensity of ultrasonic energy both visually and audibly. U.S. Pat. No. Re. 33,977 to Goodman et al. discloses an ultrasonic sensor that displays the intensity of the detected signal on an output meter operable in either linear or logarithmic mode, and also provides for audio output through headphones. U.S. Pat. No. 4,987,769 to Peacock et al. discloses an ultrasonic detector that displays the amplitude of the detected ultrasonic signal on a ten-stage logarithmic LED display. However, the detector disclosed in Peacock does not process the detected signal to produce an audible response, nor does it provide for signal attenuation after the initial pre-amplification stage.

[0013] Means have been proposed for increasing the output of the ultrasonic transducer. For example, in U.S. Pat. No. 3,374,663 to Morris it is suggested that an increase in the voltage output can be achieved by serially arranging two transducers. It has been found, however, that with such an arrangement a typical transistor pre-amplifier loads the transducers to such an extent that the gains achieved by stacking them serially are lost. The Morris patent proposes the use of a triple Darlington configuration in order to produce a sufficiently high input impedance to prevent this degradation in the signal produced by the stack of transducers. Unfortunately, the transducers in this arrangement are not placed so that they both readily receive ultrasonic energy. Thus, the Morris arrangement is not entirely satisfactory.

SUMMARY OF THE INVENTION

[0014] The present invention is directed to providing improved methods and apparatus for detecting leaks and mechanical faults by ultrasonic means. In accordance with the invention, an input transducer signal is applied to a unity gain buffer amplifier that is used to maintain the impedance level seen by the transducer. The processed signal from the unity gain buffer amplifier is supplied to a voltage control amplifier that also receives a voltage control signal that is generated by a digital-to-analog converter located on an external I/O board. The voltage control signal is used to switch the voltage controlled amplifier such that the dynamic range of the signal is expanded prior to a clip of the signal. The voltage control signal is based on a level that is programmed into the voltage control amplifier by the digital-to-analog converter located on the external I/O board. The voltage controller is thus controlled by the I/O board in response to commands sent to the external I/O board from a micro-controller.

[0015] The output from the voltage controlled amplifier is connected to a fixed gain differential amplifier. The output signal from the fixed gain amplifier is supplied to a variable gain amplifier that is switchable between two fixed levels, such as 0 dB and 20 dB. The gain level of the variable gain amplifier is toggled between the two fixed gain levels based on a level that is determined by the amount of gain that is programmed into the voltage control amplifier.

[0016] The output of the variable gain amplifier is supplied to a pair of heterodyning circuits, i.e., a dual heterodyning circuit. At each respective heterodyning circuit, the output signal from the variable gain amplifier is multiplied with a local oscillator signal that is internal to each circuit. Here, each local oscillator is nominally set to 38 kHz such that for a 40 kHz input transducer signal, a difference frequency of about 2 kHz (i.e., the audio component) is provided at the output of each heterodyning circuit.

[0017] The output signal from the first heterodyning circuit is amplified and divided into two signal branches. The first signal branch is transformer coupled to a headphone output. The second signal branch is connected to an amplifier that is also transformer coupled to a line output and also applied to an external audio amplifier. The output from the second of the heterodyning circuits is amplified and supplied to a metering circuit.

[0018] In addition, a further analog signal path is created at the second heterodyning circuit. The signal in this path is converted to a linear dB format analog signal and supplied to a micro-controller. This analog signal is converted in the microcontroller into a digital signal by an analog-to-digital converter, and is further converted in the micro-controller into a WAV file format, as well as other digital signal formats, for subsequent spectral analysis.

[0019] The present inventors have determined that a heterodyned signal that drives a meter requires a relatively large dynamic range, but a limited frequency response, while a heterodyned signal that is required for headphones or spectral analysis may have a low dynamic range, but requires high resolution. Further, it has been found that the resolution or frequency response of the input transducer signal is degraded if a single heterodyning circuit is used to drive a number of circuits or meters with competing requirements. In order to overcome these competing requirements, the present invention uses a dual heterodyning circuit in which the two individual heterodyne circuits are separately optimized so that the second results in a signal with a large dynamic range and the first results in a signal with a great resolution, and neither unduly loads the transducer array or obscures subtle frequency components. This permits the capture of particularly low level frequency components for extraction during spectral analysis.

[0020] In accordance with the invention, the first heterodyning circuit has a feed back loop filter and a transformer to provide an enhanced spectral (i.e., frequency) response. This circuit is used to drive the headphone, a wave file generator and a line output. This signal, which has a modest dynamic range but a high frequency response and a low signal to noise ratio, allows the spectrum of the signal to be analyzed in real time by an external spectrum analyzer, recorded for later analysis or listened to in real time through the headphones.

[0021] The second heterodyning circuit has a smaller frequency response but a larger dynamic range so that it can drive the meter. In accordance with the invention, the second heterodyne circuit is not required to have an optimized spectral response. If the meter were driven with the first heterodyne circuit, the impedance and dynamic range requirements of the meter would adversely affect the response. Thus, two heterodyne circuits are used, with the circuit that drives the meter being simpler, and less costly to manufacture and having a larger dynamic range.

[0022] In either mechanical analysis or electrical equipment analysis, a large number of frequencies in the low frequency range become lost. This is especially true in the case of electrical applications. After extended use of the detection equipment, operators often tend to begin to use their ears as a guide to the condition of an area of concern. However, it is extremely difficult for a person to discern with their ears the differences between inputs that are electrical in nature and inputs that are vibrational. Further, in other technologies, such as vibration analysis, infrared technologies, or where rotational equipment is used, the use of the human ear is a highly unreliable way in which to predict faults. For example, a transformer resonating at 60 Hz may cause a component in an equipment cabinet to resonate at the same 60 Hz. When an operator listens to the cabinet containing the component that is vibrating at the 60 Hz, it is impossible to determine whether the resonance is electrical or mechanical.

[0023] In a further aspect of the invention, a focal point reflector (FPR) is used in conjunction with the dual heterodyne circuit to reduce reception nulls when performing the ultrasonic measurements. The focal plane reflector is a flat non-porous, reflective material, such as printed circuit board (PCB) that is placed behind and in parallel with an ultrasonic array of piezoelectric transducers which has a series connection of multiple crystal detectors. In the preferred embodiment, three crystals are used. Such a detector is enclosed in a weather resistant or environment resistant cylinder which is open in one axial direction. The multiple crystals are aligned for maximum sensitivity along this axis and a cover is located over the opening to keep out moisture. In addition a wire screen is located over the cover to protect it from physical damage. In preferred embodiments, the cover is made from Mylar.

[0024] When detecting low level leaks, it is necessary to be close to the leak and therefore in the near field of detection. In such cases, the ultrasonic energy from a low level source may not be strong enough to stimulate the piezoelectric crystals of the ultrasonic array. If the low level ultrasonic energy is not focused directly in front of the piezoelectric crystals, they will not be excited and hence, a null in the ultrasonic reception will occur. In accordance with the invention, the FPR permits the reflection back into the crystal elements of ultrasonic energy that may originally “fall” between the crystals in the array. When the transducer array is located in close proximity to a leak, the ultrasonic signal that falls between the crystals in the array is reflected by the RFP back toward the area of detection, and is then reflected back into the crystals. As a result, the reduction of reception nulls during the detection of low level leaks is achieved.

[0025] In an alternative embodiment, a multi-transducer array is used in conjunction with the dual heterodyne circuit of the invention. This multi-transducer array has a wide sensing surface that provides significantly more “coverage” of the ultrasonic detection area. As a result, the sensor provides a wide area of high sensitivity, thereby substantially eliminating all reception nulls. In accordance with the alternative embodiment, seven crystals are used.

[0026] In certain instances, it may be necessary to locate the transducers in harsh environments. For such a purpose, the transducer is made from a material that is resistant to adverse water, wind and temperature conditions, without seriously degrading its ability to detect ultrasonic vibrations. Here, the FPR may be used in a chamber with a porous screen that is placed in front of and parallel to the transducers. As a result, the transducer are protected from adverse environmental conditions.

[0027] By using the focal point reflector with the dual heterodyning circuit of the present invention to provide the enhanced spectrum, it becomes easier to determine whether a detected resonance is mechanical or electrical. In addition, fault frequencies are also more easily discernable. In other words, the enhanced signal output having fewer reception nulls provides a lower signal-to-noise ratio, so as to increase the ease with which frequency components are analyzed.

BRIEF DESCRIPTION OF THE DRAWINGS

[0028] The foregoing and other advantages and features of the invention will become more apparent from the detailed description of the exemplary embodiments of the invention given below with reference to the accompanying drawings in which:

[0029]
FIG. 1 is an exemplary block diagram of the dual heterodyning circuit in accordance with the present invention;

[0030]
FIGS. 2A through 2K is a schematic diagram of the dual heterodyning circuit shown in FIG. 1;

[0031]
FIG. 3(a) and FIG. 3(b) form a block diagram of the I/O board, the micro-controller, and the rear panel in accordance with the invention;

[0032]
FIG. 4 is a block diagram illustrating a flash card inserted into the micro-controller of FIG. 3(a);

[0033]
FIG. 5 is a bottom view of the ultrasonic instrument of the present invention;

[0034]
FIG. 6 is a perspective view showing the flash card and rear panel of the ultrasonic instrument of the invention;

[0035]
FIG. 7 is a plan view of the rear panel of the ultrasonic instrument of the invention;

[0036]
FIG. 8 is a front view of the ultrasonic instrument of the invention;

[0037] FIGS. 9(a) and 9(b) are block diagrams of an additional aspect of the invention.

[0038] FIGS. 10(a) through 10(c) are illustrations of a focal point reflector for use with the dual heterodyning circuit of FIG. 1;

[0039]
FIG. 11 is a cross-sectional view of an ultrasonic detector that includes the focal point reflector of FIGS. 10(a) and 10(b);

[0040] FIGS. 12(a) through 12(e) are illustrations of transducer arrays that include the focal point reflector of FIGS. 10(a) through 10(c); and

[0041]
FIG. 13 is a cross-sectional view a chamber containing an ultrasonic detector that includes the focal point reflector of FIGS. 10(a) and 10(b)

DETAILED DESCRIPTION OF AN EXEMPLARY EMBODIMENT

[0042]
FIG. 6 is a perspective view of a portable ultrasonic detector 600. Toward the front of the housing there are ultrasonic transducers 95, as shown in FIG. 8. Micro-processor controlled circuits for heterodyning the ultrasonic signal to shift its frequency to the audio range are contained in the body of the housing. A display 82 is located at the back so the operation and the results can be viewed. At the back, there is also a jack 88 for headphones, so that the user can listen to the audio sound during a test, e.g., as a way of locating a leak. Other jacks and controls are located on the body or will be described subsequently.

[0043]
FIG. 1 is an exemplary block diagram of the dual heterodyning circuit in accordance with the present invention which is located in the housing of the ultrasonic detector. In FIG. 1, an input signal is applied from an ultrasonic transducer to a buffer amplifier 12 (U4B) at input 11 (P0). Typically, unity gain buffer 12 is used to maintain at a desired level the impedance level seen by the transducer. The processed signal from buffer 12 is supplied to voltage controlled amplifier (VCA) 14 (U5) that also receives a voltage control signal 15 that is generated by a digital-to-analog converter on an external I/O board shown in FIG. 3(b). The voltage control is thus controlled by the I/O board in response to commands sent to the external I/O board from a micro-controller (see FIG. 3(a)).

[0044] Voltage controlled amplifier 14 is connected to a fixed gain amplifier 16. In preferred embodiments, amplifier 16 has a fixed gain of approximately 20 dB. The output signal from amplifier 16 is supplied to variable gain amplifier 18 (VGA) that is switchable between two fixed levels, such as 0 dB and 20 dB. The gain level of amplifier 18 is toggled between the two fixed gain levels based on a signal level applied to input 17. This signal is determined at the micro-controller on the I/O board based on the amount of gain that is programmed into the voltage controlled amplifier 14.

[0045] The output of VGA 18 is supplied to a first heterodyning circuit 20 (U8). In heterodyne circuit 20, the output signal supplied to VGA 18 is multiplied in multiplier 22 by a local oscillator 21 that is internal to heterodyne circuit 20. Sum and difference frequencies are provided at the output of circuit 20. At this point, the high frequency components of the signal are filtered out and a difference signal is buffered in amplifier 24, such that an audio signal is provided. The local oscillator 21 within circuit 20 is nominally set to 38 kHz such that for a 40 kHz input transducer signal, a difference frequency is approximately 2 kHz. Amplifier 24 is used to amplify the output signal and apply it to terminal 23 which leads to a metering circuit (not shown). This signal has a large dynamic range.

[0046] The output signal from VGA 18 is also received by amplifier 30, which amplifies this signal prior to supplying it to a second heterodyning circuit 32. In exemplary embodiments, the signal supplied to amplifier 30 is amplified by approximately 10 dB. The second heterodyne circuit 32 receives the output of amplifier 30 and multiplies this signal in multiplier 33 by a local oscillator 34 that is also internal to circuit 32. Sum and difference frequencies are created at the output of heterodyne circuit 43. The high frequency components of the signal are filtered out and the low frequency signal is buffered in amplifier 35, such that an audio signal is provided. The local oscillator within circuit 32 is nominally set to 38 kHz such that for a 40 kHz input transducer signal, a difference frequency audio signal is approximately 2 kHz. The audio signal is then buffered by a unity gain amplifier 36. The output of amplifier 36 is next provided to an amplifier 37. In preferred embodiments, the signal level supplied to amplifier 37 is increased by approximately 40 dB.

[0047] In accordance with the invention, the second heterodyning circuit 32 has a feed back loop 31 from the output of amplifier 35 to the input of circuit 32. This feedback loop 31 provides an enhanced spectral (i.e., frequency) response.

[0048] The output signal from unity gain amplifier 36 is divided into two signal branches. The first branch leads to the amplifier 37. The second branch leads from unity amplifier 36 to amplifier 40 that is coupled to a headphone output by way of transformer 41. In the first signal branch, amplifier 37 is coupled to transformer 36, which in turn is coupled to a line output. The line output is subsequently applied to an audio amplifier (not shown). In addition, a further analog signal from amplifier 37 is coupled to amplifier 38, where it is attenuated by approximately −3 dB. The attenuated signal is then supplied to the micro-controller (not shown). This analog signal is converted in the micro-controller into a digital signal by an analog-to-digital converter, and is further converted in the micro-controller into a WAV file format, as well as other digital signal formats for storage and playback, and for subsequent spectral analysis.

[0049] The wideband, high resolution signal from amplifier 36, which is a result of feedback loop 31, is used to drive the headphone, a wave file generator and a line output. This signal, which has a modest dynamic range but a high frequency response and a low signal to noise ratio, allows the spectrum of the signal to be analyzed in real time by an external spectrum analyzer, recorded for later analysis or listened to in real time through the headphones.

[0050] The first heterodyning circuit 20 has a smaller frequency response but a larger dynamic range so that it can drive the meter. In accordance with the invention, the first heterodyne circuit is not required to have an optimized spectral response. If the meter is driven with the same heterodyne circuit as the headphone circuit, the impedance and dynamic range requirements of the meter would adversely affect the headphone response. Thus, two heterodyne circuits 20, 32 are used, with the circuit that drives the meter being simpler, less costly to manufacture and having a greater dynamic range. The circuit that drives the headphones has a smaller and a lower signal-to-noise ratio, which provides a better spectral response.

[0051] By way of example, FIGS. 2A-2K are an exemplary schematic diagram of the dual heterodyning circuit in accordance with the present invention. Buffer amplifier 12 is shown in FIG. 2A. A transducer signal having a frequency of approximately 40 kHz +5 kHz is applied via connector 11 (P10) by way of capacitors 210 (C27) and 211 (C21), resistors 212 (R20) and 213 (R27), diodes 214 (D2) and 215 (D3) to the buffer amplifier 12, which is used to maintain the impedance level seen by an input transducer (not shown) at a predetermined fixed level. Typically, amplifier 12 is a standard Integrated Circuit (IC), such as an OP-284ES.

[0052] The voltage divider comprising resistors 220 (R36) and 221 (R45), along with capacitor 222 (C41) are coupled to the positive input of amplifier 44 (U10) that is used to generate a 6 volt low impedance output based on the 12 volt input that is applied to resistor 220. The 6 volt low impedance output is used to provide a reference level for the analog circuitry of the invention. Amplifier 44 has a feed back loop comprised of capacitor 222 (C25) and resistor 223 (R33) to improve its response. This amplifier is typically a standard “off-the-shelf” IC, such as an AD797 manufactured by Analog Devices.

[0053] Capacitor 230 (C19) and resistor 232 (R24) are connected in series from the output of amplifier 12 to the input of voltage controlled amplifier (VCA) 14 (U5). Amplifier 14 with capacitors 234 (C24), 236 (C14), 238 (C33), resistors 240 (R21), 242 (R30) provides a means for expanding the dynamic range of the signal prior to clipping of the signal. Preferably, VCA 14 is a standard voltage controlled amplifier, such as a SSM2018T manufactured by Analog Devices. The control voltage on pin 11 of VCA 14 is generated by a digital-to-analog converter 71 (DAC) that resides on an I/O board (shown in FIG. 3(b)) that is controlled by an external micro-controller (shown in FIG. 3(a)). VOS 302 is the control signal. The output of VCA 14 is on pin 3 through capacitor 236 and resistor 240 (TP5).

[0054] As shown in FIG. 2B, the output of VCA 14 on TP5 is applied to the input of the differential amplifier 16 through capacitor 242 and resistor 244. A feed back loop of capacitor 246 and resistor 248 is connected around amplifier 16. Capacitor 249 and resistors 245, 247 form the rest of the differential amplifier 16. Amplifier 16 has a predetermined fixed gain level and because of its high common mode rejection ratio noise is reduced. The output signal from amplifier 16 is coupled to the input of variable gain amplifier 18 by way of capacitor C18 and resister R23. In preferred embodiments, amplifier 16 is typically a standard “off-the-shelf” IC, such as an OP-284ES, and has a gain level of approximately 20 dB.

[0055] Amplifier 18 is switchable between two gain levels based on the sensitivity level required by VCA 14. In preferred embodiments, amplifier 18 is switched between 0 dB and 20 dB by an analog switch 45 (U3) that is typically a standard “off-the-shelf” IC, such as a DG419DY. Resistor 254 (R19), resistor 256 (R15) and variable resistor 258 (VR1) set the gain, while resistor 260 when shorted across the other resistor by switch U3 sets the second gain level The output of amplifier 18 is connected through capacitor 262 (C20) to the output at TP3. This level is biased by a voltage from variable resistor 264 (VR2).

[0056] The micro-controller sets the digital bits DAC, CLK, DACSDO, DACLD on connector J3 (FIG. 3(a)). These bits are applied to DAC 71 (FIG. 3(b)), which in turn produces the control voltage 302 (VOS or VOF) on J13. A control voltage 302 VOS is received on P13 (FIG. 2H). As shown in FIG. 2G, VOS is then supplied to amplifier 52 (U11A) by way of resistors 270 (R57) and 272 (R58) to amplifier 57 by way of resistor 274 (R83). The output of amplifier 52 is provided to one input of differential amplifier 53. A unity gain buffer amplifier 55 has an input voltage from variable resistor 276 (VR7). Its output is applied to the other input of amplifier 53 as a reference voltage. The output of amplifier 53 is amplified in amplifier 54 and provides the signal at TP12. In effect, the amplifiers 52, 53, 54 and 55 scale and level shift the VOS signal. As can be seen from FIG. 2A, the TP12 signal is applied to the control output of VCA 14.

[0057] The VOS signal level is from approximately 0 to 5 volts. In preferred embodiments, the signal level of VOS is from 0 to 4.095 volts.

[0058] In alternative embodiments, variable resistor 280 (VR8) RT1, and RG1 (FIG. 2G) are optionally connected for providing nominal temperature compensation of the system.

[0059] Amplifier 57, which also receives the VOS signal, buffers that signal and feeds the positive input (pin 3) of amplifier 56 (U14A) through resistor 284 (R82), where amplifier 56 is connected in a comparator arrangement. Here, resistor 286 (R84) is used to provide hysteresis for noise rejection. Coupled to the negative input (pin 2) of amplifier 56 is a variable reference level that is created by variable resister 288 (VR9), which sets a reference level. Typically, amplifiers 52, 53, 54 and 55 are standard “off-the-shelf” ICs, such as a LM6134AIM.

[0060] In accordance with the invention, the reference level that is applied to the negative input (pin 2) of amplifier 56 is set during a calibration process to generate a CLIP signal that is output from pin 1 of amplifier 56. This CLIP signal is used to switch the variable gain amplifier 18 from 0 dB to +20 dBs. (See the input switch to switch 45 on FIG. 2B.) Simultaneously, the gain of the transducer pre-amps (not shown) is decreased by 20 dB. Of note, in order to extend the dynamic range of the transducer amplifier (not shown), the overall gain of the system plus the transducer pre-amp must have no net increase in gain. As a result, if the variable gain amplifier in the pre-amp located within the transducer has a 100 dB dynamic range and a pad of 20 dB is inserted, then a clean, un-clipped dynamic range of 120 dB is achieved from the entire system. The signal that controls the switching of amplifier 18 is the CLIP signal that is generated by amplifier 56.

[0061] Amplifier 56 is controlled by a sensitivity setting such that the overall sensitivity of the system is determined by the micro-controller whereby an operator using a controller 72 on a front panel of the instrument 600 can adjust the overall sensitivity (see FIG. 5 and FIG. 7). As a result, if the sensitivity of the system is lowered by a predetermined level, the clip signal output from amplifier 56 is toggled such that gain switching occurs at the transducer pre-amp and at variable gain amplifier 18. In preferred embodiments, the predetermined level is 10 dB downward from the maximum sensitivity of the system.

[0062] With reference to FIG. 2A, differential amplifier 43 (U4A) receives the output 268B of variable gain amplifier 18 (FIG. 2B) on its positive input 241 (pin 3). This signal is received through resistors 251 (R13) and 253 (R17), and capacitor 255 (C10). The output of amplifier 43 is connected to zener diode 259 (D1) at TP1. Amplifier 43 functions as a positive rectifier circuit outputting a positive DC voltage proportional to the amplitude of the signal. Zener diode D1 clamps the output of amplifier 43 to a voltage of approximately 5 volts to prevent the micro-controller from being subjected to excessive voltage levels. As a result, a DC voltage is generated which the micro-controller compares to a predetermined value. If the DC voltage is greater than the predetermined value then the micro-controller indicates saturation on the LCD display by displaying an over range condition.

[0063] The output of amplifier 18 is also applied to the first of a pair of function generator circuits that form the dual heterodyne circuits 20 (U8), 32 (U99), FIG. 2F. The output of amplifier 18 is further connected to resistor 130 (R8) that is connected in series with capacitor 135 (C5), which is subsequently connected to the base of transistor 134 (Q1) (FIG. 2C). The collector of transistor 134 is capacitively connected to the input (pin 1) of the second of the pair of function generator circuits, i.e., heterodyne circuit 32 (U99) (see FIG. 2F) by way of capacitor 136 (C3). As shown in FIG. 2C, a feed back loop comprising capacitors 140 (C12), 142 (C11), transistor 47 (Q2) and variable resistor 144 (VR14) provides a feedback signal at pin 1 of function generator (heterodyne) circuit 32 (see FIG. 2F). In accordance with the invention, transistor 46 amplifies the output signal from amplifier 18 by a predetermined amount. In the preferred embodiment, the predetermined amount is 10 dB.

[0064] Ultrasonic signals leaking from a container (not shown) are detected by the transducer (not shown), amplified and frequency shifted such that a user is provided with an indication of the existence of a leak by way of the sound heard in a pair of headphones (see FIG. 2K). The actual frequency shift of the ultrasonic signal is accomplished in the function generator 32. The generator (FIG. 2F) may be a commercially-available integrated circuit, such as the EXAR 2206, which has been wired to produce sine wave outputs at a frequency determined by tuning resistor 180 (VR3) connected to pin 7 of circuit 32, resistors 181 (R46) and 182 (R49) connected from pins 15 and 16 to ground, capacitors 183(C38), 184 (C43), and resistor 186 (R52). One characteristic of circuit 32 is that a particular bias applied to its input (pin 1) will cause it to produce an amplitude-modulated (AM), suppressed-carrier output. The bias to obtain this suppressed-carrier modulation is derived from variable resistor 144 (VR14) (FIG. 2C). If capacitor 183 (C37) and resistor 180 (VR3) are selected to produce a carrier signal that differs from the ultrasonic signal by a frequency in the audio band, the output of heterodyne circuit 32 will be an audio signal related to the input ultrasonic signal and a much higher signal. In particular, the output signal will be equivalent to the sum and difference frequencies of the ultrasonic signal and the carrier signal generated within circuit 32, but the carrier signal itself will not be present in the output. If, for example, variable resistor 180 (VR3) is set such that circuit 32 generates a 42 kHz signal and the ultrasonic signal applied through capacitors C3 to circuit 32 is at 40 kHz, the output will be at 2 kHz and at 82 kHz. In preferred embodiments, the oscillator in circuit 32 is adjusted between a range of 20 kHz and 100 kHz.

[0065] Although a proper bias on the input to circuit 32 will eliminate or suppress the carrier generated by that circuit, it has been found that this adjustment is critical and some carrier may leak through due to temperature and voltage variations. Also, as the carrier frequency is changed due to changes in the setting of resistor 180 (VR3), there are changes in the circuit operation that may cause the carrier to appear in the output unless there is an adjustment of the bias. In order to provide this adjustment, a servo or feedback network is provided.

[0066] In particular, the output of circuit 32 is also capacitively coupled to the base of transistor 35 (Q3) by way of capacitor 190 (C36), and resistor 192 (R40), as shown in FIG. 2F. Together these components provide an input signal for the feedback network formed by transistor 47 that biases pin 1 of circuit 32 (see FIG. 2C). Here, transistor 35 provides amplification of the output signal from pin 2 of circuit 32 by a predetermined amount. In preferred embodiments, the predetermined amount of amplification is 40 dB.

[0067] The output from pin 2 of circuit 32 is also fed to voltage amplifier 36 (FIG. 2E), where the signal from pin 2 is buffered. By way of resistor 171 (R140), the output signal from amplifier 36 feeds the base of transistor 37 (Q6) over line 304, by way of capacitor 305 (C66) (FIG. 21). Here, the output signal from amplifier 36 is coupled to transformer 39 to thereby generate a low frequency output (“LFO”). The audio signal on line 304 is also applied to summing amplifier 68 (FIG. 2K) which in turn drives amplifier 40. Amplifier 40 drives transformer 41 which is used to power the headphones. In preferred embodiments of the invention, transformer 39 has a turns ratio of approximately 1:0.05, and the output signal is used to drive low impedance loads. The transformer 41 has a turns ratio of 1:0.175.

[0068] The output signal from amplifier 36 (FIG. 2E) is also provided to amplifier 50, where it is attenuated by approximately −3 dB, based on resistors 300 (R34) and 302 (R26). Amplifier 50 (U1B) and amplifier 36 are typically standard “off-the-shelf” ICs, such as an OP-284ES. The output from amplifier 50 is supplied to the micro-controller for conversion into a digital signal by an analog-to-digital converter located in the micro-controller (not shown). This digital signal is converted into a digital format, such as a WAV file, for subsequent image processing.

[0069] Signals VR and +12VR are applied from a power supply (FIG. 2K) to the circuit of FIG. 21. These signals are applied to the negative and positive terminals of differential amplifier 59 (U12B). Capacitor 306 (C48) and resistor 308 (R70) are connected to form a feedback loop about amplifier 59. The signal +12VR is applied to the positive input of amplifier 59 through resistors 310, 311 and 312 (R81). A zener diode 314 (D5) is connected between resistors 310 and 311. VR is connected to the negative input of amplifier 59 through resisters 315 (R68) and 316 (R69). The output of amplifier 59 is connected to zener diode 318 (D4), and through resister 319 (R76) to the positive input of amplifier 60 (U12A). A variable resistor 320 (VR6) is connected to amplifier 60 and serves to establish a reference point of amplifier 60.

[0070] Signal +12V1 is applied from the power supply (FIG. 2K) to the bias amplifier shown in FIG. 2D. This 12V signal is applied to the VIN terminal of voltage regulator 48 (U2). The output (VOUT) of voltage regulator 48 provides a +5 volt TTL signal that is supplied to amplifier 49 (U1A) by way of resistors 360 (R6), 362 (R9), and capacitors 364 (C7) and 365 (C8). Amplifier 49 provides a regulated +2.5V voltage that is used as a reference voltage in accordance with the invention. The +5V voltage is also used to provide a TTL reference level to all other circuit ICs where required.

[0071] With further reference to FIG. 21, amplifiers 59 and 60 provide a comparator circuit that generates a low battery monitor. By way of zener diode D5, a regulated reference voltage is generated and applied to the positive input (pin 5) of amplifier 59. Concurrently with application of the regulated reference voltage, a battery voltage is applied to the resistive divider (315, 311) on the negative side (pin 6) of amplifier 59. The reference voltage at zener diode D5 remains relatively constant due to the clamping action of the zener diode D5.

[0072] Zener diode D4 in FIG. 21 is connected to the output of amplifier 59, and clamps the output voltage to approximately 5 volts such that the micro-controller is not subjected to excessive voltage levels. If the battery voltage falls below a predetermined level, then the input voltage at the negative input of amplifier 59 will also fall below the reference level. In accordance with the invention, the output of amplifier 59 is zero to indicate a fully charged battery, and approximately 3.5 volts to 4 volts (nominal) to indicate that the battery capacity is low and needs to be recharged. The output of amplifier 59 is inverted in amplifier 60 and produces the OFF signal used in the circuit of FIG. 2H, as will be explained subsequently. As a result, a means is provided for the micro-controller to indicate on an LCD whether or not the battery is adequately charged. In preferred embodiments, amplifiers U12A and U12B are standard “off-the-shelf” ICs, such as an LM6132.

[0073] In the contemplated embodiments of the invention, the LCD is a screen that is large so that the display can easily be seen by the operator. In accordance with the contemplated embodiments, this would include a time series display of the heterodyned ultrasonic signal to permit the viewing of measurement trends in real time.

[0074] Returning to FIG. 2E, when the battery level falls below the optimum operating level, the base of transistor 73 (Q4) is pulled high by the output of amplifier 60 (FIG. 21) on line 322. This causes the plus input of amplifier 36 to be low. As a result, the output signal from amplifier 50 is also low.

[0075] As stated previously in connection with FIG. 1, the first output from amplifier 18 is applied to the first of the pair of function generator circuits, e.g., circuit 20 (see FIG. 2F). This generator may also be a commercially available integrated circuit, such as the EXAR 2206, which has also been wired to produce sine wave outputs at a frequency determined by tuning resistor 330 (VR5) connected to pin 7 of the circuit 20, resistor 331 (R51), capacitor 332 (C42), as well as capacitor 333 (C37) connected between pins 5 and 6, and resistors 334 (R43) and 335 (R48) connected to ground from pins 15, 16 of circuit 20.

[0076] Function generator circuit 20 multiples the first output signal using an oscillator that is internal to circuit 20. In a manner similar to circuit 32, the sum and difference frequencies of the ultrasonic signal are also generated at the output pin 2 of circuit 20. In preferred embodiments, the local oscillators in circuit 20 and circuit 32 are nominally set to 38 kHz. As with the tuning resistor 180 (VR3) that is connected to circuit 32, if tuning resistor 330 (VR5) is set such that circuit 20 generates a 42 kHz signal and the ultrasonic signal applied is at 40 kHz, the output at pin 2 of circuit 20 will be at 2 kHz and at 82 kHz. Since only the audio band signal is desired, the filter circuit comprising resistors R38, R39, R42 and R44, capacitors C35, C40 and C39 will eliminate the 82 kHz sum signal. In preferred embodiments the oscillator in circuit 20 is adjusted between a range of 20 kHz and 100 kHz.

[0077] Frequency control of function generator circuits 20 and 32 is achieved by the micro-controller 80 (see FIGS. 3(a) and 3(b)). As shown in FIG. 2F, an input signal 302 VOF is applied to the positive input (pin 5) of amplifier 51 (U7B). VOF 302 originates from the DAC 71 which is on the I/O board (FIG. 3(b)). The voltage level of VOF is from approximately 0 to 4.095 volts. The oscillation frequency of circuit 20 and circuit 32 is set during a calibration process by way of variable resistors 330 (VR5) and 180 (VR3) (see FIG. 2F). In accordance with the invention, when the frequency of the system is tuned, voltage VOF is changed, i.e., the voltage applied to pin 5 of amplifier 51 is changed (FIG. 2F). As a result, the frequency of the local oscillators of circuit 20 and circuit 32 can be changed in the range from approximately 20 kHz to 100 kHz.

[0078] In accordance with the invention, the output from heterodyne circuit 20 (FIG. 2F) is provided to amplifier 24 on line 340, as shown in FIG. 2H. Connected to amplifier 24 are resistors 345 (R73), 344 (R65), and capacitors 342 (C47), and 346 (C53). The output signal meter (pin 1) of amplifier 24 is provided to an additional circuit for conversion into RMS units and dB units (see FIG. 2J). The collector of transistor 74 (Q5) (FIG. 2H) is connected to the positive input of amplifier 24, while the base of transistor 74 is connected through resistor 348 (R80) to OFF signal output from amplifier 60FIG. 21). As a result, when the battery level falls below the optimum operating level, the base of transistor 74 is pulled high and the output signal from amplifier 24 is terminated. Typically, amplifier 24 is a standard “off-the-shelf” IC, such has an OP-284ES.

[0079] The output signal meter (pin 1) of amplifier 24 shown in FIG. 2H is provided to the input of amplifier 61 (U9B) by way of connector J11 (FIG. 2J). Connected to the positive input (pin 5) of amplifier 61 are resistors 400 (R106) and 405 (R107). A low pass filtered output signal from amplifier 61 is provided to the positive input (pin 3) of amplifier 62 (U9A) through capacitor 411 (C74) where it is buffered and output from pin 1 of amplifier 62 over resistor 420 (R105) and capacitor 423 (C72) to pin 15 of RMS-to-DC converter 65 (U19). Typically, amplifiers 61 and 62 are standard ICs, such as an OP-284-ES. RMS-to-DC converter 65 is typically a standard “off-the-shelf” IC, such as an AD637 manufactured by Analog Devices.

[0080] With further reference to FIG. 2J, RMS-to-DC converter 65 computes the root-mean-square, or the mean square of the absolute value of the input signal at pin 15 of converter 65 and provides an equivalent de output voltage at pin 16, as well as an RMS output at pin 11. The DC output voltage at pin 16 of converter 65 varies linearly to the dB level of the input signal's amplitude at pin 15 of converter 65. Here, the de output voltage is a buffered output that is provided to amplifier 67 (U17A) by way of resistor 426 (R110) and resistor temperature compensator 429 (RT1).

[0081] Resistors 432 (R111), 435 (R108) and variable resistor 438 (VR10) are coupled to amplifier 67. Together, these resistors control the gain of amplifier 67 to thereby scale the dB level of the output signal that is seen on connector J11. Here, R108 is not installed so VR10 completely controls the scaling of the dB output signal from amplifier 67. This output signal is forwarded by way of pin 1 (TP21) on connector J11 to the I/O board shown in FIG. 3(b) and the micro-controller shown in FIG. 3(a).

[0082] As further shown in FIG. 2J, voltage regulator 64 (U20) is connected to BUFIN (pin 1) of the RMS-to-DC converter 65. The voltage regulator 64 receives +12V2 that is supplied on connector J11 from the power supply (FIG. 2K) and converts this 12 volt input voltage to a regulated output voltage that is output on pin 2 of regulator 64. Resistors 441 (R113), 444 (R113), and 447 (R112) set the level of a regulated output voltage from regulator 64, where variable resistor 450 (VR11) provides a means to adjust the output current and set the 0 dB reference level for converter 65 of this regulator. Typically, the voltage regulator 64 is a standard “off-the-shelf” IC, such as a LM317 manufactured by National Semiconductor Corporation

[0083] Coupled to output offset (pin 4) and analog common (pin 3) of the RMS-to-DC converter 65 is a voltage regulator 66 (U21) that also receives the +12V2 voltage from the power supply. The voltage regulator 66 provides a +5 volt output that is also supplied to the positive input (pin 3) of amplifier 67. Voltage regulator 66 is typically a standard “off-the-shelf” IC, such as a LM78L05CM.

[0084] RMS output (pin 11) of the RMS-to-DC converter 65 is provided to the positive input (pin 5) of amplifier 63 through resistor 453 (R102). Averaging capacitor 464 (C75) is connected across pins 11 and 10 of converter 65 and is used to determine the averaging error that occurs during the calculation of the true RMS of the input signal supplied to pin 15 of the converter 65. The magnitude of the error is dependent on the value of capacitor 464. As shown in FIG. 2J, the RMS output from pin 7 of amplifier 63 is forwarded by way of pin 2 of connector J11 to the I/O board shown in FIG. 3(b) and the micro-controller shown in FIG. 3(a). Typically amplifiers 63 and 67 are standard “off-the-shelf” ICs, such as a LM6132AIM.

[0085] The dB output signal at J11 pin 1 (FIG. 2J) has a 50 dB dynamic range, a 0-5V DC scale for direct input to the micro-controller, and an accurate linear dB format. These provide an elimination of the need for expensive DSP processors or math co-processors, a limitation or reduction of the memory requirements for data and code, and because of the accurate analog preprocessing, an elimination of the need for elaborate signal analysis or data conversion algorithms. In addition, a reduction of signal processing time is also provided, as well as reduced processor clock speeds which in turn lower power consumption. It should be noted that this invention performs real time analog signal processing on the heterodyned signal only.

[0086] Turning now to FIG. 2K, therein shown is an audio amplifier that is used to provide an audio output signal that is supplied to a pair of headphones connected to the jack 88 (J12) on the rear panel of the housing (FIG. 7).

[0087] As stated previously, the audio signal on line 304 is applied to one input of the inputs of the summing amplifier 68. An input alarm signal is supplied to the second input of the summing amplifier through capacitor 509 (C62), resistors 500 (R90), 503 (R91), and variable resistor 506 (VR15). Voltage follower amplifier 69 (U15A) utilizes the +12V voltage from the power supply to create a 6 volt reference level (pin 1) that is supplied to the positive input (pin 5) of summing amplifier 68.

[0088] The output signal from the summing amplifier 68 is applied to audio amplifier 40 (U16) which is transformer coupled by transformer 41 (T1) to the jack 88 on the rear panel of the housing (FIG. 7). Control of the audio volume is achieved by a signal VOL that is provided on connector J7 through resistors 518 (R96), 521 (R97) to pin 4 of amplifier 40. In preferred embodiments, amplifier 40 is a standard “off-the-shelf” IC, such as a TDA7052A manufactured by Philips Semiconductors.

[0089] FIGS. 9(a) and 9(b) are block diagrams of an additional aspect of the invention. In FIG. 9(b), a digital camera 90 is used to make a picture of the device being ultrasonically measured. The camera 90 is typically mounted on the detector housing (FIG. 9(a)). The picture signal and the signal from the dual heterodyne circuit may be combined in a circuit 75, but the camera may be activated independently of the system. The combiner 75 may be connected to a printer 76 and transmits print information directly to the printer from a user in a manner that is known. In preferred embodiments, the camera is a digital camera that stores image files. Thus, pictures of the device under test may be printed, as well as text results.

[0090] In certain embodiments, the camera utilizes a laser beam to pinpoint the location of the image. The recorded image is then “coupled” or “linked” to the stored information for that location, e.g., ultrasonic data, WAV file, and atmospheric conditions. The recorded image and the stored information for the image location is then uploaded to a suitable portable storage device in the instrument, such as a flash card 83 (see FIG. 4 and FIG. 6), smart media or memory stick. The recorded image and the stored information is then downloaded to a data base computer and incorporated into a data base program that generates a report for determining the condition of the device being measured.

[0091] With specific reference to FIG. 9(b), when an ultrasonic measurement of a device is performed, a picture can be captured and stored in memory using the camera 77. The picture can then be forwarded to micro-controller 80 where it is combined with the WAV and line output from the second heterodyne circuit 32 (see FIG. 1) in combiner 75 for output to the printer 76. In preferred embodiments, the printout comprises a spectral display of the line output and a graphical display of the WAV file information from the second heterodyne circuit 32 (see FIG. 1), as well as a picture of the device under test.

[0092] With reference to FIG. 3(a), sensitivity encoder 100 is used to increase or decrease the sensitivity level of the dual heterodyne circuit in accordance with the present invention. As shown in FIGS. 6 and 7, the sensitivity is adjusted by turning a rotational knob 72 that is located at the back of the housing. In preferred embodiments, the sensitivity encoder 100 is a rotational optical encoder.

[0093] Rotation of the sensitivity encoder 100 by way of knob 72 changes a signal on P24 (FIG. 3(b) which causes D/A converter 71 to the change the output level of the control voltage VOS 302 on connecter J13 (FIG. 3(b)) that controls the gain of VCA 14 (see FIG. 2A). Consequently, changes in the gain of VCA 14 produce proportional changes in the sensitivity level of the dual heterodyne circuit.

[0094] With reference to FIG. 7, LCD 82 provides a display of data that is used to distinguish between trends or deviations in readings. As a result, a user is provided with the means to bypass valve analysis and pinpoint an ultrasonic source, such as an internal leak in a tank or vessel, or an underground leak in gas piping or electrical transmission lines.

[0095] Sensitivity level indicator 105, shown on the LCD 82, provides the user with the ability to view the sensitivity level setting of the dual heterodyne circuit. As a result, the user can consistently set the sensitivity level of the circuit to permit repeated comparative frequency spectrum measurements, where repeatability is critical. As shown in FIG. 7, LCD 82 displays the sensitivity level setting as a range of integer numbers. In the preferred embodiment, this range is from 0 to 70, where S is an abbreviation for sensitivity.

[0096] In accordance with the invention, the integer numbers represent the adjustment range of VCA 14 (FIG. 2A), where each integer value corresponds to one decibel in the change of the gain of VCA 14. In accordance with the preferred embodiment, a sensitivity level setting of 70 corresponds to maximum sensitivity while a sensitivity level setting of 0 corresponds to a minimum sensitivity setting (70 dB below maximum sensitivity). In accordance with the invention, the sensitivity setting is also a field in the memory of the portable ultrasonic detector so that when the user presses the store button 85, the sensitivity level setting value is stored. In certain embodiments of the invention, the user can also annotate data files that are stored, and by way of voice recognition incorporate them into a final report.

[0097] In accordance with the invention, “Spin and Click™” controls are used to provide an end user interface that is simple and intuitive. With reference to FIG. 7, knob 72 acts as a cursor control. As knob 72 is clicked, the cursor moves in a set pattern around the display screen 82. If a “function field” is blinking, knob 72 is then spun to change the values within the function field. Once a function is selected, knob 72 is then clicked to set the selected value.

[0098] In accordance with the preferred embodiment of the invention, multiple applications can be displayed. In the preferred embodiment, 6 applications can be displayed, i.e., GENERIC, LEAKS, STEAM TRAPS, VALVES, BEARINGS AND ELECTRICAL. Each application has two screens, i.e., MAIN and STORAGE. In addition, the screens VALVES AND BEARINGS have an ABCD SCREEN. The “Click” on knob 72 moves the “cursor” to “FIXED” positions on each screen. In certain embodiments, the number of controls are minimized. In the preferred embodiment, two controls are used to permit the user to “navigate” through the various display screens, and change multiple operational settings.

[0099] In another embodiment of the invention, a focal point reflector 1000 (FPR) shown in FIGS. 10(a) and 10(b) is used in conjunction with the dual heterodyne circuit to reduce reception nulls that can occur when performing the ultrasonic measurements. The focal plane reflector 1000 is a flat non-porous, ultrasonically reflective material, such as printed circuit board (PCB) that is placed behind and in parallel with an ultrasonic array of piezoelectric transducers. The array is a series connection of multiple crystal detectors (FIG. 11 and FIGS. 12(a) thru 12(e)).

[0100] In a preferred embodiment, three crystals detectors are used. Here, the three crystal detectors, only one of which is visible in FIG. 11, are set on pedestals 1010 and are electrically connected to a printed circuit board 1020 which is within a container for the sensor electronics. The FPR 1000 includes through holes 1001 that are evenly spaced for permitting each pedestal of the crystal detectors to be inserted pass through the FPR to permit the electrical connection of the crystal detectors to the printed circuit board 1020.

[0101] As shown in FIG. 12(c), such a detector is enclosed in a weather resistant or environmentally resistant cylinder 1030 which is open in one axial direction. In this arrangement, the three crystal elements are positioned in the cylindrical housing 1030 and are aligned with the axis X of the housing 1030 for maximum sensitivity. In FIG. 11, only one of the crystal detectors on the far side of the X axis can be seen. The other two are spaced about the X axis at 120° angles from the one seen in FIG. 11. A cover (not shown) is positioned over the opening to keep out moisture. In addition a wire screen (not shown) is located over the cover to protect it from physical damage.

[0102] The connection between the dual heterodyning circuit and the multiple crystal array 1040 comprises three wires. These wires are used to supply the clip signal (pin 2) supplied to pin 6 of analog switch 45 (FIG. 2B), the ultrasonic signal (pin 3), and +12V (pin 1) via connector 11 (P10) (see FIG. 2A).

[0103] When detecting low level leaks, it is necessary to be close to the leak and therefore in the near field of detection. In such cases, the ultrasonic energy from a low level source may not be strong enough to stimulate the piezoelectric crystals of the Trisonic™ array 1050 shown in FIG. 12(a). If the low level ultrasonic energy is not focused directly in front of the piezoelectric crystals 1040, they will not be excited and hence, a null in the ultrasonic reception will occur. In accordance with the present embodiment, the addition of the FPR 1000 to the transducer array, as shown in FIG. 11, permits the reflection of ultrasonic energy that may “fall” between the crystals in the array and then back into the crystal elements. (See the area shown in FIG. 12(d)). When the transducer array is located in close proximity to a leak, the ultrasonic signal that falls between the crystals in the array is reflected by the RFP back toward the area of detection, and is then reflected back into the crystals. As a result, the reduction of reception nulls during the detection of low level leaks is achieved.

[0104] In an alternative embodiment, a multi-transducer array 1060 (shown in FIG. 12(e)) is used in conjunction with the dual heterodyne circuit of the invention. This multi-transducer array 1060 has a wide sensing surface that provides significantly more “coverage” by the crystal elements of the array. As a result, the sensor is provided with a wide area of high sensitivity to thereby substantially eliminate all reception nulls. In accordance with the alternative embodiment, seven crystal elements are used. Here the crystal detectors are spaced about the X axis at 60° angles, with a single crystal detector being arranged in the center of the FPR as seen in FIG. 12(e).

[0105] In certain instances, it may be necessary to locate the transducers in harsh environments. For such a purpose, the transducer is made from a material that is resistant to adverse water, wind and temperature conditions, without seriously degrading its ability to detect ultrasonic vibrations. Here, the FPR 1000 may be used in a chamber 1065 with a porous screen 1060 that is placed in front of and parallel to the crystal detectors. As shown in FIG. 13, the cylindrical metal housing 1030 is retained in one end wall 1070 of a closed weather-tight metal container which holds the remaining elements of the sensor. An “O”-ring 1080 provides a secure seal between the housing 1030 and the end wall 1070. The other end of the housing is open. Further, “O”-ring 1075 seals housing 1030 against a flange 1035 surrounding the FRP 1000.

[0106] Ultrasonic vibrations are generally received along the axis X through the opening 1090 in the cylindrical housing. These vibrations, however, must pass through the metal screen 1060 and a sheet of flexible, tough, water resistant material 1100, such as a polyester film. One suitable materials is Mylar, which is electrically conductive. The Mylar completes the covering of the open end of the cylindrical housing 1030 so it becomes moisture proof, but allows the ultrasonic vibrations to reach crystal detectors 1040 without substantial attenuation. The metal screen 1060 and the conductive Mylar barrier 1100, both of which are connected to the metal container at wall 1070, attenuate RF interference, while still allowing free passage of the ultrasonic frequencies of interests. Further, screen 1060 provides protection to the Mylar against physical impact.

[0107] The use of the focal point reflector with the dual heterodyning circuit minimizes reception nulls to thereby provide an enhanced output spectrum. As a result, it is easier to determine whether the resonance is mechanical or electrical. In addition, fault frequencies are also more easily detected. The enhanced signal output provides a lower signal-to-noise ratio, so as to increase the ease with which frequency components are analyzed.

[0108] Although the invention has been described and illustrated in detail, it is to be clearly understood that the same is by way of illustration and example, and is not to be taken by way of limitation. The spirit and scope of the present invention are to be limited only by the terms of the appended claims.

	Number	Date	Country
Parent	10292799	Nov 2002	US
Child	10386008	Mar 2003	US

Apparatus and method for minimizing reception nulls in heterodyned ultrasonic signals

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