In the last decade, environmental, industrial, and military monitoring applications have motivated research related to wireless sensing. Some applications, such as structural health monitoring, would benefit from sensing that is deeply embedded in the environment for an extended period of time. For such applications, it is desirable that the sensing device has an indefinite operating lifetime and a compact geometry for deployment ease and to ensure that the integrity of the structure being monitored is not compromised. The former constraint necessitates either that the sensing device has sufficient on-board energy resources to sustain an active device for the deployment lifetime or that the node be passive. For use in civil infrastructure, the approach of using an on-board energy resource may not be tenable. Because of this, there has been a large amount of recent work on passive wireless sensing systems.
In addition to on-board direct current (DC) power considerations, a passive device should require very low externally provided power for activation to ensure sufficient wireless transmission range. One approach for providing such power is through radio frequency (RF) interrogation. Both passive radio frequency identification (RFID) and surface acoustic wave (SAW) sensors operate on this premise. Unfortunately, such devices are typically constrained to short-range implementations, typically a few feet, because of their requisite activation power.
In view of the above discussion, it can be appreciated that it would be desirable to have a compact passive device for wireless sensing that requires relatively low externally provided power for activation and that provides sufficient wireless transmission range.
The present disclosure may be better understood with reference to the following figures. Matching reference numerals designate corresponding parts throughout the figures, which are not necessarily drawn to scale.
As described above, it would be desirable to have a compact passive device for wireless sensing that requires relatively low externally provided power for activation and that provides sufficient wireless transmission range. Disclosed herein are repeaters for wireless sensing applications. In some embodiments, the repeaters are harmonic repeaters that comprise a frequency multiplier and conjugate-matched three-dimensional receive and transmit antennas. In some embodiments, the repeaters are configured to receive an interrogation signal and re-radiate a return signal that has a frequency that is twice the frequency of the interrogation signal. The return signal can be modulated by a passive sensor to which the repeater is electrically coupled. In some embodiments, the repeater has a communication range of greater than approximately 50 m using a 2 W source in a free-space environment.
In the following disclosure, various specific embodiments are described. It is to be understood that those embodiments are example implementations of the disclosed inventions and that alternative embodiments are possible. All such embodiments are intended to fall within the scope of this disclosure.
As identified above, the disclosed repeaters comprise frequency multipliers. In some embodiments, the frequency multipliers are diode-based multipliers that re-radiate return signals having a frequency that is twice that of the incoming interrogation signal. In such a case, the repeater can be classified as a frequency doubling reflectenna (FDR).
Schottky barrier diodes are desirable in passive frequency multipliers because of their strong non-linear current-voltage characteristics. A GaAs Schottky diode can be used in developing an embodiment of an FDR. Such a diode has low signal loss at frequencies of interest and turns on at a very low induced voltage because of its low barrier junction.
The main parameter of interest in characterizing a multiplier design is the conversion gain (CG). The CG of a diode doubler is expressed as follows:
CG=Pout(dBm)−Pin(dBm) (Equation 1)
where in this case Pout is the output power of the multiplier at the second harmonic (e.g., 4.8 GHz) and Pin is the input power at the fundamental frequency (e.g., 2.4 GHz).
A simulation was performed to predict a multiplier response using Agilent's Advanced Design System (ADS) software.
A DC bias connection was included in the frequency doubler design. The bias was supplied to the diode input through an 18 nH series inductor (Coilcraft 0402) and 8.2 pF shunt capacitor (Johanson 0201). The purpose of including the DC bias connection was to impart amplitude modulation on the retransmitted signal. In addition, this connection provided a DC return path on the input side of the diode. For the output side, the DC path was provided by a shunt stub that was added to the transmit antenna feeding network.
The receive antenna 12 is a three-dimensional half-wave dipole antenna. In some embodiments, the antenna operates at 2.4 GHz. As shown in
With reference to
The second antenna element 34 of the receive antenna 12 is anti-symmetrical to the first antenna element 26. As is shown in
As is further shown in
Based on the diode doubler ADS simulation, the receive antenna 12 has an impedance of 69-F317i Ω at 2.4 GHz for maximum CG at −30 dBm input power. To match the antenna input impedance to this desired impedance, Ansoft HFSS 11 software was used to optimize parameters, such as the dimensions of the meandered portions 32, 40 of the antenna elements 26, 34, the width and length of the non-meandered sections of the antenna elements, the width of the parallel plate balun, the length of the 50Ω microstrip line 46, and the width and length of the microstrip matching line 44. Table I shows example 2.4 GHz receive antenna dimensions for variables illustrated in
An approximate equivalent circuit model of the receive antenna 12 is illustrated in
With reference back to
As shown in
As is further shown in
For optimal frequency conversion gain at −30 dBm input power, the ADS doubler simulation shows that the transmit antenna 14 should have an impedance of 15-F152i Ω at 4.8 GHz. This result was obtained from simulations of the transmit antenna 14 in the presence of the receive antenna 12. In order to match the 4.8 GHz antenna input impedance to the desired impedance, the width and length of the dipole arms 54, 62, parallel plate balun width, and the meandered matching line width and length were all optimized. In addition, the length and width of the shunt stub 72 were subsequently tuned. The shunt stub 72 was added to increase the reactive part of the input impedance and to give a DC return path to the diode doubler. The final approximate equivalent circuit model of the receive antenna is shown in
The simulated transmit antenna input impedance and the reflection coefficient between the transmit antenna input and the doubler output at −30 dBm are shown in
With reference back to
Two prototype repeaters, designated FDR1 and FDR2, were fabricated for testing purposes. The repeaters were formed using a printed circuit board milling machine and were assembled manually. Silver epoxy was used to attach the diode and lumped components, and copper wire was used to create the via connections to the ground plane.
Measurements were performed inside an anechoic chamber where the FDRs received the transmitted f1 signal, doubled the frequency, and re-radiated a 2f1 signal. A vector network analyzer (VNA) was used to send the transmitted signal and a spectrum analyzer (SA) was used for measuring the received signal power level. Path spreading loss was calculated using the Friis transmission equation. The measurements were executed over a 1 m distance and the FDR was oriented in the direction where the maximum CG was recorded. The two interrogator antennas (transmit and receive) were placed 1.3 m apart.
The CG versus input power at 2.4 GHz was measured for the two FDRs and was compared with the simulated data in
The measured and simulated FDR parameters at f1 of 2.4 GHz are listed in Table III. Good agreement was observed between the measured and predicted data.
Outdoor measurements were performed to validate the FDR performance in a long-range, free-space environment. The return signal was measured as the FDR was moved to vary the distance from the interrogator, as shown in
In order to facilitate the use of the FDR in a sensing application, the ability to modulate the return signal has been included. Using the DC network that connects to the diode input, a bias voltage can be applied to change the impedance match with the receive antenna causing a change in the CG and allowing for amplitude modulation of the re-transmitted signal from the transceiver.
As mentioned above, the three-dimensional approach provides the capability of housing the sensor electronics inside the structure. This requires that the FDR performance is insensitive to the insertion of objects (e.g. sensors) inside the cube. To demonstrate this, the FDR was tested with metallic and lossy dielectric (∈r of 10 and loss tangent of 0.1) blocks measuring 2×5×2 mm3 placed inside on the bottom side of the cube. Experimental results show that the transceiver performance was not degraded with these blocks, which occupied a larger volume than many practical sensors.
The ability to calibrate the propagation channel between the interrogator and a remote sensor node, effectively determining the round-trip path loss absent sensor stimulus effects at the node, can be important for practical implementation of this passive sensing technology. Such calibration may be needed when the exact distance to the FDR or the characteristics of the embedding environment are unknown. In one embodiment, remote calibration can include the use of two, orthogonally polarized FDRs that are in close proximity and operate as a single unit. One of the FDRs has no local stimulus and is referred to as the reference node, while the second FDR operates as the sensing node and has the local stimulus (DC bias in this demonstration) connected to the diode multiplier. To minimize the coupling between the FDRs and distortion in their radiation patterns, one of the FDRs is placed in the radiation pattern null of the other. Through simulation and experimental testing it was found that a separation distance of 8 mm between the nodes resulted in minimal change in the performance of either FDR relative to the single, isolated node performance.
The measurement setup used to perform the remote calibration is shown in
The above-described remote calibration approach is of practical value for different sensing scenarios. When using sensors that provide an absolute measure of a potentially slowly varying stimulus, such as temperature, the calibration process can involve the use of a reference signal that sets a constant baseline level against which the signal from the sensor node is compared. For sensors providing a relative measure of a stimulus that may change relatively fast over time, such as vibration, the constant calibration signal can be used to locate the node and ensure it is physically intact.
As described above, one goal is to minimize the size of the repeater. This size can be reduced by removing the portions of the substrates that are not needed to support the antennas or other electrical features of the repeater. In some embodiments, the substrates can be machined to remove these portions. With such an approach, it is possible to fit the repeater within a sphere having a diameter of approximately 21 mm, which is equal to λ1/6, which constitutes a 32% size reduction relative to the embodiment shown in
The receive antenna 102 comprises two separate antenna elements that together form the dipole, including a first antenna element 116 and a second antenna element 118. The first antenna element 116 includes a vertical portion 120 that extends upward from the base substrate 108 along the inner surface of the front substrate 110 and an arm 122 that extends laterally from a top end of the vertical portion. The arm 122 extends horizontally along the inner surfaces of both the front substrate 110 and the first lateral substrate 112. Provided at a distal end of the arm 122 on the inner surface of the first lateral substrate 112 is a meandered portion 124 that extends downward from the distal end of the arm.
The second antenna element 118 of the receive antenna 12 is anti-symmetrical to the first antenna element 116 and comprises a vertical portion 128 that extends upward from the base substrate 108 along the outer surface of the front substrate 110. The second antenna element 118 also comprises an arm 130 that extends outward from a top end of the vertical portion 128. That arm 130 extends horizontally along the outer surface of both the front substrate 110 and the second lateral substrate 114. Provided at a distal end of the arm 130 on the outer surface of the second lateral substrate 114 is a further meandered portion 132. As shown in
The first antenna element 116 is coupled to a feeding network 134 that includes a microstrip matching line having an inductor 136 and a 50Ω microstrip line 138, which are both formed on the inner surface of the base substrate 108. The second antenna element 118 is coupled to a ground plane 140 that is formed on the outer (bottom) surface of the base substrate 108.
The transmit antenna 104 is also a three-dimensional half-wave dipole antenna and is formed on the base substrate 108 and a rear substrate 142. As shown in
As is further shown in
As is further illustrated in
Unlike the substrates of the repeater 10, the substrates of the repeater 100 have been machined to remove unneeded material. As indicated in
A repeater similar to that shown in
Experiments were performed to detect vibration using a sensor node that comprised a repeater similar to that shown in
It was found in prior work that, by applying DC voltage to the diode doubler input, amplitude modulation can be imparted to the return signal. The voltage changes the diode impedance match with the antennas thus changing the CG of the transceiver. To test the capability to modulate the return signal, the CG was measured for different applied DC voltages and different power levels. As shown in
A piezoelectric thin-film vibration sensor (MiniSense 100 from Measurement Specialties) was used during the testing and was excited at a vibration frequency of 40 Hz and amplitude of 9 g (deflection of 2.5 mm). The output voltage waveform from the sensor was measured using an oscilloscope with a load impedance of 1 M Ω. The voltage waveform is illustrated in
Additional testing was performed to ensure that the integration of the piezoelectric sensor with the transceiver would not produce adverse proximity effects. The sensor was attached inside the repeater structure (i.e., within the cube defined by the repeater) and no degradation in the CG was observed. However, placing this sensor inside the node restricted the fluctuation of the sensor cantilever film that would reduce the generated voltage. In order to increase the voltage delivered to the doubler at a given vibration level, enabling lower vibration frequencies and amplitudes to be detected more easily, the sensor can be designed to have a larger capacitive input.
For the structural health monitoring applications, the vibration frequency and amplitude are both usually very low. Although piezoelectric thin-film sensors are suitable for detecting high frequency stimuli, it was found through analysis and experimental results that mechanical switches can be effectively used to provide low vibration frequency and low amplitude detection capability. The sensors used in the demonstration were the CM1344-1, AG2401-1, and ASLS-2 from Comus International. These sensors offer the additional benefit of being hermitically sealed for long lifetime operation.
The mechanical switches can be integrated into the sensor node in different manners to impart modulation to the return signal. In some embodiments, the switch can be used to open and close the diode DC current return path at the doubler input. This approach decouples the mechanical switch from the RF path, thereby alleviating the need to accurately model switch behavior at microwave frequencies and providing a more controlled transceiver response. As seen appreciated from
The AG2401-1 sensor is a low-cost ball switch that has the ability to detect very low vibration frequencies and amplitudes. Depending on its position and orientation, this sensor can rest in the open or closed state and, once it is driven into vibration, it continually changes its state as long as the motion continues. When it is in the closed state, the contact resistance is less than 1Ω. The sensor's cylindrical case has a diameter of 4 mm and height of 4.5 mm.
In order to detect stronger shocks, as might be excited by an earthquake, the ASLS-2 spring-damped ball switch proved to be effective. This switch needs to be driven by vibration amplitudes between 2-4.9 g in order to change its state. Unlike the AG2401-1 sensor, the ASLS-2 sensor responds only when the vibration waveform is above or below the zero state and not to both positive and negative excursions.
A similarly-sized mercury switch, model CM1344-1, was also tested with the sensor node. This class of switches has a longer operating lifetime than ball switches, and a higher cost. The mercury switch also offers the additional benefit of being responsive to vibration along any axis and may therefore be advantageous for embedded sensing applications where the orientation of the deployed sensor nodes is difficult to control. This switch is normally in the open state and when it is subjected to vibration it intermittently changes its state near the peaks of the excitation wave. Experimental testing with this switch showed that vibration frequencies from 3-40 Hz were detectable, although when the frequency is greater than 10 Hz the vibration frequency is difficult to correlate with the excitation signal frequency.
In some embodiments, the sensor node can simultaneously house multiple sensors without considerable performance degradation. In many situations it is desirable to use a single sensor node for different concurrent applications, thus reducing the size, cost, and identification complexity. Because the AG2401-1 and the ASLS-2 switch sensors change state only when the vibration is within specific non-overlapping amplitude and frequency ranges, they can be integrated in parallel giving the capability of detecting low and high amplitude acceleration. The sensor node was simulated and tested with those mechanical switches in place and no performance sensitivity was observed. In order to have the sensors oriented to be normally in the open state, a specific node orientation might be required. This limitation can be overcome by integrating the sensors in a different topology or using other sensor candidates.
This application is a continuation of U.S. Non-Provisional Application entitled “Compact Repeaters For Wireless Sensing,” having Ser. No. 14/073,256, filed Nov. 6, 2013, which claims priority to U.S. Provisional Application entitled “Compact Repeaters For Wireless Sensing”, having Ser. No. 61/758,545, filed Jan. 30, 2012, both which are hereby incorporated by reference herein in their entirety.
This invention was made with Government support under grant contract numbers ECCS-0925728 and ECCS-0925929 awarded by the National Science Foundation (NSF). The Government has certain rights in the invention.
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Number | Date | Country | |
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61758545 | Jan 2013 | US |
Number | Date | Country | |
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Parent | 14073256 | Nov 2013 | US |
Child | 14222139 | US |