Patent Application
20150244551
1. Field of the Invention
The present invention relates to down-conversion of electromagnetic (EM) signals. More particularly, the present invention relates to down-conversion of EM signals to intermediate frequency signals, to direct down-conversion of EM modulated carrier signals to demodulated baseband signals, and to conversion of FM signals to non-FM signals. The present invention also relates to under-sampling and to transferring energy at aliasing rates.
2. Related Art
Electromagnetic (EM) information signals (baseband signals) include, but are not limited to, video baseband signals, voice baseband signals, computer baseband signals, etc. Baseband signals include analog baseband signals and digital baseband signals.
It is often beneficial to propagate EM signals at higher frequencies. This is generally true regardless of whether the propagation medium is wire, optic fiber, space, air, liquid, etc. To enhance efficiency and practicality, such as improved ability to radiate and added ability for multiple channels of baseband signals, up-conversion to a higher frequency is utilized. Conventional up-conversion processes modulate higher frequency carrier signals with baseband signals. Modulation refers to a variety of techniques for impressing information from the baseband signals onto the higher frequency carrier signals. The resultant signals are referred to herein as modulated carrier signals. For example, the amplitude of an AM carrier signal varies in relation to changes in the baseband signal, the frequency of an FM carrier signal varies in relation to changes in the baseband signal, and the phase of a PM carrier signal varies in relation to changes in the baseband signal.
In order to process the information that was in the baseband signal, the information must be extracted, or demodulated, from the modulated carrier signal. However, because conventional signal processing technology is limited in operational speed, conventional signal processing technology cannot easily demodulate a baseband signal from higher frequency modulated carrier signal directly. Instead, higher frequency modulated carrier signals must be down-converted to an intermediate frequency (IF), from where a conventional demodulator can demodulate the baseband signal.
Conventional down-converters include electrical components whose properties are frequency dependent. As a result, conventional down-converters are designed around specific frequencies or frequency ranges and do not work well outside their designed frequency range.
Conventional down-converters generate unwanted image signals and thus must include filters for filtering the unwanted image signals. However, such filters reduce the power level of the modulated carrier signals. As a result, conventional down-converters include power amplifiers, which require external energy sources.
When a received modulated carrier signal is relatively weak, as in, for example, a radio receiver, conventional down-converters include additional power amplifiers, which require additional external energy.
What is needed includes, without limitation:
an improved method and system for down-converting EM signals;
a method and system for directly down-converting modulated carrier signals to demodulated baseband signals;
a method and system for transferring energy and for augmenting such energy transfer when down-converting EM signals;
a controlled impedance method and system for down-converting an EM signal;
a controlled aperture under-sampling method and system for down-converting an EM signal;
a method and system for down-converting EM signals using a universal down-converter design that can be easily configured for different frequencies;
a method and system for down-converting EM signals using a local oscillator frequency that is substantially lower than the carrier frequency;
a method and system for down-converting EM signals using only one local oscillator;
a method and system for down-converting EM signals that uses fewer filters than conventional down-converters;
a method and system for down-converting EM signals using less power than conventional down-converters;
a method and system for down-converting EM signals that uses less space than conventional down-converters;
a method and system for down-converting EM signals that uses fewer components than conventional down-converters;
a method and system for down-converting EM signals that can be implemented on an integrated circuit (IC); and
a method and system for down-converting EM signals that can also be used as a method and system for up-converting a baseband signal.
Briefly stated, the present invention is directed to methods, systems, and apparatuses for down-converting an electromagnetic (EM), and applications thereof.
Generally, in an embodiment, the invention operates by receiving an EM signal and recursively operating on approximate half cycles of a carrier signal. The recursive operations are typically performed at a sub-harmonic rate of the carrier signal. The invention accumulates the results of the recursive operations and uses the accumulated results to form a down-converted signal.
In an embodiment, the invention down-converts the EM signal to an intermediate frequency (IF) signal.
In another embodiment, the invention down-converts the EM signal to a demodulated baseband information signal.
In another embodiment, the EM signal is a frequency modulated (FM) signal, which is down-converted to a non-FM signal, such as a phase modulated (PM) signal or an amplitude modulated (AM) signal.
The invention is applicable to any type of EM signal, including but not limited to, modulated carrier signals (the invention is applicable to any modulation scheme or combination thereof) and unmodulated carrier signals.
Further features and advantages of the invention, as well as the structure and operation of various embodiments of the invention, are described in detail below with reference to the accompanying drawings. It is noted that the invention is not limited to the specific embodiments described herein. Such embodiments are presented herein for illustrative purposes only. Additional embodiments will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
The drawing in which an element first appears is typically indicated by the leftmost digit(s) in the corresponding reference number.
The present invention will be described with reference to the accompanying drawings wherein:
For illustrative purposes, the operation of the invention is often represented by flowcharts, such as flowchart 1201 in
Various terms used in this application are generally described in this section. The description in this section is provided for illustrative and convenience purposes only, and is not limiting. The meaning of these terms will be apparent to persons skilled in the relevant art(s) based on the entirety of the teachings provided herein. These terms may be discussed throughout the specification with additional detail.
The term modulated carrier signal, when used herein, refers to a carrier signal that is modulated by a baseband signal.
The term unmodulated carrier signal, when used herein, refers to a signal having an amplitude that oscillates at a substantially uniform frequency and phase.
The term baseband signal, when used herein, refers to an information signal including, but not limited to, analog information signals, digital information signals and direct current (DC) information signals.
The term carrier signal, when used herein, and unless otherwise specified when used herein, refers to modulated carrier signals and unmodulated carrier signals, information signals, digital information signals, and direct current (DC) information signals.
The term electromagnetic (EM) signal, when used herein, refers to a signal in the EM spectrum. EM spectrum includes all frequencies greater than zero hertz. EM signals generally include waves characterized by variations in electric and magnetic fields. Such waves may be propagated in any medium, both natural and manmade, including but not limited to air, space, wire, cable, liquid, waveguide, micro-strip, strip-line, optical fiber, etc. Unless stated otherwise, all signals discussed herein are EM signals, even when not explicitly designated as such.
The term intermediate frequency (IF) signal, when used herein, refers to an EM signal that is substantially similar to another EM signal except that the IF signal has a lower frequency than the other signal. An IF signal frequency can be any frequency above zero HZ. Unless otherwise stated, the terms lower frequency, intermediate frequency, intermediate and IF are used interchangeably herein.
The term analog signal, when used herein, refers to a signal that is constant or continuously variable, as contrasted to a signal that changes between discrete states.
The term baseband, when used herein, refers to a frequency band occupied by any generic information signal desired for transmission and/or reception.
The term baseband signal, when used herein, refers to any generic information signal desired for transmission and/or reception.
The term carrier frequency, when used herein, refers to the frequency of a carrier signal. Typically, it is the center frequency of a transmission signal that is generally modulated.
The term carrier signal, when used herein, refers to an EM wave having at least one characteristic that may be varied by modulation, that is capable of carrying information via modulation.
The term demodulated baseband signal, when used herein, refers to a signal that results from processing a modulated signal. In some cases, for example, the demodulated baseband signal results from demodulating an intermediate frequency (IF) modulated signal, which results from down converting a modulated carrier signal. In another case, a signal that results from a combined downconversion and demodulation step.
The term digital signal, when used herein, refers to a signal that changes between discrete states, as contrasted to a signal that is continuous. For example, the voltage of a digital signal may shift between discrete levels.
The term electromagnetic (EM) spectrum, when used herein, refers to a spectrum comprising waves characterized by variations in electric and magnetic fields. Such waves may be propagated in any communication medium, both natural and manmade, including but not limited to air, space, wire, cable, liquid, waveguide, microstrip, stripline, optical fiber, etc. The EM spectrum includes all frequencies greater than zero hertz.
The term electromagnetic (EM) signal, when used herein, refers to a signal in the EM spectrum. Also generally called an EM wave. Unless stated otherwise, all signals discussed herein are EM signals, even when not explicitly designated as such.
The term modulating baseband signal, when used herein, refers to any generic information signal that is used to modulate an oscillating signal, or carrier signal.
It is often beneficial to propagate electromagnetic (EM) signals at higher frequencies. This includes baseband signals, such as digital data information signals and analog information signals. A baseband signal can be up-converted to a higher frequency EM signal by using the baseband signal to modulate a higher frequency carrier signal, FC. When used in this manner, such a baseband signal is herein called a modulating baseband signal FMB.
Modulation imparts changes to the carrier signal FC that represent information in the modulating baseband signal FMB. The changes can be in the form of amplitude changes, frequency changes, phase changes, etc., or any combination thereof. The resultant signal is referred to herein as a modulated carrier signal FMC. The modulated carrier signal FMC includes the carrier signal FC modulated by the modulating baseband signal, FMB, as in:
F
MB combined with FC→FMC
The modulated carrier signal FMC oscillates at, or near the frequency of the carrier signal FC and can thus be efficiently propagated.
Modulating baseband signal FMB can be an analog baseband signal, a digital baseband signal, or a combination thereof.
Digital information includes a plurality of discrete states. For ease of explanation, digital information signals are discussed below as having two discrete states. But the invention is not limited to this embodiment.
Digital modulating baseband signal 310 can change between first state 312 and second state 314 at a data rate, or baud rate, measured as bits per second.
Carrier signal FC is modulated by the modulating baseband signal FMB, by any modulation technique, including, but not limited to, amplitude modulation (AM), frequency modulation (FM), phase modulation (PM), etc., or any combination thereof. Examples are provided below for amplitude modulating, frequency modulating, and phase modulating the analog modulating baseband signal 210 and the digital modulating baseband signal 310, on the carrier signal FC. The examples are used to assist in the description of the invention. The invention is not limited to, or by, the examples.
In amplitude modulation (AM), the amplitude of the modulated carrier signal FMC is a function of the amplitude of the modulating baseband signal FMB.
The analog AM carrier signal 516 oscillates at the frequency of carrier signal 410. The amplitude of the analog AM carrier signal 516 tracks the amplitude of analog modulating baseband signal 210, illustrating that the information contained in the analog modulating baseband signal 210 is retained in the analog AM carrier signal 516.
The digital AM carrier signal 616 oscillates at the frequency of carrier signal 410. The amplitude of the digital AM carrier signal 616 tracks the amplitude of digital modulating baseband signal 310, illustrating that the information contained in the digital modulating baseband signal 310 is retained in the digital AM signal 616. As the digital modulating baseband signal 310 changes states, the digital AM signal 616 shifts amplitudes. Digital amplitude modulation is often referred to as amplitude shift keying (ASK), and the two terms are used interchangeably throughout the specification.
In frequency modulation (FM), the frequency of the modulated carrier signal FMC varies as a function of the amplitude of the modulating baseband signal FMB.
The frequency of the analog FM carrier signal 716 varies as a function of amplitude changes on the analog baseband signal 210. In the illustrated example, the frequency of the analog FM carrier signal 716 varies in proportion to the amplitude of the analog modulating baseband signal 210. Thus, at time t1, the amplitude of the analog baseband signal 210 and the frequency of the analog FM carrier signal 716 are at maximums. At time t3, the amplitude of the analog baseband signal 210 and the frequency of the analog AM carrier signal 716 are at minimums.
The frequency of the analog FM carrier signal 716 is typically centered around the frequency of the carrier signal 410. Thus, at time t2, for example, when the amplitude of the analog baseband signal 210 is at a mid-point, illustrated here as zero volts, the frequency of the analog FM carrier signal 716 is substantially the same as the frequency of the carrier signal 410.
The frequency of the digital FM carrier signal 816 varies as a function of amplitude changes on the digital modulating baseband signal 310. In the illustrated example, the frequency of the digital FM carrier signal 816 varies in proportion to the amplitude of the digital modulating baseband signal 310. Thus, between times t0 and t1, and between times t2 and t4, when the amplitude of the digital baseband signal 310 is at the higher amplitude second state, the frequency of the digital FM carrier signal 816 is at a maximum. Between times t1 and t2, when the amplitude of the digital baseband signal 310 is at the lower amplitude first state, the frequency of the digital FM carrier signal 816 is at a minimum. Digital frequency modulation is often referred to as frequency shift keying (FSK), and the terms are used interchangeably throughout the specification.
Typically, the frequency of the digital FM carrier signal 816 is centered about the frequency of the carrier signal 410, and the maximum and minimum frequencies are equally offset from the center frequency. Other variations can be employed but, for ease of illustration, this convention will be followed herein.
In phase modulation (PM), the phase of the modulated carrier signal FMC varies as a function of the amplitude of the modulating baseband signal FMB.
Generally, the frequency of the analog PM carrier signal 916 is substantially the same as the frequency of carrier signal 410. But the phase of the analog PM carrier signal 916 varies with amplitude changes on the analog modulating baseband signal 210. For relative comparison, the carrier signal 410 is illustrated in
The phase of the analog PM carrier signal 916 varies as a function of amplitude changes of the analog baseband signal 210. In the illustrated example, the phase of the analog PM signal 916 lags by a varying amount as determined by the amplitude of the baseband signal 210. For example, at time t1, when the amplitude of the analog baseband signal 210 is at a maximum, the analog PM carrier signal 916 is in phase with the carrier signal 410. Between times t1 and t3, when the amplitude of the analog baseband signal 210 decreases to a minimum amplitude, the phase of the analog PM carrier signal 916 lags the phase of the carrier signal 410, until it reaches a maximum out of phase value at time t3. In the illustrated example, the phase change is illustrated as approximately 180 degrees. Any suitable amount of phase change, varied in any manner that is a function of the baseband signal, can be utilized.
The frequency of the digital PM carrier signal 1016 is substantially the same as the frequency of carrier signal 410. The phase of the digital PM carrier signal 1016 varies as a function of amplitude changes on the digital baseband signal 310. In the illustrated example, when the digital baseband signal 310 is at the first state 312, the digital PM carrier signal 1016 is out of phase with the carrier signal 410. When the digital baseband signal 310 is at the second state 314, the digital PM carrier signal 1016 is in-phase with the carrier signal 410. Thus, between times t1 and t2, when the amplitude of the digital baseband signal 310 is at the first state 312, the digital PM carrier signal 1016 is out of phase with the carrier signal 410. Between times t0 and t1, and between times t2 and t4, when the amplitude of the digital baseband signal 310 is at the second state 314, the digital PM carrier signal 1016 is in phase with the carrier signal 410.
In the illustrated example, the out of phase value between times t1 and t3 is illustrated as approximately 180 degrees out of phase. Any suitable amount of phase change, varied in any manner that is a function of the baseband signal, can be utilized. Digital phase modulation is often referred to as phase shift keying (PSK), and the terms are used interchangeably throughout the specification.
When the modulated carrier signal FMC is received, it can be demodulated to extract the modulating baseband signal FMB. Because of the typically high frequency of modulated carrier signal FMC, however, it is generally impractical to demodulate the baseband signal FMB directly from the modulated carrier signal FMC. Instead, the modulated carrier signal FMC must be down-converted to a lower frequency signal that contains the original modulating baseband signal.
When a modulated carrier signal is down-converted to a lower frequency signal, the lower frequency signal is referred to herein as an intermediate frequency (IF) signal FIF. The IF signal FIF oscillates at any frequency, or frequency band, below the frequency of the modulated carrier frequency FMC. Down-conversion of FMC to FIF is illustrated as:
FMC→FIF
After FMC is down-converted to the IF modulated carrier signal FIF, FIF can be demodulated to a baseband signal FDMB, as illustrated by:
F
IF
→F
DMB
FDMB is intended to be substantially similar to the modulating baseband signal FMB, illustrating that the modulating baseband signal FMB can be substantially recovered.
It will be emphasized throughout the disclosure that the present invention can be implemented with any type of EM signal, including, but not limited to, modulated carrier signals and unmodulated carrier signals. The above examples of modulated carrier signals are provided for illustrative purposes only. Many variations to the examples are possible. For example, a carrier signal can be modulated with a plurality of the modulation types described above. A carrier signal can also be modulated with a plurality of baseband signals, including analog baseband signals, digital baseband signals, and combinations of both analog and digital baseband signals.
Conventional signal processing techniques follow the Nyquist sampling theorem, which states that, in order to faithfully reproduce a sampled signal, the signal must be sampled at a rate that is greater than twice the frequency of the signal being sampled. When a signal is sampled at less than or equal to twice the frequency of the signal, the signal is said to be under-sampled, or aliased. Conventional signal processing thus teaches away from under-sampling and aliasing, in order to faithfully reproduce a sampled signal.
Contrary to conventional wisdom, the present invention is a method and system for down-converting an electromagnetic (EM) signal by aliasing the EM signal. Aliasing is represented generally in
By taking a carrier and aliasing it at an aliasing rate, the invention can down-convert that carrier to lower frequencies. One aspect that can be exploited by this invention is realizing that the carrier is not the item of interest, the lower baseband signal is of interest to reproduce sufficiently. This baseband signal's frequency content, even though its carrier may be aliased, does satisfy the Nyquist criteria and as a result, the baseband information can be sufficiently reproduced.
In an embodiment, the invention down-converts the EM signal to an intermediate frequency (IF) signal.
In another embodiment, the invention down-converts the EM signal to a demodulated baseband information signal.
In another embodiment, the EM signal is a frequency modulated (FM) signal, which is down-converted to a non-FM signal, such as a phase modulated (PM) signal or an amplitude modulated (AM) signal.
The invention down-converts any type of EM signal, including, but not limited to, modulated carrier signals and unmodulated carrier signals. For ease of discussion, the invention is further described herein using modulated carrier signals for examples. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert signals other than carrier signals as well. The invention is not limited to the example embodiments described above.
In an embodiment, down-conversion is accomplished by under-sampling an EM signal. This is described generally in Section 1.2.2. below and in detail in Section II and its sub-sections. In another embodiment, down-conversion is achieved by transferring non-negligible amounts of energy from an EM signal. This is described generally in Section 1.2.3. below and in detail in Section III.
The term aliasing, as used herein, refers both to down-converting an EM signal by under-sampling the EM signal at an aliasing rate and to down-converting an EM signal by transferring energy from the EM signal at the aliasing rate. Methods for under-sampling an EM signal to down-convert the EM signal are now described at an overview level.
Down-converting by under-sampling is illustrated by 4504 in
In an embodiment, an EM signal is under-sampled at an aliasing rate to down-convert the EM signal to a lower, or intermediate frequency (IF) signal. The EM signal can be a modulated carrier signal or an unmodulated carrier signal. In an exemplary example, a modulated carrier signal FMC is down-converted to an IF signal FIF.
F
MC
→F
IF
This embodiment is illustrated generally by 4508 in
In another embodiment, an EM signal is directly down-converted to a demodulated baseband signal (direct-to-data down-conversion), by under-sampling the EM signal at an aliasing rate. The EM signal can be a modulated EM signal or an unmodulated EM signal. In an exemplary embodiment, the EM signal is the modulated carrier signal FMC, and is directly down-converted to a demodulated baseband signal FDMB.
F
MC
→F
DMB
This embodiment is illustrated generally by 4510 in
In another embodiment, a frequency modulated (FM) carrier signal FFMC is converted to a non-FM signal F(NON-FM), by under-sampling the FM carrier signal FFMC.
F
FMC
→F
(NON-FM)
This embodiment is illustrated generally by 4512 in
The term aliasing, as used herein, refers both to down-converting an EM signal by under-sampling the EM signal at an aliasing rate and to down-converting an EM signal by transferring non-negligible amounts energy from the EM signal at the aliasing rate. Methods for transferring energy from an EM signal to down-convert the EM signal are now described at an overview level. More detailed descriptions are provided in Section III.
Down-converting by transferring energy is illustrated by 4506 in
In an embodiment, EM signal is down-converted to a lower, or intermediate frequency (IF) signal, by transferring energy from the EM signal at an aliasing rate. The EM signal can be a modulated carrier signal or an unmodulated carrier signal. In an exemplary example, a modulated carrier signal FMC is down-converted to an IF signal FIF.
F
MC
→F
IF
This embodiment is illustrated generally by 4514 in
In another embodiment, an EM signal is down-converted to a demodulated baseband signal by transferring energy from the EM signal at an aliasing rate. This embodiment is referred to herein as direct-to-data down-conversion. The EM signal can be a modulated EM signal or an unmodulated EM signal. In an exemplary embodiment, the EM signal is the modulated carrier signal FMC, and is directly down-converted to a demodulated baseband signal FDMB.
F
MC
→F
DMB
This embodiment is illustrated generally by 4516 in
In another embodiment, a frequency modulated (FM) carrier signal FFMC is converted to a non-FM signal F(NON-FM), by transferring energy from the FM carrier signal FFMC at an aliasing rate.
F
FMC
→F
(NON-FM)
The FM carrier signal FFMC can be converted to, for example, a phase modulated (PM) signal or an amplitude modulated (AM) signal.
This embodiment is illustrated generally by 4518 in
In accordance with the definition of aliasing, the aliasing rate is equal to, or less than, twice the frequency of the EM carrier signal. Preferably, the aliasing rate is much less than the frequency of the carrier signal. The aliasing rate is preferably more than twice the highest frequency component of the modulating baseband signal FMB that is to be reproduced. The above requirements are illustrated in EQ. (1).
2·FMC≧FAR>2·(Highest Freq. Component of FMB) EQ. (1)
In other words, by taking a carrier and aliasing it at an aliasing rate, the invention can down-convert that carrier to lower frequencies. One aspect that can be exploited by this invention is that the carrier is not the item of interest; instead the lower baseband signal is of interest to be reproduced sufficiently. The baseband signal's frequency content, even though its carrier may be aliased, satisfies the Nyquist criteria and as a result, the baseband information can be sufficiently reproduced, either as the intermediate modulating carrier signal FIF or as the demodulated direct-to-data baseband signal FDMB.
In accordance with the invention, relationships between the frequency of an EM carrier signal, the aliasing rate, and the intermediate frequency of the down-converted signal, are illustrated in EQ. (2).
F
C
=n·F
AR
±F
IF EQ. (2)
FC is the frequency of the EM carrier signal that is to be aliased;
FAR is the aliasing rate;
n identifies a harmonic or sub-harmonic of the aliasing rate (generally, n=0.5, 1, 2, 3, 4, . . . ); and
FIF is the intermediate frequency of the down-converted signal.
Note that as (n·FAR) approaches FC, FIF approaches zero. This is a special case where an EM signal is directly down-converted to a demodulated baseband signal. This special case is referred to herein as Direct-to-Data down-conversion. Direct-to-Data down-conversion is described in later sections.
High level descriptions, exemplary embodiments and exemplary implementations of the above and other embodiments of the invention are provided in sections below.
The example conventional receiver system 1102 receives an electromagnetic (EM) signal 1104 via an antenna 1106. The EM signal 1104 can include a plurality of EM signals such as modulated carrier signals. For example, the EM signal 1104 includes one or more radio frequency (RF) EM signals, such as a 900 MHZ modulated carrier signal. Higher frequency RF signals, such as 900 MHZ signals, generally cannot be directly processed by conventional signal processors. Instead, higher frequency RF signals are typically down-converted to lower intermediate frequencies (IF) for processing. The receiver system 1102 down-converts the EM signal 1104 to an intermediate frequency (IF) signal 1108n, which can be provided to a signal processor 1110. When the EM signal 1104 includes a modulated carrier signal, the signal processor 1110 usually includes a demodulator that demodulates the IF signal 1108n to a baseband information signal (demodulated baseband signal).
Receiver system 1102 includes an RF stage 1112 and one or more IF stages 1114. The RF stage 1112 receives the EM signal 1104. The RF stage 1112 includes the antenna 1106 that receives the EM signal 1104.
The one or more IF stages 1114a-1114n down-convert the EM signal 1104 to consecutively lower intermediate frequencies. Each of the one or more IF sections 1114a-1114n includes a mixer 1118a-1118n that down-converts an input EM signal 1116 to a lower frequency IF signal 1108. By cascading the one or more mixers 1118a-1118n, the EM signal 1104 is incrementally down-converted to a desired IF signal 1108n.
In operation, each of the one or more mixers 1118 mixes an input EM signal 1116 with a local oscillator (LO) signal 1119, which is generated by a local oscillator (LO) 1120. Mixing generates sum and difference signals from the input EM signal 1116 and the LO signal 1119. For example, mixing an input EM signal 1116a, having a frequency of 900 MHZ, with a LO signal 1119a, having a frequency of 830 MHZ, results in a sum signal, having a frequency of 900 MHZ+830 MHZ=1.73 GHZ, and a difference signal, having a frequency of 900 MHZ-830 MHZ=70 MHZ.
Specifically, in the example of
Generally, it is very difficult, if not impossible, to separate the two 70 MHZ signals. Instead, one or more filters 1122 and 1123 are provided upstream from each mixer 1118 to filter the unwanted frequencies, also known as image frequencies. The filters 1122 and 1123 can include various filter topologies and arrangements such as bandpass filters, one or more high pass filters, one or more low pass filters, combinations thereof, etc.
Typically, the one or more mixers 1118 and the one or more filters 1122 and 1123 attenuate or reduce the strength of the EM signal 1104. For example, a typical mixer reduces the EM signal strength by 8 to 12 dB. A typical filter reduces the EM signal strength by 3 to 6 dB.
As a result, one or more low noise amplifiers (LNAs) 1121 and 1124a-1124n are provided upstream of the one or more filters 1123 and 1122a-1122n. The LNAs and filters can be in reversed order. The LNAs compensate for losses in the mixers 1118, the filters 1122 and 1123, and other components by increasing the EM signal strength prior to filtering and mixing. Typically, for example, each LNA contributes 15 to 20 dB of amplification.
However, LNAs require substantial power to operate. Higher frequency LNAs require more power than lower frequency LNAs. When the receiver system 1102 is intended to be portable, such as a cellular telephone receiver, for example, the LNAs require a substantial portion of the total power.
At higher frequencies, impedance mismatches between the various stages further reduce the strength of the EM signal 1104. In order to optimize power transferred through the receiver system 1102, each component should be impedance matched with adjacent components. Since no two components have the exact same impedance characteristics, even for components that were manufactured with high tolerances, impedance matching must often be individually fine tuned for each receiver system 1102. As a result, impedance matching in conventional receivers tends to be labor intensive and more art than science. Impedance matching requires a significant amount of added time and expense to both the design and manufacture of conventional receivers. Since many of the components, such as LNA, filters, and impedance matching circuits, are highly frequency dependent, a receiver designed for one application is generally not suitable for other applications. Instead, a new receiver must be designed, which requires new impedance matching circuits between many of the components.
Conventional receiver components are typically positioned over multiple IC substrates instead of on a single IC substrate. This is partly because there is no single substrate that is optimal for both RF, IF, and baseband frequencies. Other factors may include the sheer number of components, their various sizes and different inherent impedance characteristics, etc. Additional signal amplification is often required when going from chip to chip. Implementation over multiple substrates thus involves many costs in addition to the cost of the ICs themselves.
Conventional receivers thus require many components, are difficult and time consuming to design and manufacture, and require substantial external power to maintain sufficient signal levels. Conventional receivers are thus expensive to design, build, and use.
In an embodiment, the present invention is implemented to replace many, if not all, of the components between the antenna 1106 and the signal processor 1110, with an aliasing module that includes a universal frequency translator (UFT) module. (More generally, the phrase “universal frequency translator,” “universal frequency translation,” “UFT,” “UFT transform,” and “UFT technology” (or similar phrases) are used herein to refer to the frequency translation technology/concepts described herein.) The UFT is able to down-convert a wide range of EM signal frequencies using very few components. The UFT is easy to design and build, and requires very little external power. The UFT design can be easily tailored for different frequencies or frequency ranges. For example, UFT design can be easily impedance matched with relatively little tuning. In a direct-to-data embodiment of the invention, where an EM signal is directly down-converted to a demodulated baseband signal, the invention also eliminates the need for a demodulator in the signal processor 1110.
When the invention is implemented in a receiver system, such as the receiver system 1102, power consumption is significantly reduced and signal to noise ratio is significantly increased.
In an embodiment, the invention can be implemented and tailored for specific applications with easy to calculate and easy to implement impedance matching circuits. As a result, when the invention is implemented as a receiver, such as the receiver 1102, specialized impedance matching experience is not required.
In conventional receivers, components in the IF sections comprise roughly eighty to ninety percent of the total components of the receivers. The UFT design eliminates the IF section(s) and thus eliminates the roughly eighty to ninety percent of the total components of conventional receivers.
Other advantages of the invention include, but are not limited to:
The invention can be implemented as a receiver with only a single local oscillator;
The invention can be implemented as a receiver with only a single, lower frequency, local oscillator;
The invention can be implemented as a receiver using few filters;
The invention can be implemented as a receiver using unit delay filters;
The invention can be implemented as a receiver that can change frequencies and receive different modulation formats with no hardware changes;
The invention can be also be implemented as frequency up-converter in an EM signal transmitter;
The invention can be also be implemented as a combination up-converter (transmitter) and down-converter (receiver), referred to herein as a transceiver;
The invention can be implemented as a method and system for ensuring reception of a communications signal, as disclosed in co-pending patent application titled, “Method and System for Ensuring Reception of a Communications Signal,” Ser. No. 09/176,415 (now U.S. Pat. No. 6,091,940), incorporated herein by reference in its entirety;
The invention can be implemented in a differential configuration, whereby signal to noise ratios are increased;
A receiver designed in accordance with the invention can be implemented on a single IC substrate, such as a silicon-based IC substrate;
A receiver designed in accordance with the invention and implemented on a single IC substrate, such as a silicon-based IC substrate, can down-convert EM signals from frequencies in the giga Hertz range;
A receiver built in accordance with the invention has a relatively flat response over a wide range of frequencies. For example, in an embodiment, a receiver built in accordance with the invention to operate around 800 MHZ has a substantially flat response (i.e., plus or minus a few dB of power) from 100 MHZ to 1 GHZ. This is referred to herein as a wide-band receiver; and
A receiver built in accordance with the invention can include multiple, user-selectable, Impedance match modules, each designed for a different wide-band of frequencies, which can be used to scan an ultra-wide-band of frequencies.
In an embodiment, the invention down-converts an EM signal to an IF signal by under-sampling the EM signal. This embodiment is illustrated by 4508 in
This embodiment can be implemented with modulated and unmodulated EM signals. This embodiment is described herein using the modulated carrier signal FMC in
The following sections describe example methods for down-converting the modulated carrier signal FMC to the IF signal FIF, according to embodiments of the invention. Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of down-converting an EM signal to an IF signal FIF, according to an embodiment of the invention. In particular, an operational process of under-sampling a modulated carrier signal FMC to down-convert it to the IF signal FIF, is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. This structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all combinations of modulation techniques are valid for this invention. For ease of discussion, the digital AM carrier signal 616 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed flowcharts and descriptions for AM, FM and PM example embodiments. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and unmodulated carrier signals.
The method illustrated in the flowchart 1407 is now described at a high level using the digital AM carrier signal 616 of
The process begins at step 1408, which includes receiving an EM signal. Step 1408 is represented by the digital AM carrier signal 616.
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. When down-converting an EM signal to an IF signal, the frequency or aliasing rate of the pulses 1504 sets the IF.
The intermediate frequency of the down-converted signal FIF, which in this example is the AM intermediate signal 1506, can be determined from EQ. (2), which is reproduced below for convenience.
F
C
=n·F
AR
±F
IF EQ. (2)
A suitable aliasing rate FAR can be determined in a variety of ways. An example method for determining the aliasing rate FAR, is provided below. After reading the description herein, one skilled in the relevant art(s) will understand how to determine appropriate aliasing rates for EM signals, including ones in addition to the modulated carrier signals specifically illustrated herein.
In
Step 1704 includes determining, or selecting, the intermediate frequency. This is the frequency to which the EM signal will be down-converted. The intermediate frequency can be determined, or selected, to match a frequency requirement of a down-stream demodulator. The intermediate frequency can be, for example, 1 MHZ.
Step 1706 includes determining the aliasing rate or rates that will down-convert the EM signal to the IF specified in step 1704.
EQ. (2) can be rewritten as EQ. (3):
n·F
AR
=F
C
±F
IF EQ. (3)
Which can be rewritten as EQ. (4):
(FC±FIF) can be defined as a difference value FDIFF, as illustrated in EQ. (6):
(FC±FIF)=FDIFF EQ. (6)
EQ. (4) can be rewritten as EQ. (7):
From EQ. (7), it can be seen that, for a given n and a constant FAR, FDIFF is constant. For the case of FDIFF=FC−FIF, and for a constant FDIFF, as FC increases, FIF necessarily increases. For the case of FDIFF=FC+FIF, and for a constant FDIFF, as FC increases, FIF necessarily decreases. In the latter case of FDIFF=FC+FIF, any phase or frequency changes on FC correspond to reversed or inverted phase or frequency changes on FIF. This is mentioned to teach the reader that if FDIFF=FC+FIF is used, the above effect will affect the phase and frequency response of the modulated intermediate signal FIF.
EQs. (2) through (7) can be solved for any valid n. A suitable n can be determined for any given difference frequency FDIFF and for any desired aliasing rate FAR(Desired). EQs. (2) through (7) can be utilized to identify a specific harmonic closest to a desired aliasing rate FAR(Desired) that will generate the desired intermediate signal FIF.
An example is now provided for determining a suitable n for a given difference frequency FDIFF and for a desired aliasing rate FAR(Desired). For ease of illustration, only the case of (FC−FIF) is illustrated in the example below.
The desired aliasing rate FAR(Desired) can be, for example, 140 MHZ. Using the previous examples, where the carrier frequency is 901 MHZ and the IF is 1 MHZ, an initial value of n is determined as:
The initial value 6.4 can be rounded up or down to the valid nearest n, which was defined above as including (0.5, 1, 2, 3, . . . ). In this example, 6.4 is rounded down to 6.0, which is inserted into EQ. (5) for the case of (FC−FIF)=FDIFF:
In other words, under-sampling a 901 MHZ EM carrier signal at 150 MHZ generates an intermediate signal at 1 MHZ. When the under-sampled EM carrier signal is a modulated carrier signal, the intermediate signal will also substantially include the modulation. The modulated intermediate signal can be demodulated through any conventional demodulation technique.
Alternatively, instead of starting from a desired aliasing rate, a list of suitable aliasing rates can be determined from the modified form of EQ. (5), by solving for various values of n. Example solutions are listed below.
Solving for n=0.5, 1, 2, 3, 4, 5 and 6:
900 MHZ/0.5=1.8 GHZ (i.e., second harmonic, illustrated in
900 MHZ/1=900 MHZ (i.e., fundamental frequency, illustrated in
900 MHZ/2=450 MHZ (i.e., second sub-harmonic, illustrated in
900 MHZ/3=300 MHZ (i.e., third sub-harmonic, illustrated in
900 MHZ/4=225 MHZ (i.e., fourth sub-harmonic, illustrated in
900 MHZ/5=180 MHZ (i.e., fifth sub-harmonic, illustrated in
The steps described above can be performed for the case of (FC+FIF) in a similar fashion. The results can be compared to the results obtained from the case of (FC−FIF) to determine which provides better result for an application.
In an embodiment, the invention down-converts an EM signal to a relatively standard IF in the range of, for example, 100 KHZ to 200 MHZ. In another embodiment, referred to herein as a small off-set implementation, the invention down-converts an EM signal to a relatively low frequency of, for example, less than 100 KHZ. In another embodiment, referred to herein as a large off-set implementation, the invention down-converts an EM signal to a relatively higher IF signal, such as, for example, above 200 MHZ.
The various off-set implementations provide selectivity for different applications. Generally, lower data rate applications can operate at lower intermediate frequencies. But higher intermediate frequencies can allow more information to be supported for a given modulation technique.
In accordance with the invention, a designer picks an optimum information bandwidth for an application and an optimum intermediate frequency to support the baseband signal. The intermediate frequency should be high enough to support the bandwidth of the modulating baseband signal FMB.
Generally, as the aliasing rate approaches a harmonic or sub-harmonic frequency of the EM signal, the frequency of the down-converted IF signal decreases. Similarly, as the aliasing rate moves away from a harmonic or sub-harmonic frequency of the EM signal, the IF increases.
Aliased frequencies occur above and below every harmonic of the aliasing frequency. In order to avoid mapping other aliasing frequencies in the band of the aliasing frequency (IF) of interest, the IF of interest is preferably not near one half the aliasing rate.
As described in example implementations below, an aliasing module, including a universal frequency translator (UFT) module built in accordance with the invention, provides a wide range of flexibility in frequency selection and can thus be implemented in a wide range of applications. Conventional systems cannot easily offer, or do not allow, this level of flexibility in frequency selection.
Preferably, the under-sampling module 1606 under-samples the EM signal 1304 to down-convert it to the intermediate signal FIF in the manner shown in the operational flowchart 1407 of
The operation of the under-sampling system 1602 is now described with reference to the flowchart 1407 and to the timing diagrams in
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting the EM signal 1304 to the intermediate signal FIF, illustrated in the flowchart 1407 of
1.2.1.1 Operational Description
Operation of the exemplary process of the flowchart 1407 in
1.2.1.1.1 Analog AM Carrier Signal
A process for down-converting the analog AM carrier signal 516 in
The process begins at step 1408, which includes receiving the EM signal. This is represented by the analog AM carrier signal 516 in
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the AM carrier signal 516, the under-sample points 1905 “walk through” the analog AM carrier signal 516. In this example, the under-sample points 1905 “walk through” the analog AM carrier signal 516 at approximately a one megahertz rate. In other words, the under-sample points 1905 occur at different locations on subsequent cycles of the AM carrier signal 516. As a result, the under-sample points 1905 capture varying amplitudes of the analog AM signal 516. For example, under-sample point 1905A has a larger amplitude than under-sample point 1905B.
In
In
The AM intermediate signal 1912 is substantially similar to the AM carrier signal 516, except that the AM intermediate signal 1912 is at the 1 MHZ intermediate frequency. The AM intermediate signal 1912 can be demodulated through any conventional AM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the AM intermediate signal 1910 in
1.2.1.1.2 Digital AM Carrier Signal
A process for down-converting the digital AM carrier signal 616 in
The process begins at step 1408, which includes receiving an EM signal. This is represented by the AM signal 616 in
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the AM carrier signal 616, the under-sample points 1805 walk through the AM carrier signal 616. In other words, the under-sample points 1805 occur at different locations of subsequent cycles of the AM signal 616. As a result, the under-sample points 1805 capture various amplitudes of the AM signal 616. In this example, the under-sample points 1805 walk through the AM carrier signal 616 at approximately a 1 MHZ rate. For example, under-sample point 1805A has a larger amplitude than under-sample point 1805B.
In
In
The AM intermediate signal 1812 is substantially similar to the AM carrier signal 616, except that the AM intermediate signal 1812 is at the 1 MHZ intermediate frequency. The AM intermediate signal 1812 can be demodulated through any conventional AM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the AM intermediate signal 1810 in
1.2.1.2 Structural Description
The operation of the under-sampling system 1602 is now described for the analog AM carrier signal 516, with reference to the flowchart 1407 and to the timing diagrams of
The operation of the under-sampling system 1602 is now described for the digital AM carrier signal 616, with reference to the flowchart 1407 and to the timing diagrams of
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
1.2.2.1 Operational Description
Operation of the exemplary process of the flowchart 1407 in
1.2.2.1.1 Analog FM Carrier Signal
A process for down-converting the analog FM carrier signal 716 to an analog FM intermediate signal is now described with reference to the flowchart 1407 in
The process begins at step 1408, which includes receiving an EM signal. This is represented in
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the FM carrier signal 716, the under-sample points 2005 occur at different locations of subsequent cycles of the under-sampled signal 716. In other words, the under-sample points 2005 walk through the signal 716. As a result, the under-sample points 2005 capture various amplitudes of the FM carrier signal 716.
In
In
The FM intermediate signal 2012 is substantially similar to the FM carrier signal 716, except that the FM intermediate signal 2012 is at the 1 MHZ intermediate frequency. The FM intermediate signal 2012 can be demodulated through any conventional FM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the FM intermediate signal 2010 in
1.2.2.1.2 Digital FM Carrier Signal
A process for down-converting the digital FM carrier signal 816 to a digital FM intermediate signal is now described with reference to the flowchart 1407 in
The process begins at step 1408, which includes receiving an EM signal. This is represented in
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to an intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the FM carrier signal 816, the under-sample points 2105 occur at different locations of subsequent cycles of the FM carrier signal 816. In other words, the under-sample points 2105 walk through the signal 816. As a result, the under-sample points 2105 capture various amplitudes of the signal 816.
In
In
The FM intermediate signal 2112 is substantially similar to the FM carrier signal 816, except that the FM intermediate signal 2112 is at the 1 MHZ intermediate frequency. The FM intermediate signal 2112 can be demodulated through any conventional FM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the FM intermediate signal 2110 in
1.2.2.2 Structural Description
The operation of the under-sampling system 1602 is now described for the analog FM carrier signal 716, with reference to the flowchart 1407 and the timing diagrams of
The operation of the under-sampling system 1602 is now described for the digital FM carrier signal 816, with reference to the flowchart 1407 and the timing diagrams of
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
1.2.3.1 Operational Description
Operation of the exemplary process of the flowchart 1407 in
1.2.3.1.1 Analog PM Carrier Signal
A process for down-converting the analog PM carrier signal 916 to an analog PM intermediate signal is now described with reference to the flowchart 1407 in
The process of down-converting the PM carrier signal 916 to a PM intermediate signal begins at step 1408, which includes receiving an EM signal. This is represented in
Step 1410 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the PM carrier signal 916, the under-sample points 2305 occur at different locations of subsequent cycles of the PM carrier signal 916. As a result, the under-sample points capture various amplitudes of the PM carrier signal 916.
In
In
The analog PM intermediate signal 2312 is substantially similar to the analog PM carrier signal 916, except that the analog PM intermediate signal 2312 is at the 1 MHZ intermediate frequency. The analog PM intermediate signal 2312 can be demodulated through any conventional PM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the analog PM intermediate signal 2310 in
1.2.3.1.2 Digital PM Carrier Signal
A process for down-converting the digital PM carrier signal 1016 to a digital PM intermediate signal is now described with reference to the flowchart 1407 in
The process begins at step 1408, which includes receiving an EM signal. This is represented in
Step 1408 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1412 includes under-sampling the EM signal at the aliasing rate to down-convert the EM signal to an intermediate signal FIF. Step 1412 is illustrated in
Because a harmonic of the aliasing rate is off-set from the PM carrier signal 1016, the under-sample points 2205 occur at different locations of subsequent cycles of the PM carrier signal 1016.
In
In
The digital PM intermediate signal 2212 is substantially similar to the digital PM carrier signal 1016, except that the digital PM intermediate signal 2212 is at the 1 MHZ intermediate frequency. The digital PM carrier signal 2212 can be demodulated through any conventional PM demodulation technique.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the digital PM intermediate signal 2210 in
1.2.3.2 Structural Description
The operation of the under-sampling system 1602 is now described for the analog PM carrier signal 916, with reference to the flowchart 1407 and the timing diagrams of
The operation of the under-sampling system 1602 is now described for the digital PM carrier signal 1016, with reference to the flowchart 1407 and the timing diagrams of
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention. Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. The implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
In an embodiment, the invention directly down-converts an EM signal to a baseband signal, by under-sampling the EM signal. This embodiment is referred to herein as direct-to-data down-conversion and is illustrated in
This embodiment can be implemented with modulated and unmodulated EM signals. This embodiment is described herein using the modulated carrier signal FMC in
The following sections describe example methods for directly down-converting the modulated carrier signal FMC to the demodulated baseband signal FDMB. Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of directly down-converting the modulated carrier signal FMC to the demodulated baseband signal FDMB, according to the invention. In particular, an operational process of directly down-converting the modulated carrier signal FMC to the demodulated baseband signal FDMB is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. The structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all combinations of modulation techniques are valid for this invention. For ease of discussion, the digital AM carrier signal 616 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed descriptions for AM and PM example embodiments. FM presents special considerations that are dealt with separately in Section II.3, below. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and unmodulated carrier signals.
The method illustrated in the flowchart 1413 is now described at a high level using the digital AM carrier signal 616, from
The process of the flowchart 1413 begins at step 1414, which includes receiving an EM signal. Step 1414 is represented by the digital AM carrier signal 616 in
Step 1416 includes receiving an under-sampling signal having an aliasing rate FAR.
F
C
=n·F
AR
±F
IF EQ. (2)
When directly down-converting an EM signal to baseband (i.e., zero IF), EQ. (2) becomes:
F
C
=n·F
AR EQ. (8)
Thus, to directly down-convert the AM signal 616 to a demodulated baseband signal, the aliasing rate is substantially equal to the frequency of the AM signal 616 or to a harmonic or sub-harmonic thereof. Although the aliasing rate is too low to permit reconstruction of higher frequency components of the AM signal 616 (i.e., the carrier frequency), it is high enough to permit substantial reconstruction of the lower frequency modulating baseband signal 310.
Step 1418 includes under-sampling the EM signal at the aliasing rate to directly down-convert it to the demodulated baseband signal FDMB.
In a direct to data embodiment, the frequency of the under-sampling signal 1604 is substantially equal to a harmonic of the EM signal 1304 or, more typically, a sub-harmonic thereof Preferably, the under-sampling module 1606 under-samples the EM signal 1304 to directly down-convert it to the demodulated baseband signal FDMB, in the manner shown in the operational flowchart 1413. But it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 1413. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
The operation of the aliasing system 1602 is now described for the digital AM carrier signal 616, with reference to the flowchart 1413 and to the timing diagrams in
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting the EM signal 1304 to the demodulated baseband signal FDMB, illustrated in the flowchart 1413 of
2.2.1.1 Operational Description
Operation of the exemplary process of the flowchart 1413 in
2.2.1.1.1 Analog AM Carrier Signal
A process for directly down-converting the analog AM carrier signal 516 to a demodulated baseband signal is now described with reference to the flowchart 1413 in
The process begins at step 1414, which includes receiving an EM signal. This is represented by the analog AM carrier signal 516.
Step 1416 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1418 includes under-sampling the EM signal at the aliasing rate to directly down-convert it to the demodulated baseband signal FDMB. Step 1418 is illustrated in
In
In
The demodulated baseband signal 3512 is substantially similar to the modulating baseband signal 210. The demodulated baseband signal 3512 can be processed using any signal processing technique(s) without further down-conversion or demodulation.
The aliasing rate of the under-sampling signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and polarity, as desired.
In the example above, the under-sample points 3505 occur at positive locations of the AM carrier signal 516. Alternatively, the under-sample points 3505 can occur at other locations including negative points of the analog AM carrier signal 516. When the under-sample points 3505 occur at negative locations of the AM carrier signal 516, the resultant demodulated baseband signal is inverted relative to the modulating baseband signal 210.
The drawings referred to herein illustrate direct to data down-conversion in accordance with the invention. For example, the demodulated baseband signal 3510 in
2.2.1.1.2 Digital AM Carrier Signal
A process for directly down-converting the digital AM carrier signal 616 to a demodulated baseband signal is now described with reference to the flowchart 1413 in
The process begins at step 1414, which includes receiving an EM signal. This is represented by the digital AM carrier signal 616.
Step 1416 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1418 includes under-sampling the EM signal at the aliasing rate to directly down-convert it to the demodulated baseband signal FDMB. Step 1418 is illustrated in
In
In
The demodulated baseband signal 3612 is substantially similar to the digital modulating baseband signal 310. The demodulated analog baseband signal 3612 can be processed using any signal processing technique(s) without further down-conversion or demodulation.
The aliasing rate of the under-sampling signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and polarity, as desired.
In the example above, the under-sample points 3605 occur at positive locations of signal portion 3604. Alternatively, the under-sample points 3605 can occur at other locations including negative locations of the signal portion 3604. When the under-sample points 3605 occur at negative points, the resultant demodulated baseband signal is inverted with respect to the modulating baseband signal 310.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the demodulated baseband signal 3610 in
2.2.1.2 Structural Description
The operation of the under-sampling module 1606 is now described for the analog AM carrier signal 516, with reference to the flowchart 1413 and the timing diagrams of
The operation of the under-sampling system 1602 is now described for the digital AM carrier signal 616, with reference to the flowchart 1413 and the timing diagrams of
Example implementations of the under-sampling module 1606 are provided in Sections 4 and 5 below.
2.2.2.1 Operational Description
Operation of the exemplary process of the flowchart 1413 in
2.2.2.1.1 Analog PM Carrier Signal
A process for directly down-converting the analog PM carrier signal 916 to a demodulated baseband signal is now described with reference to the flowchart 1413 in
The process begins at step 1414, which includes receiving an EM signal. This is represented by the analog PM signal 916.
Step 1416 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1418 includes under-sampling the analog PM carrier signal 916 at the aliasing rate to directly down-convert it to a demodulated baseband signal. Step 1418 is illustrated in
Because a harmonic of the aliasing rate is substantially equal to the frequency of the signal 916, or substantially equal to a harmonic or sub-harmonic thereof, essentially no IF is produced. The only substantial aliased component is the baseband signal.
In
In
The demodulated baseband signal 3712 is substantially similar to the analog modulating baseband signal 210. The demodulated baseband signal 3712 can be processed without further down-conversion or demodulation.
The aliasing rate of the under-sampling signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and polarity, as desired.
In the example above, the under-sample points 3705 occur at positive locations of the analog PM carrier signal 916. Alternatively, the under-sample points 3705 can occur at other locations include negative points of the analog PM carrier signal 916. When the under-sample points 3705 occur at negative locations of the analog PM carrier signal 916, the resultant demodulated baseband signal is inverted relative to the modulating baseband signal 210.
The drawings referred to herein illustrate direct to data down-conversion in accordance with the invention. For example, the demodulated baseband signal 3710 in
2.2.2.1.2 Digital PM Carrier Signal
A process for directly down-converting the digital PM carrier signal 1016 to a demodulated baseband signal is now described with reference to the flowchart 1413 in
The process begins at step 1414, which includes receiving an EM signal. This is represented by the digital PM signal 1016.
Step 1416 includes receiving an under-sampling signal having an aliasing rate FAR.
Step 1418 includes under-sampling the digital PM carrier signal 1016 at the aliasing rate to directly down-convert it to a demodulated baseband signal. This is illustrated in
Because a harmonic of the aliasing rate is substantially equal to the frequency of the signal 1016, essentially no IF is produced. The only substantial aliased component is the baseband signal.
In
In
The demodulated baseband signal 3812 is substantially similar to the digital modulating baseband signal 310. The demodulated baseband signal 3812 can be processed without further down-conversion or demodulation.
The aliasing rate of the under-sampling signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and polarity, as desired.
In the example above, the under-sample points 3805 occur at positive locations of the digital PM carrier signal 1016. Alternatively, the under-sample points 3805 can occur at other locations include negative points of the digital PM carrier signal 1016. When the under-sample points 3805 occur at negative locations of the digital PM carrier signal 1016, the resultant demodulated baseband signal is inverted relative to the modulating baseband signal 310.
The drawings referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the demodulated baseband signal 3810 in
2.2.2.2 Structural Description
The operation of the under-sampling system 1602 is now described for the analog PM carrier signal 916, with reference to the flowchart 1413 and the timing diagrams of
The operation of the under-sampling system 1602 is now described for the digital PM carrier signal 1016, with reference to the flowchart 1413 and the timing diagrams of
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. These implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
In an embodiment, the invention down-converts an FM carrier signal FFMC to a non-FM signal F(NON-FM), by under-sampling the FM carrier signal FFMC. This embodiment is illustrated in
In an example embodiment, the FM carrier signal FFMC is down-converted to a phase modulated (PM) signal FPM. In another example embodiment, the FM carrier signal FFMC is down-converted to an amplitude modulated (AM) signal FAM. The invention is not limited to these embodiments. The down-converted signal can be demodulated with any conventional demodulation technique to obtain a demodulated baseband signal FDMB.
The invention can be implemented with any type of FM signal. Exemplary embodiments are provided below for down-converting a frequency shift keying (FSK) signal to a non-FSK signal. FSK is a sub-set of FM, wherein an FM signal shifts or switches between two or more frequencies. FSK is typically used for digital modulating baseband signals, such as the digital modulating baseband signal 310 in
In a first example embodiment, the FSK signal 816 is under-sampled at an aliasing rate that is based on a mid-point between the upper and lower frequencies of the FSK signal 816. When the aliasing rate is based on the mid-point, the FSK signal 816 is down-converted to a phase shift keying (PSK) signal. PSK is a sub-set of phase modulation, wherein a PM signal shifts or switches between two or more phases. PSK is typically used for digital modulating baseband signals. For example, in
In a second example embodiment, the FSK signal 816 is under-sampled at an aliasing rate that is based upon either the upper frequency or the lower frequency of the FSK signal 816. When the aliasing rate is based upon the upper frequency or the lower frequency of the FSK signal 816, the FSK signal 816 is down-converted to an amplitude shift keying (ASK) signal. ASK is a sub-set of amplitude modulation, wherein an AM signal shifts or switches between two or more amplitudes. ASK is typically used for digital modulating baseband signals. For example, in
The following sections describe methods for under-sampling an FM carrier signal FFMC to down-convert it to the non-FM signal F(NON-FM). Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of under-sampling the FM carrier signal FFM to down-convert it to the non-FM signal F(NON-FM), according to the invention. In particular, an operational process for down-converting the FM carrier signal FFM to the non-FM signal F(NON-FM) is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. The structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all forms of frequency modulation techniques are valid for this invention. For ease of discussion, the digital FM carrier (FSK) signal 816 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed flowcharts and descriptions for the FSK signal 816. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of FM signal.
The method illustrated in the flowchart 1419 is described below at a high level for down-converting the FSK signal 816 in
The process of the flowchart 1419 begins at step 1420, which includes receiving an FM signal. This is represented by the FSK signal 816. The FSK signal 816 shifts between an upper frequency 3910 and a lower frequency 3912. In an exemplary embodiment, the upper frequency 3910 is approximately 901 MHZ and the lower frequency 3912 is approximately 899 MHZ.
Step 1422 includes receiving an under-sampling signal having an aliasing rate FAR.
When down-converting an FM carrier signal FFMC to a non-FM signal F(NON-FM), the aliasing rate is substantially equal to a frequency contained within the FM signal, or substantially equal to a harmonic or sub-harmonic thereof. In this example overview embodiment, where the FSK signal 816 is to be down-converted to a PSK signal, the aliasing rate is based on a mid-point between the upper frequency 3910 and the lower frequency 3912. For this example, the mid-point is approximately 900 MHZ. In another embodiment described below, where the FSK signal 816 is to be down-converted to an ASK signal, the aliasing rate is based on either the upper frequency 3910 or the lower frequency 3912, not the mid-point.
Step 1424 includes under-sampling the FM signal FFMC at the aliasing rate to down-convert the FM carrier signal FFMC to the non-FM signal F(NON-FM). Step 1424 is illustrated in
When the upper frequency 3910 is under-sampled, the PSK signal 3904 has a frequency of approximately 1 MHZ and is used as a phase reference. When the lower frequency 3912 is under-sampled, the PSK signal 3904 has a frequency of 1 MHZ and is phase shifted 180 degrees from the phase reference.
The aliasing rate of the under-sampling signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
Detailed exemplary embodiments for down-converting an FSK signal to a PSK signal and for down-converting an FSK signal to an ASK signal are provided below.
In a modulation conversion embodiment, the EM signal 1304 is an FM carrier signal and the under-sampling module 1606 under-samples the FM carrier signal at a frequency that is substantially equal to a harmonic of a frequency within the FM signal or, more typically, substantially equal to a sub-harmonic of a frequency within the FM signal. Preferably, the under-sampling module 1606 under-samples the FM carrier signal FFMC to down-convert it to a non-FM signal F(NON-FM) in the manner shown in the operational flowchart 1419. But it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 1419. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
The operation of the under-sampling system 1602 shall now be described with reference to the flowchart 1419 and the timing diagrams of
Example implementations of the under-sampling module 1606 are provided in Section 4 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting an FM carrier signal FFMC to a non-FM signal, F(NON-FM), illustrated in the flowchart 1419 of
Operation of the exemplary process of the flowchart 1419 in
The FSK signal 816 shifts between a first frequency 4006 and a second frequency 4008. In the exemplary embodiment, the first frequency 4006 is lower than the second frequency 4008. In an alternative embodiment, the first frequency 4006 is higher than the second frequency 4008. For this example, the first frequency 4006 is approximately 899 MHZ and the second frequency 4008 is approximately 901 MHZ.
The process of down-converting the FSK signal 816 to a PSK signal begins at step 1420, which includes receiving an FM signal. This is represented by the FSK signal 816.
Step 1422 includes receiving an under-sampling signal having an aliasing rate FAR.
In this example, where an FSK signal is being down-converted to a PSK signal, the aliasing rate is substantially equal to a harmonic of the mid-point between the frequencies 4006 and 4008 or, more typically, substantially equal to a sub-harmonic of the mid-point between the frequencies 4006 and 4008. In this example, where the first frequency 4006 is 899 MHZ and second frequency 4008 is 901 MHZ, the mid-point is approximately 900 MHZ. Suitable aliasing rates include 1.8 GHZ, 900 MHZ, 450 MHZ, etc. In this example, the aliasing rate of the under-sampling signal 4008 is approximately 450 MHZ.
Step 1424 includes under-sampling the FM signal at the aliasing rate to down-convert it to the non-FM signal F(NON-FM). Step 1424 is illustrated in
In
When the first frequency 4006 is under-sampled, the PSK signal 4012 has a frequency of approximately 1 MHZ and is used as a phase reference. When the second frequency 4008 is under-sampled, the PSK signal 4012 has a frequency of 1 MHZ and is phase shifted 180 degrees from the phase reference.
In
The aliasing rate of the under-sampling signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
In the example above, the under-sample points 4005 occur at positive locations of the FSK signal 816. Alternatively, the under-sample points 4005 can occur at other locations including negative points of the FSK signal 816. When the under-sample points 4005 occur at negative locations of the FSK signal 816, the resultant PSK signal is inverted relative to the PSK signal 4014.
The drawings referred to herein illustrate modulation conversion in accordance with the invention. For example, the PSK signal 4014 in
3.2.1.2 Structural Description
The operation of the under-sampling system 1602 is now described for down-converting the FSK signal 816 to a PSK signal, with reference to the flowchart 1419 and to the timing diagrams of
3.2.2.1 Operational Description
Operation of the exemplary process of
The FSK signal 816 shifts between a first frequency 4106 and a second frequency 4108. In the exemplary embodiment, the first frequency 4106 is lower than the second frequency 4108. In an alternative embodiment, the first frequency 4106 is higher than the second frequency 4108. For this example, the first frequency 4106 is approximately 899 MHZ and the second frequency 4108 is approximately 901 MHZ.
The process of down-converting the FSK signal 816 to an ASK signal begins at step 1420, which includes receiving an FM signal. This is represented by the FSK signal 816.
Step 1422 includes receiving an under-sampling signal having an aliasing rate FAR.
Generally, when down-converting an FM signal to a non-FM signal, the aliasing rate is substantially equal to a harmonic of a frequency within the FM signal or, more typically, to a sub-harmonic of a frequency within the FM signal. When an FSK signal 816 is being down-converted to an ASK signal, the aliasing rate is substantially equal to a harmonic of the first frequency 4106 or the second frequency 4108 or, more typically, substantially equal to a sub-harmonic of the first frequency 4106 or the second frequency 4108. In this example, where the first frequency 4106 is 899 MHZ and the second frequency 4108 is 901 MHZ, the aliasing rate can be substantially equal to a harmonic or sub-harmonic of 899 MHZ or 901 MHZ. In this example the aliasing rate is approximately 449.5 MHZ, which is a sub-harmonic of the first frequency 4106.
Step 1424 includes under-sampling the FM signal at the aliasing rate to down-convert it to a non-FM signal F(NON-FM). Step 1424 is illustrated in
In
In
When down-converting from FM to AM, the aliasing rate of the under-sampling signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and/or polarity, as desired.
In an alternative embodiment, the aliasing rate is based on the second frequency and the resultant ASK signal is reversed relative to the ASK signal 4114.
The drawings referred to herein illustrate modulation conversion in accordance with the invention. For example, the ASK signal 4114 in
3.2.2.2 Structural Description
The operation of the under-sampling system 1602 is now described for down-converting the FSK signal 816 to an ASK signal, with reference to the flowchart 1419 and to the timing diagrams of
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. These implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described in the Sub-Sections above are presented in this section (and its subsections). These implementations are presented herein for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described herein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
The sample and hold system 2602 includes a sample and hold module 2604, which receives the EM signal 1304 and the under-sampling signal 1604. The sample and hold module 2604 under-samples the EM signal at the aliasing rate of the under-sampling signal 1604, as described in the sections above with respect to the flowcharts 1401 in
Preferably, the switch module 2702 and the holding module 2706 under-sample the EM signal 1304 to down-convert it in any of the manners shown in the operation flowcharts 1401, 1407, 1413 and 1419. For example, the sample and hold module 2604 can receive and under-sample any of the modulated carrier signal signals described above, including, but not limited to, the analog AM signal 516, the digital AM signal 616, the analog FM signal 716, the digital FM signal 816, the analog PM signal 916, the digital PM signal 1016, etc., and any combinations thereof.
The switch module 2702 and the holding module 2706 down-convert the EM signal 1304 to an intermediate signal, to a demodulated baseband or to a different modulation scheme, depending upon the aliasing rate.
For example, operation of the switch module 2702 and the holding module 2706 are now described for down-converting the EM signal 1304 to an intermediate signal, with reference to the flowchart 1407 and the example timing diagrams in
In step 1408, the switch module 2702 receives the EM signal 1304 (
The holding module 2706 substantially holds or maintains each under-sampled amplitude until a subsequent under-sample. (
Preferably, the break-before-make under-sampling system 2401 under-samples the EM signal 1304 to down-convert it in any of the manners shown in the operation flowcharts 1401, 1407, 1413 and 1419. For example, the sample and hold module 2604 can receive and under-sample any of the unmodulated or modulated carrier signal signals described above, including, but not limited to, the analog AM signal 516, the digital AM signal 616, the analog FM signal 716, the digital FM signal 816, the analog PM signal 916, the digital PM signal 1016, etc., and combinations thereof.
The break-before-make under-sampling system 2401 down-converts the EM signal 1304 to an intermediate signal, to a demodulated baseband or to a different modulation scheme, depending upon the aliasing rate.
Prior to time t0, the normally open switch 2404 and the normally closed switch 2406 are at their normal states.
At time t0, the isolation signal 2412 in
At time t1, the under-sampling signal 1604 in
Prior to t2, the under-sampling signal 1604 in
At time t2, the isolation signal 2412 in
The break-before-make under-sampling system 2401 includes a holding module 2416, which can be similar to the holding module 2706 in
The switch module 2702 in
The switch device 2810 (e.g., switch modules 2702, 2404 and 2406) can be implemented with any type of suitable switch device, including, but not limited to mechanical switch devices and electrical switch devices, optical switch devices, etc., and combinations thereof. Such devices include, but are not limited to transistor switch devices, diode switch devices, relay switch devices, optical switch devices, micro-machine switch devices, etc.
In an embodiment, the switch module 2810 can be implemented as a transistor, such as, for example, a field effect transistor (FET), a bi-polar transistor, or any other suitable circuit switching device.
In
It should be understood that the illustration of the switch module 2810 as a FET 2802 in
In
In
The holding modules 2706 and 2416 preferably captures and holds the amplitude of the original, unaffected, EM signal 1304 within the short time frame of each negligible aperture under-sampling signal pulse.
In an exemplary embodiment, holding modules 2706 and 2416 are implemented as a reactive holding module 2901 in
In an embodiment, the holding modules 2706 and 2416 include one or more capacitive holding elements, illustrated in
In an alternative embodiment, the holding modules 2706 and 2416 include one or more inductive holding elements, illustrated in
In an alternative embodiment, the holding modules 2706 and 2416 include a combination of one or more capacitive holding elements and one or more inductive holding elements, illustrated in
In an embodiment, the optional under-sampling signal module 3002 includes an aperture generator, an example of which is illustrated in
The width or aperture of the pulses 2926 is determined by delay through the branch 2922 of the aperture generator 2920. Generally, as the desired pulse width decreases, the tolerance requirements of the aperture generator 2920 increase. In other words, to generate negligible aperture pulses for a given input EM frequency, the components utilized in the example aperture generator 2920 require greater reaction times, which are typically obtained with more expensive elements, such as gallium arsenide (GaAs), etc.
The example logic and implementation shown in the aperture generator 2920 are provided for illustrative purposes only, and are not limiting. The actual logic employed can take many forms. The example aperture generator 2920 includes an optional inverter 2928, which is shown for polarity consistency with other examples provided herein. An example implementation of the aperture generator 2920 is illustrated in
Additional examples of aperture generation logic is provided in
In an embodiment, the input signal 2924 is generated externally of the under-sampling signal module 3002, as illustrated in
The type of down-conversion performed by the under-sampling system 3001 depends upon the aliasing rate of the under-sampling signal 1604, which is determined by the frequency of the pulses 2926. The frequency of the pulses 2926 is determined by the frequency of the input signal 2924. For example, when the frequency of the input signal 2924 is substantially equal to a harmonic or a sub-harmonic of the EM signal 1304, the EM signal 1304 is directly down-converted to baseband (e.g. when the EM signal is an AM signal or a PM signal), or converted from FM to a non-FM signal. When the frequency of the input signal 2924 is substantially equal to a harmonic or a sub-harmonic of a difference frequency, the EM signal 1304 is down-converted to an intermediate signal.
The optional under-sampling signal module 3002 can be implemented in hardware, software, firmware, or any combination thereof.
The holding module 4206 can be implemented as described above with reference to
Operation of the inverted sample and hold system 4201 is illustrated in
The inverted sample and hold system 4201 can be used to down-convert any type of EM signal, including modulated carrier signals and unmodulated carrier signals, to IF signals and to demodulated baseband signals.
The implementations described above are provided for purposes of illustration. These implementations are not intended to limit the invention. Alternate implementations, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
The methods and systems described in sections above can be optionally optimized with one or more of the optimization methods or systems described below.
In an embodiment, the optional under-sampling signal module 3002 in
The aliasing rate is twice the frequency of the input oscillating signal Fosc 3106, as shown by EQ. (9) below.
F
AR=2·Fosc EQ. (9)
The aperture width of the aliasing pulses is determined by the delay through a first inverter 3108 of
The invention can be implemented in a variety of differential configurations. Differential configurations are useful for reducing common mode noise. This can be very useful in receiver systems where common mode interference can be caused by intentional or unintentional radiators such as cellular phones, CB radios, electrical appliances etc. Differential configurations are also useful in reducing any common mode noise due to charge injection of the switch in the switch module or due to the design and layout of the system in which the invention is used. Any spurious signal that is induced in equal magnitude and equal phase in both input leads of the invention will be substantially reduced or eliminated. Some differential configurations, including some of the configurations below, are also useful for increasing the voltage and/or for increasing the power of the down-converted signal 1308A. While an example of a differential under-sampling module is shown below, the example is shown for the purpose of illustration, not limitation. Alternate embodiments (including equivalents, extensions, variations, deviations, etc.) of the embodiment described herein will be apparent to those skilled in the relevant art based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
One or both of the inputs 4404 and 4406 are coupled to an EM signal source. For example, the inputs can be coupled to an EM signal source, wherein the input voltages at the inputs 4404 and 4406 are substantially equal in amplitude but 180 degrees out of phase with one another. Alternatively, where dual inputs are unavailable, one of the inputs 4404 and 4406 can be coupled to ground.
In operation, when the switch module 4416 is closed, the holding modules 4414 and 4420 are in series and, provided they have similar capacitive values, they charge to equal amplitudes but opposite polarities. When the switch module 4416 is open, the voltage at the output 4408 is relative to the input 4404, and the voltage at the output 4410 is relative to the voltage at the input 4406.
Portions of the voltages at the outputs 4408 and 4410 include voltage resulting from charge stored in the holding modules 4414 and 4420, respectively, when the switch module 4416 was closed. The portions of the voltages at the outputs 4408 and 4410 resulting from the stored charge are generally equal in amplitude to one another but 180 degrees out of phase.
Portions of the voltages at the outputs 4408 and 4410 also include ripple voltage or noise resulting from the switching action of the switch module 4416. But because the switch module is positioned between the two outputs, the noise introduced by the switch module appears at the outputs 4408 and 4410 as substantially equal and in-phase with one another. As a result, the ripple voltage can be substantially filtered out by inverting the voltage at one of the outputs 4408 or 4410 and adding it to the other remaining output. Additionally, any noise that is impressed with substantially equal amplitude and equal phase onto the input terminals 4404 and 4406 by any other noise sources will tend to be canceled in the same way.
The differential system 4402 is effective when used with a differential front end (inputs) and a differential back end (outputs). It can also be utilized in the following configurations, for example:
a) A single-input front end and a differential back end; and
b) A differential front end and single-output back end.
Examples of these system are provided below.
The outputs 4408 and 4410 are coupled to a differential circuit 4444 such as a filter, which preferably inverts one of the outputs 4408 or 4410 and adds it to the other output 4408 or 4410. This substantially cancels common mode noise generated by the switch module 4416. The differential circuit 4444 preferably filters the higher frequency components of the EM signal 1304 that pass through the holding modules 4414 and 4420. The resultant filtered signal is output as the down-converted signal 1308A.
The down-converted signal 1308A may be smoothed by filtering as desired. The differential circuit 4444 implemented as a filter in
Some of the characteristics of the down-converted signal 1308A depend upon characteristics of a load placed on the down-converted signal 1308A. For example, in an embodiment, when the down-converted signal 1308A is coupled to a high impedance load, the charge that is applied to a holding module such as holding module 2706 in
The down-converted signal 1308A can be buffered with a high impedance amplifier, if desired.
Alternatively, or in addition to buffering the down-converted signal 1308A, the input EM signal may be buffered or amplified by a low noise amplifier.
Generally, the amplitude of the down-converted signal 1308A varies as a function of the frequency and phase differences between the EM signal 1304 and the under-sampling signal 1604. In an embodiment, the down-converted signal 1308A is used as the feedback 3006 to control the frequency and phase relationship between the EM signal 1304 and the under-sampling signal 1604. This can be accomplished using the example block diagram shown in
In the example of
DAC 3206 controls an input to a voltage controlled oscillator, VCO 3208. VCO 3208 controls a frequency input of a pulse generator 3210, which, in an embodiment, is substantially similar to the pulse generator shown in
In an embodiment, the state machine 3204 operates in accordance with the state machine flowchart 3220 in
The amplitude of the down-converted signal 1308A can be made to vary with the amplitude of the under-sampling signal 1604. In an embodiment where Switch Module 2702 is a FET as shown in
The energy transfer embodiments of the invention provide enhanced signal to noise ratios and sensitivity to very small signals, as well as permitting the down-converted signal to drive lower impedance loads unassisted. The energy transfer aspects of the invention are represented generally by 4506 in
Section II above disclosed methods and systems for down-converting an EM signal by under-sampling. The under-sampling systems utilize a sample and hold system controlled by an under-sampling signal. The under-sampling signal includes a train of pulses having negligible apertures that tend towards zero time in duration. The negligible aperture pulses minimize the amount of energy transferred from the EM signal. This protects the under-sampled EM signal from distortion or destruction. The negligible aperture pulses also make the sample and hold system a high impedance system. An advantage of under-sampling is that the high impedance input allows accurate voltage reproduction of the under-sampled EM signal. The methods and systems disclosed in Section II are thus useful for many situations including, but not limited to, monitoring EM signals without distorting or destroying them.
Because the under-sampling systems disclosed in Section II transfer only negligible amounts of energy, they are not suitable for all situations. For example, in radio communications, received radio frequency (RF) signals are typically very weak and must be amplified in order to distinguish them over noise. The negligible amounts of energy transferred by the under-sampling systems disclosed in Section II may not be sufficient to distinguish received RF signals over noise.
In accordance with an aspect of the invention, methods and systems are disclosed below for down-converting EM signals by transferring non-negligible amounts of energy from the EM signals. The resultant down-converted signals have sufficient energy to allow the down-converted signals to be distinguishable from noise. The resultant down-converted signals also have sufficient energy to drive lower impedance circuits without buffering.
Down-converting by transferring energy is introduced below in an incremental fashion to distinguish it from under-sampling. The introduction begins with further descriptions of under-sampling.
In an under-sampling environment, the holding capacitance 7808 preferably has a small capacitance value. This allows the holding capacitance 7808 to substantially charge to the voltage of the input EM signal 7804 during the negligible apertures of the under-sampling signal pulses. For example, in an embodiment, the holding capacitance 7808 has a value in the range of 1 pF. Other suitable capacitance values can be used to achieve substantially the voltage of the original unaffected input signal. Various capacitances can be employed for certain effects, which are described below. The under-sampling system is coupled to a load 7812. In
When the load 7812 is a high impedance load, the holding capacitance 7808 does not significantly discharge between pulses 7904. As a result, charge that is transferred to the holding capacitance 7808 during a pulse 7904 tends to “hold” the voltage value sampled constant at the terminal 7816 until the next pulse 7904. When voltage of the input EM signal 7804 changes between pulses 7904, the holding capacitance 7808 substantially attains the new voltage and the resultant voltage at the terminal 7816 forms a stair step pattern, as illustrated in
Note that the voltage level of the down-converted signals illustrated in
0.1.1.1 Effects of Lowering the Impedance of the Load
Effects of lowering the impedance of the load 7812 are now described.
When the load 7812 is a low impedance load, the holding capacitance 7808 is significantly discharged by the load between pulses 8004 (
0.1.1.2 Effects of Increasing the Value of the Holding Capacitance
Effects of increasing the value of the holding capacitance 7808, while having to drive a low impedance load 7812, is now described.
Recall that when the load 7812 is a low impedance load, the holding capacitance 7808 is significantly discharged by the load between pulses 8104 (
In
In summary, under-sampling systems, such as the under-sampling system 7802 illustrated in
In an embodiment, the present invention transfers energy from an EM signal by utilizing an energy transfer signal instead of an under-sampling signal. Unlike under-sampling signals that have negligible aperture pulses, the energy transfer signal includes a train of pulses having non-negligible apertures that tend away from zero. This provides more time to transfer energy from an EM input signal. One direct benefit is that the input impedance of the system is reduced so that practical impedance matching circuits can be implemented to further improve energy transfer and thus overall efficiency. The non-negligible transferred energy significantly improves the signal to noise ratio and sensitivity to very small signals, as well as permitting the down-converted signal to drive lower impedance loads unassisted. Signals that especially benefit include low power ones typified by RF signals. One benefit of a non-negligible aperture is that phase noise within the energy transfer signal does not have as drastic of an effect on the down-converted output signal as under-sampling signal phase noise or conventional sampling signal phase noise does on their respective outputs.
The energy transfer system 8202 receives an energy transfer signal 8210, which controls the switch module 8206. The energy transfer signal 8210 includes a train of energy transfer pulses having non-negligible pulse widths that tend away from zero time in duration. The non-negligible pulse widths can be any non-negligible amount. For example, the non-negligible pulse widths can be ½ of a period of the input EM signal. Alternatively, the non-negligible pulse widths can be any other fraction of a period of the input EM signal, or a multiple of a period plus a fraction. In an example embodiment, the input EM signal is approximately 900 MHZ and the non-negligible pulse width is approximately 550 pico seconds. Any other suitable non-negligible pulse duration can be used.
In an energy transfer environment, the storage module, illustrated in
One benefit of the energy transfer system 8202 is that, even when the input EM signal 8204 is a very small signal, the energy transfer system 8202 transfers enough energy from the input EM signal 8204 that the input EM signal can be efficiently down-converted.
The energy transfer system 8202 is coupled to a load 8212. Recall from the overview of under-sampling that loads can be classified as high impedance loads or low impedance loads. A high impedance load is one that is relatively insignificant to an output drive impedance of the system for a given output frequency. A low impedance load is one that is relatively significant. Another benefit of the energy transfer system 8202 is that the non-negligible amounts of transferred energy permit the energy transfer system 8202 to effectively drive loads that would otherwise be classified as low impedance loads in under-sampling systems and conventional sampling systems. In other words, the non-negligible amounts of transferred energy ensure that, even for lower impedance loads, the storage capacitance 8208 accepts and maintains sufficient energy or charge to drive the load 8202. This is illustrated below in the timing diagrams of
The energy transfer aspects of the invention are represented generally by 4506 in
In an embodiment, the invention down-converts an EM signal to an IF signal by transferring energy from the EM signal at an aliasing rate. This embodiment is illustrated by 4514 in
This embodiment can be implemented with any type of EM signal, including, but not limited to, modulated carrier signals and unmodulated carrier signals. This embodiment is described herein using the modulated carrier signal FMC in
The following sections describe methods for down-converting an EM signal to an IF signal FIF by transferring energy from the EM signal at an aliasing rate. Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of down-converting an EM signal to an IF signal FIF by transferring energy, according to the invention. In particular, an operational process of down-converting the modulated carrier signal FMC to the IF modulated carrier signal FIF, by transferring energy, is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. This structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all combinations of modulation techniques are valid for this invention. For ease of discussion, the digital AM carrier signal 616 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed flowcharts and descriptions for AM, FM and PM example embodiments. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and unmodulated carrier signals.
The method illustrated in the flowchart 4607 is now described at a high level using the digital AM carrier signal 616 of
The process begins at step 4608, which includes receiving an EM signal. Step 4608 is illustrated by the digital AM carrier signal 616. The digital AM carrier signal 616 of
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
The energy transfer pulses 4704 repeat at the aliasing rate. A suitable aliasing rate can be determined or selected as described below. Generally, when down-converting an EM signal to an intermediate signal, the aliasing rate is substantially equal to a difference frequency, which is described below, or substantially equal to a harmonic or, more typically, a sub-harmonic of the difference frequency.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF.
The intermediate frequency of the down-converted signal FIF, which, in this example, is the intermediate signal 4706 and 4708, can be determined from EQ. (2), which is reproduced below for convenience.
F
C
=n·F
AR
±F
IF EQ. (2)
A suitable aliasing rate FAR can be determined in a variety of ways. An example method for determining the aliasing rate FAR, is provided below. After reading the description herein, one skilled in the relevant art(s) will understand how to determine appropriate aliasing rates for EM signals, including ones in addition to the modulated carrier signals specifically illustrated herein.
In
Step 4804 includes determining, or selecting, the intermediate frequency. This is the frequency to which the EM signal will be down-converted The intermediate frequency can be determined, or selected, to match a frequency requirement of a down-stream demodulator. The intermediate frequency can be, for example, 1 MHZ.
Step 4806 includes determining the aliasing rate or rates that will down-convert the EM signal to the IF specified in step 4804.
EQ. (2) can be rewritten as EQ. (3):
n·F
AR
−F
C
±F
IF EQ. (3)
Which can be rewritten as EQ. (4):
(FC±FIF) can be defined as a difference value FDIFF, as illustrated in EQ. (6):
(FC±FIF)=FDIFF EQ. (6)
EQ. (4) can be rewritten as EQ. (7):
From EQ. (7), it can be seen that, for a given n and a constant FAR, FDIFF is constant. For the case of FDIFF=FC−FIF, and for a constant FDIFF, as FC increases, FIF necessarily increases. For the case of FDIFF=FC→FIF, and for a constant FDIFF, as FC increases, FIF necessarily decreases. In the latter case of FDIFF=FC+FIF, any phase or frequency changes on FC correspond to reversed or inverted phase or frequency changes on FIF. This is mentioned to teach the reader that if FDIFF=FC+FIF is used, the above effect will occur to the phase and frequency response of the modulated intermediate signal FIF.
EQs. (2) through (7) can be solved for any valid n. A suitable n can be determined for any given difference frequency FDIFF and for any desired aliasing rate FAR(Desired). EQs. (2) through (7) can be utilized to identify a specific harmonic closest to a desired aliasing rate FAR(Desired) that will generate the desired intermediate signal FIF.
An example is now provided for determining a suitable n for a given difference frequency FDIFF and for a desired aliasing rate FAR(Desired). For ease of illustration, only the case of (FC−FIF) is illustrated in the example below.
The desired aliasing rate FAR(Desired) can be, for example, 140 MHZ. Using the previous examples, where the carrier frequency is 901 MHZ and the IF is 1 MHZ, an initial value of n is determined as:
The initial value 6.4 can be rounded up or down to the valid nearest n, which was defined above as including (0.5, 1, 2, 3, . . . ). In this example, 6.4 is rounded down to 6.0, which is inserted into EQ. (5) for the case of (FC−FIF)=
In other words, transferring energy from a 901 MHZ EM carrier signal at 150 MHZ generates an intermediate signal at 1 MHZ. When the EM carrier signal is a modulated carrier signal, the intermediate signal will also substantially include the modulation. The modulated intermediate signal can be demodulated through any conventional demodulation technique.
Alternatively, instead of starting from a desired aliasing rate, a list of suitable aliasing rates can be determined from the modified form of EQ. (5), by solving for various values of n. Example solutions are listed below.
Solving for n=0.5, 1, 2, 3, 4, 5 and 6:
900 MHZ/0.5=1.8 GHZ (i.e., second harmonic);
900 MHZ/1=900 MHZ (i.e., fundamental frequency);
900 MHZ/2=450 MHZ (i.e., second sub-harmonic);
900 MHZ/3=300 MHZ (i.e., third sub-harmonic);
900 MHZ/4=225 MHZ (i.e., fourth sub-harmonic);
900 MHZ/5=180 MHZ (i.e., fifth sub-harmonic); and
900 MHZ/6=150 MHZ (i.e., sixth sub-harmonic).
The steps described above can be performed for the case of (FC→FIF) in a similar fashion. The results can be compared to the results obtained from the case of (FC−FIF) to determine which provides better result for an application.
In an embodiment, the invention down-converts an EM signal to a relatively standard IF in the range of, for example, 100 KHZ to 200 MHZ. In another embodiment, referred to herein as a small off-set implementation, the invention down-converts an EM signal to a relatively low frequency of, for example, less than 100 KHZ. In another embodiment, referred to herein as a large off-set implementation, the invention down-converts an EM signal to a relatively higher IF signal, such as, for example, above 200 MHZ.
The various off-set implementations provide selectivity for different applications. Generally, lower data rate applications can operate at lower intermediate frequencies. But higher intermediate frequencies can allow more information to be supported for a given modulation technique.
In accordance with the invention, a designer picks an optimum information bandwidth for an application and an optimum intermediate frequency to support the baseband signal. The intermediate frequency should be high enough to support the bandwidth of the modulating baseband signal FMB.
Generally, as the aliasing rate approaches a harmonic or sub-harmonic frequency of the EM signal, the frequency of the down-converted IF signal decreases. Similarly, as the aliasing rate moves away from a harmonic or sub-harmonic frequency of the EM signal, the IF increases.
Aliased frequencies occur above and below every harmonic of the aliasing frequency. In order to avoid mapping other aliasing frequencies in the band of the aliasing frequency (IF) of interest, the IF of interest should not be near one half the aliasing rate.
As described in example implementations below, an aliasing module, including a universal frequency translator (UFT) module built in accordance with the invention provides a wide range of flexibility in frequency selection and can thus be implemented in a wide range of applications. Conventional systems cannot easily offer, or do not allow, this level of flexibility in frequency selection.
Preferably, the energy transfer module 6304 transfers energy from the EM signal 1304 to down-convert it to the intermediate signal FIF in the manner shown in the operational flowchart 4607 of
The operation of the energy transfer system 6302 is now described in detail with reference to the flowchart 4607 and to the timing diagrams illustrated in
Example implementations of the energy transfer system 6302 are provided in Sections 4 and 5 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting the EM signal 1304 by transferring energy can be implemented with any type of EM signal, including modulated carrier signals and unmodulated carrier signals. For example, the method of the flowchart 4601 can be implemented to down-convert AM signals, FM signals, PM signals, etc., or any combination thereof. Operation of the flowchart 4601 of
1.2.1.1 Operational Description
Operation of the exemplary process of the flowchart 4607 in
1.2.1.1.1 Analog AM Carrier Signal
A process for down-converting the analog AM carrier signal 516 in
The process begins at step 4608, which includes receiving the EM signal. This is represented by the analog AM carrier signal 516.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate signal FR. In
The down-converted signal 5012 includes portions 5010A, which correlate with the energy transfer pulses 5007 in
Because a harmonic of the aliasing rate is off-set from the analog AM carrier signal 516, the energy transfer pulses 5007 “walk through” the analog AM carrier signal 516 at the difference frequency FDIFF. In other words, the energy transfer pulses 5007 occur at different locations of subsequent cycles of the AM carrier signal 516. As a result, the energy transfer pulses 5007 capture varying amounts of energy from the analog AM carrier signal 516, as illustrated by portions 5010A, which provides the AM intermediate signal 5012 with an oscillating frequency FIF.
In
The present invention can output the unfiltered AM intermediate signal 5014, the filtered AM intermediate signal 5016, a partially filtered AM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the AM intermediate signals 5014 in
1.2.1.1.2 Digital AM Carrier Signal
A process for down-converting the digital AM carrier signal 616 to a digital AM intermediate signal is now described for the flowchart 4607 in
The process begins at step 4608, which includes receiving an EM signal. This is represented by the digital AM carrier signal 616.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to the intermediate signal FIF. In
The down-converted signal 5112 includes portions 5110A, which correlate with the energy transfer pulses 5107 in
Because a harmonic of the aliasing rate is off-set from the frequency of the digital AM carrier signal 616, the energy transfer pulses 5107 “walk through” the digital AM signal 616 at the difference frequency FDIFF. In other words, the energy transfer pulse 5107 occur at different locations of subsequent cycles of the digital AM carrier signal 616. As a result, the energy transfer pulses 5107 capture varying amounts of energy from the digital AM carrier signal 616, as illustrated by portions 5110, which provides the AM intermediate signal 5112 with an oscillating frequency FIF.
In
The present invention can output the unfiltered AM intermediate signal 5114, the filtered AM intermediate signal 5116, a partially filtered AM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the AM intermediate signals 5114 in
1.2.1.2 Structural Description
The operation of the energy transfer system 6302 is now described for the analog AM carrier signal 516, with reference to the flowchart 4607 and to the timing diagrams in
The operation of the energy transfer system 6302 is now described for the digital AM carrier signal 616, with reference to the flowchart 1401 and the timing diagrams in
Example embodiments of the energy transfer module 6304 are disclosed in Sections 4 and 5 below.
1.2.2.1 Operational Description
Operation of the exemplary process of the flowchart 4607 in
1.2.2.1.1 Analog FM Carrier Signal
A process for down-converting the analog FM carrier signal 716 in
The process begins at step 4608, which includes receiving an EM signal. This is represented by the analog FM carrier signal 716.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate signal FIF. In
The down-converted signal 5212 includes portions 5210A, which correlate with the energy transfer pulses 5207 in
Because a harmonic of the aliasing rate is off-set from the frequency of the analog FM carrier signal 716, the energy transfer pulses 5207 “walk through” the analog FM carrier signal 716 at the difference frequency FDIFF. In other words, the energy transfer pulse 5207 occur at different locations of subsequent cycles of the analog FM carrier signal 716. As a result, the energy transfer pulses 5207 capture varying amounts of energy from the analog FM carrier signal 716, as illustrated by portions 5210, which provides the FM intermediate signal 5212 with an oscillating frequency FIF.
In
The present invention can output the unfiltered FM intermediate signal 5214, the filtered FM intermediate signal 5216, a partially filtered FM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the FM intermediate signals 5214 in
1.2.2.1.2 Digital FM Carrier Signal
A process for down-converting the digital FM carrier signal 816 in
The process begins at step 4608, which includes receiving an EM signal. This is represented by the digital FM carrier signal 816.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to the an intermediate signal FIF. In
Portions 5310A represent energy transferred from the digital FM carrier signal 816 to a storage device, while simultaneously driving an output load. The portions 5310A occur when a switching module is closed by the energy transfer pulses 5307.
Portions 5310B represent energy stored in a storage device continuing to drive the load. Portions 5310B occur when the switching module is opened after energy transfer pulses 5307.
Because a harmonic of the aliasing rate is off-set from the frequency of the digital FM carrier signal 816, the energy transfer pulses 5307 “walk through” the digital FM carrier signal 816 at the difference frequency FDIFF. In other words, the energy transfer pulse 5307 occur at different locations of subsequent cycles of the digital FM carrier signal 816. As a result, the energy transfer pulses 5307 capture varying amounts of energy from the digital FM carrier signal 816, as illustrated by portions 5310, which provides the FM intermediate signal 5312 with an oscillating frequency FIF.
In
The present invention can output the unfiltered FM intermediate signal 5314, the filtered FM intermediate signal 5316, a partially filtered FM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the FM intermediate signals 5314 in
1.2.2.2 Structural Description
The operation of the energy transfer system 6302 is now described for the analog FM carrier signal 716, with reference to the flowchart 4607 and the timing diagrams in
The operation of the energy transfer system 6302 is now described for the digital FM carrier signal 816, with reference to the flowchart 4607 and the timing diagrams in
Example embodiments of the energy transfer module 6304 are disclosed in Sections 4 and 5 below.
1.2.3.1 Operational Description
Operation of the exemplary process of the flowchart 4607 in
1.2.3.1.1 Analog PM Carrier Signal
A process for down-converting the analog PM carrier signal 916 in
The process begins at step 4608, which includes receiving an EM signal. This is represented by the analog PM carrier signal 916.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to the IF signal FIF. In
Portions 5410A represent energy transferred from the analog PM carrier signal 916 to a storage device, while simultaneously driving an output load. The portions 5410A occur when a switching module is closed by the energy transfer pulses 5407.
Portions 5410B represent energy stored in a storage device continuing to drive the load. Portions 5410B occur when the switching module is opened after energy transfer pulses 5407.
Because a harmonic of the aliasing rate is off-set from the frequency of the analog PM carrier signal 716, the energy transfer pulses 5407 “walk through” the analog PM carrier signal 916 at the difference frequency FDIFF. In other words, the energy transfer pulse 5407 occur at different locations of subsequent cycles of the analog PM carrier signal 916. As a result, the energy transfer pulses 5407 capture varying amounts of energy from the analog PM carrier signal 916, as illustrated by portions 5410, which provides the PM intermediate signal 5412 with an oscillating frequency FIF.
In
The present invention can output the unfiltered PM intermediate signal 5414, the filtered PM intermediate signal 5416, a partially filtered PM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the PM intermediate signals 5414 in
1.2.3.1.2 Digital PM Carrier Signal
A process for down-converting the digital PM carrier signal 1016 in
The process begins at step 4608, which includes receiving an EM signal. This is represented by the digital PM carrier signal 1016.
Step 4610 includes receiving an energy transfer signal having an aliasing rate FAR.
Step 4612 includes transferring energy from the EM signal at the aliasing rate to down-convert the EM signal to an intermediate signal FIF. In
Portions 5510A represent energy transferred from the digital PM carrier signal 1016 to a storage device, while simultaneously driving an output load. The portions 5510A occur when a switching module is closed by the energy transfer pulses 5507.
Portions 5510B represent energy stored in a storage device continuing to drive the load. Portions 5510B occur when the switching module is opened after energy transfer pulses 5507.
Because a harmonic of the aliasing rate is off-set from the frequency of the digital PM carrier signal 716, the energy transfer pulses 5507 “walk through” the digital PM carrier signal 1016 at the difference frequency FDIFF. In other words, the energy transfer pulse 5507 occur at different locations of subsequent cycles of the digital PM carrier signal 1016. As a result, the energy transfer pulses 5507 capture varying amounts of energy from the digital PM carrier signal 1016, as illustrated by portions 5510, which provides the PM intermediate signal 5512 with an oscillating frequency FIF.
In
The present invention can output the unfiltered PM intermediate signal 5514, the filtered PM intermediate signal 5516, a partially filtered PM intermediate signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The signals referred to herein illustrate frequency down-conversion in accordance with the invention. For example, the PM intermediate signals 5514 in
1.2.3.2 Structural Description
Operation of the energy transfer system 6302 is now described for the analog PM carrier signal 916, with reference to the flowchart 4607 and the timing diagrams in
Operation of the energy transfer system 6302 is now described for the digital PM carrier signal 1016, with reference to the flowchart 1401 and the timing diagrams in
Example embodiments of the energy transfer module 6304 are disclosed in Sections 4 and 5 below.
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention. Example implementations of the energy transfer module 6304 are disclosed in Sections 4 and 5 below.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. These implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
In an embodiment, the invention directly down-converts an EM signal to a baseband signal, by transferring energy from the EM signal. This embodiment is referred to herein as direct-to-data down-conversion and is illustrated by 4516 in
This embodiment can be implemented with modulated and unmodulated EM signals. This embodiment is described herein using the modulated carrier signal FMC in
The following sections describe methods for directly down-converting the modulated carrier signal FMC to the demodulated baseband signal FDMB. Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of transferring energy from the modulated carrier signal FMC to directly down-convert the modulated carrier signal FMC to the demodulated baseband signal FDMB, according to the invention. In particular, an operational process of directly down-converting the modulated carrier signal FMC to the demodulated baseband signal FDMB is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. The structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all combinations of modulation techniques are valid for this invention. For ease of discussion, the digital AM carrier signal 616 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed flowcharts and descriptions for AM and PM example embodiments. FM presents special considerations that are dealt with separately in Section III.3. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of EM signal, including any form of modulated carrier signal and unmodulated carrier signals.
The high-level process illustrated in the flowchart 4613 is now described at a high level using the digital AM carrier signal 616, from
The process of the flowchart 4613 begins at step 4614, which includes receiving an EM signal. Step 4613 is represented by the digital AM carrier signal 616.
Step 4616 includes receiving an energy transfer signal having an aliasing rate FAR.
The non-negligible apertures 5606 can be any width other than the period of the EM signal, or a multiple thereof. For example, the non-negligible apertures 5606 can be less than the period of the signal 616 such as, ⅛, ¼, ½, ¾, etc., of the period of the signal 616. Alternatively, the non-negligible apertures 5606 can be greater than the period of the signal 616. The width and amplitude of the apertures 5606 can be optimized based on one or more of a variety of criteria, as described in sections below.
The energy transfer pulses 5604 repeat at the aliasing rate or pulse repetition rate. The aliasing rate is determined in accordance with EQ. (2), reproduced below for convenience.
F
C
=n·F
AR
±F
IF EQ. (2)
When directly down-converting an EM signal to baseband (i.e., zero IF), EQ. (2) becomes:
F
C
=n·F
AR EQ. (8)
Thus, to directly down-convert the AM signal 616 to a demodulated baseband signal, the aliasing rate is substantially equal to the frequency of the AM signal 616 or to a harmonic or sub-harmonic thereof. Although the aliasing rate is too low to permit reconstruction of higher frequency components of the AM signal 616 (i.e., the carrier frequency), it is high enough to permit substantial reconstruction of the lower frequency modulating baseband signal 310.
Step 4618 includes transferring energy from the EM signal at the aliasing rate to directly down-convert the EM signal to a demodulated baseband signal FDMB.
In an embodiment, the energy transfer system 6302 transfers energy from any type of EM signal, including modulated carrier signals and unmodulated carrier signal, to directly down-convert the EM signal to a demodulated baseband signal. Preferably, the energy transfer system 6302 transfers energy from the EM signal 1304 to down-convert it to demodulated baseband signal in the manner shown in the operational flowchart 4613. However, it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 4613. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
Operation of the energy transfer system 6302 is now described in at a high level for the digital AM carrier signal 616, with reference to the flowchart 4613 and the timing diagrams illustrated in
Example implementations of the energy transfer module 6302 are disclosed in Sections 4 and 5 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting the EM signal to the demodulated baseband signal FDMB, illustrated in the flowchart 4613 of
2.2.1.1 Operational Description
Operation of the exemplary process of the flowchart 4613 in
2.2.1.1.1 Analog AM Carrier Signal
A process for directly down-converting the analog AM carrier signal 516 in
The process begins at step 4614, which includes receiving an EM signal. This is represented by the analog AM carrier signal 516.
Step 4616 includes receiving an energy transfer signal having an aliasing rate FAR. In
Step 4618 includes transferring energy from the EM signal at the aliasing rate to directly down-convert the EM signal to the demodulated baseband signal FDMB. In
The demodulated baseband signal 5712 includes portions 5710A, which correlate with the energy transfer pulses 5707 in
In
The present invention can output the unfiltered demodulated baseband signal 5712, the filtered demodulated baseband signal 5716, a partially filtered demodulated baseband signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The aliasing rate of the energy transfer signal is preferably controlled to optimize the demodulated baseband signal for amplitude output and polarity, as desired.
The drawings referred to herein illustrate direct down-conversion in accordance with the invention. For example, the demodulated baseband signals 5712 in
2.2.1.1.2 Digital AM Carrier Signal
A process for directly down-converting the digital AM carrier signal 616 in
The process begins at step 4614, which includes receiving an EM signal. This is represented by the digital AM carrier signal 616.
Step 4616 includes receiving an energy transfer signal having an aliasing rate FAR. In
Step 4618 includes transferring energy from the EM signal at the aliasing rate to directly down-convert the EM signal to the demodulated baseband signal FDMB. In
The demodulated baseband signal 5812 includes portions 5810A, which correlate with the energy transfer pulses 5807 in
In
The present invention can output the unfiltered demodulated baseband signal 5812, the filtered demodulated baseband signal 5816, a partially filtered demodulated baseband signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
The drawings referred to herein illustrate direct down-conversion in accordance with the invention. For example, the demodulated baseband signals 5812 in
2.2.1.2 Structural Description
In an embodiment, the energy transfer module 6304 preferably transfers energy from the EM signal to directly down-convert it to a demodulated baseband signal in the manner shown in the operational flowchart 4613. But it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 1413. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
Operation of the energy transfer system 6302 is now described for the digital AM carrier signal 516, with reference to the flowchart 4613 and the timing diagrams in
The operation of the energy transfer system 6402 is now described for the digital AM carrier signal 616, with reference to the flowchart 4613 and the timing diagrams in
Example implementations of the energy transfer module 6302 are disclosed in Sections 4 and 5 below.
2.2.2.1 Operational Description
Operation of the exemplary process of flowchart 4613 in
2.2.2.1.1 Analog PM Carrier Signal
A process for directly down-converting the analog PM carrier signal 916 to a demodulated baseband signal is now described for the flowchart 4613 in
The process begins at step 4614, which includes receiving an EM signal. This is represented by the analog PM carrier signal 916.
Step 4616 includes receiving an energy transfer signal having an aliasing rate FAR. In
Step 4618 includes transferring energy from the EM signal at the aliasing rate to directly down-convert the EM signal to the demodulated baseband signal FDMB. In
The demodulated baseband signal 5912 includes portions 5910A, which correlate with the energy transfer pulses 5907 in
In
The present invention can output the unfiltered demodulated baseband 5912, the filtered demodulated baseband signal 5916, a partially filtered demodulated baseband signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
The drawings referred to herein illustrate direct down-conversion in accordance with the invention. For example, the demodulated baseband signals 5912 in
2.2.2.1.2 Digital PM Carrier Signal
A process for directly down-converting the digital PM carrier signal 1016 in
Step 4616 includes receiving an energy transfer signal FAR. In
Step 4618 includes transferring energy from the EM signal at the aliasing rate to directly down-convert the EM signal to the demodulated baseband signal FDMB. In
The demodulated baseband signal 6012 includes portions 6010A, which correlate with the energy transfer pulses 6007 in
In
The present invention can output the unfiltered demodulated baseband signal 6012, the filtered demodulated baseband signal 6016, a partially filtered demodulated baseband signal, a stair step output signal, etc. The choice between these embodiments is generally a design choice that depends upon the application of the invention.
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
The drawings referred to herein illustrate direct down-conversion in accordance with the invention. For example, the demodulated baseband signals 6012 in
2.2.2.2 Structural Description
In an embodiment, the energy transfer system 6302 preferably transfers energy from an EM signal to directly down-convert it to a demodulated baseband signal in the manner shown in the operational flowchart 4613. But it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 1413. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
Operation of the energy transfer system 6302 is now described for the analog PM carrier signal 916, with reference to the flowchart 4613 and the timing diagrams in
Operation of the energy transfer system 6302 is now described for the digital PM carrier signal 1016, with reference to the flowchart 4613 and to the timing diagrams in
Example implementations of the energy transfer module 6302 are disclosed in Sections 4 and 5 below.
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention. Example implementations of the energy transfer module 6302 are disclosed in Sections 4 and 5 below.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. These implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
In an embodiment, the invention down-converts an FM carrier signal FFMC to a non-FM signal F(NON-FM), by transferring energy from the FM carrier signal FFMC at an aliasing rate. This embodiment is illustrated in
In an example embodiment, the FM carrier signal FFMC is down-converted to a phase modulated (PM) signal FPM. In another example embodiment, the FM carrier signal FFMC is down-converted to an amplitude modulated (AM) signal FAM. The down-converted signal can be demodulated with any conventional demodulation technique to obtain a demodulated baseband signal FDMB.
The invention can be implemented with any type of FM signal. Exemplary embodiments are provided below for down-converting a frequency shift keying (FSK) signal to a non-FSK signal. FSK is a sub-set of FM, wherein an FM signal shifts or switches between two or more frequencies. FSK is typically used for digital modulating baseband signals, such as the digital modulating baseband signal 310 in
In a first example embodiment, energy is transferred from the FSK signal 816 at an aliasing rate that is based on a mid-point between the upper and lower frequencies of the FSK signal 816. When the aliasing rate is based on the mid-point, the FSK signal 816 is down-converted to a phase shift keying (PSK) signal. PSK is a sub-set of phase modulation, wherein a PM signal shifts or switches between two or more phases. PSK is typically used for digital modulating baseband signals. For example, in
In a second example embodiment, energy is transferred from the FSK signal 816 at an aliasing rate that is based upon either the upper frequency or the lower frequency of the FSK signal 816. When the aliasing rate is based upon the upper frequency or the lower frequency of the FSK signal 816, the FSK signal 816 is down-converted to an amplitude shift keying (ASK) signal. ASK is a sub-set of amplitude modulation, wherein an AM signal shifts or switches between two or more amplitudes. ASK is typically used for digital modulating baseband signals. For example, in
The following sections describe methods for transferring energy from an FM carrier signal FFMC to down-convert it to the non-FM signal F(NON-FM). Exemplary structural embodiments for implementing the methods are also described. It should be understood that the invention is not limited to the particular embodiments described below. Equivalents, extensions, variations, deviations, etc., of the following will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such equivalents, extensions, variations, deviations, etc., are within the scope and spirit of the present invention.
The following sections include a high level discussion, example embodiments, and implementation examples.
This section (including its subsections) provides a high-level description of transferring energy from the FM carrier signal FFM to down-convert it to the non-FM signal F(NON-FM), according to the invention. In particular, an operational process for down-converting the FM carrier signal FFM to the non-FM signal F(NON-FM) is described at a high-level. Also, a structural implementation for implementing this process is described at a high-level. The structural implementation is described herein for illustrative purposes, and is not limiting. In particular, the process described in this section can be achieved using any number of structural implementations, one of which is described in this section. The details of such structural implementations will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein.
Any and all forms of frequency modulation techniques are valid for this invention. For ease of discussion, the digital FM carrier (FSK) signal 816 is used to illustrate a high level operational description of the invention. Subsequent sections provide detailed flowcharts and descriptions for the FSK signal 816. Upon reading the disclosure and examples therein, one skilled in the relevant art(s) will understand that the invention can be implemented to down-convert any type of FM signal.
The method illustrated in the flowchart 4619 is described below at a high level for down-converting the FSK signal 816 in
The process of the flowchart 4619 begins at step 4620, which includes receiving an FM signal. This is represented by the FSK signal 816. The FSK signal 816 shifts between a first frequency 8410 and a second frequency 8412. The first frequency 8410 can be higher or lower than the second frequency 8412. In an exemplary embodiment, the first frequency 8410 is approximately 899 MHZ and the second frequency 8412 is approximately 901 MHZ.
Step 4622 includes receiving an energy transfer signal having an aliasing rate FAR.
The energy transfer pulses 8403 repeat at the aliasing rate FAR, which is determined or selected as previously described. Generally, when down-converting an FM carrier signal FFMC to a non-FM signal F(NON-FM), the aliasing rate is substantially equal to a harmonic or, more typically, a sub-harmonic of a frequency within the FM signal. In this example overview embodiment, where the FSK signal 816 is to be down-converted to a PSK signal, the aliasing rate is substantially equal to a harmonic or, more typically, a sub-harmonic of the mid-point between the first frequency 8410 and the second frequency 8412. For the present example, the mid-point is approximately 900 MHZ.
Step 4624 includes transferring energy from the FM carrier signal FFMC at the aliasing rate to down-convert the FM carrier signal FFMC to the non-FM signal F(NON-FM).
When the second frequency 8412 is under-sampled, the PSK signal 8404 has a frequency of approximately 1 MHZ and is used as a phase reference. When the first frequency 8410 is under-sampled, the PSK signal 8404 has a frequency of 1 MHZ and is phase shifted 180 degrees from the phase reference.
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
Detailed exemplary embodiments for down-converting an FSK signal to a PSK signal and for down-converting an FSK signal to an ASK signal are provided below.
In a modulation conversion embodiment, the EM signal 1304 is an FM carrier signal FFMC and the energy transfer module 6304 transfers energy from FM carrier signal at a harmonic or, more typically, a sub-harmonic of a frequency within the FM frequency band. Preferably, the energy transfer module 6304 transfers energy from the FM carrier signal FFMC to down-convert it to a non-FM signal F(NON-FM) in the manner shown in the operational flowchart 4619. But it should be understood that the scope and spirit of the invention includes other structural embodiments for performing the steps of the flowchart 4619. The specifics of the other structural embodiments will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
The operation of the energy transfer system 6302 shall now be described with reference to the flowchart 4619 and the timing diagrams of
Example implementations of the energy transfer module 6302 are provided in Section 4 below.
Various embodiments related to the method(s) and structure(s) described above are presented in this section (and its subsections). These embodiments are described herein for purposes of illustration, and not limitation. The invention is not limited to these embodiments. Alternate embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
The method for down-converting an FM carrier signal FFMC to a non-FM signal, F(NON-FM), illustrated in the flowchart 4619 of
3.2.1.1 Operational Description
A process for down-converting the FSK signal 816 in
The FSK signal 816 is re-illustrated in
The process begins at step 4620, which includes receiving an FM signal. This is represented by the FSK signal 816.
Step 4622 includes receiving an energy transfer signal having an aliasing rate FAR.
In this example, where an FSK signal is being down-converted to a PSK signal, the aliasing rate is substantially equal to a harmonic or, more typically, a sub-harmonic, of the mid-point between the frequencies 6106 and 6108. In this example, where the first frequency 6106 is 899 MHZ and second frequency 6108 is 901 MHZ, the mid-point is approximately 900 MHZ. Suitable aliasing rates thus include 1.8 GHZ, 900 MHZ, 450 MHZ, etc.
Step 4624 includes transferring energy from the FM signal at the aliasing rate to down-convert it to the non-FM signal F(NON-FM). In
The PSK signal 6112 includes portions 6110A, which correlate with the energy transfer pulses 6107 in
In
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and polarity, as desired.
The drawings referred to herein illustrate modulation conversion in accordance with the invention. For example, the PSK signals 6112 in
3.2.1.2 Structural Description
The operation of the energy transfer system 1602 is now described for down-converting the FSK signal 816 to a PSK signal, with reference to the flowchart 4619 and to the timing diagrams of
3.2.2. Second Example Embodiment
3.2.2.1 Operational Description
A process for down-converting the FSK signal 816 in
The FSK signal 816 is re-illustrated in
The process begins at step 4620, which includes receiving an FM signal. This is represented by the FSK signal 816.
Step 4622 includes receiving an energy transfer signal having an aliasing rate FAR.
In this example, where an FSK signal is being down-converted to an ASK signal, the aliasing rate is substantially equal to a harmonic or, more typically, a sub-harmonic, of either the first frequency 6206 or the second frequency 6208. In this example, where the first frequency 6206 is 899 MHZ and the second frequency 6208 is 901 MHZ, the aliasing rate can be substantially equal to a harmonic or sub-harmonic of 899 MHZ or 901 MHZ.
Step 4624 includes transferring energy from the FM signal at the aliasing rate to down-convert it to the non-FM signal F(NON-FM). In
The ASK signal 6212 includes portions 6210A, which correlate with the energy transfer pulses 6209 in
In
The aliasing rate of the energy transfer signal is preferably controlled to optimize the down-converted signal for amplitude output and/or polarity, as desired.
The drawings referred to herein illustrate modulation conversion in accordance with the invention. For example, the ASK signals 6212 in
3.2.2.2 Structural Description
The operation of the energy transfer system 1602 is now described for down-converting the FSK signal 816 to an ASK signal, with reference to the flowchart 4619 and to the timing diagrams of
The embodiments described above are provided for purposes of illustration. These embodiments are not intended to limit the invention. Alternate embodiments, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate embodiments fall within the scope and spirit of the present invention.
Example implementations of the energy transfer module 6302 are disclosed in Sections 4 and 5 below.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in Sections 4 and 5 below. These implementations are presented for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described therein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
Exemplary operational and/or structural implementations related to the method(s), structure(s), and/or embodiments described above are presented in this section (and its subsections). These implementations are presented herein for purposes of illustration, and not limitation. The invention is not limited to the particular implementation examples described herein. Alternate implementations (including equivalents, extensions, variations, deviations, etc., of those described herein) will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
The energy transfer module 6304 transfers energy from the EM signal 1304 at the aliasing rate of the energy transfer signal 6306, as described in the sections above with respect to the flowcharts 4601 in
The gated transfer module 6404 transfers energy from the EM signal 1304 at the aliasing rate of the energy transfer signal 6306, as described in the sections above with respect to the flowcharts 4601 in
For example, operation of the switch module 6502 and the storage module 6506 is now described for down-converting the EM signal 1304 to an intermediate signal, with reference to the flowchart 4607 and the example timing diagrams in
In step 4608, the switch module 6502 receives the EM signal 1304 (
In
Prior to time t0, the normally open switch 6704 and the normally closed switch 6706 are at their normal states.
At time t0, the isolation signal 6712 in
At time t1, the energy transfer signal 6306 in
Prior to t2, the energy transfer signal 6306 in
At time t2, the isolation signal 6712 in
The storage module 6716, is similar to the storage module 6506
The switch module 6502 in
In an embodiment, the switch module 6610 can be implemented as a transistor, such as, for example, a field effect transistor (FET), a bi-polar transistor, or any other suitable circuit switching device.
In
It should be understood that the illustration of the switch module 6610 as a FET 6602 in
In
In
The storage modules 6506 and 6716 store non-negligible amounts of energy from the EM signal 1304. In an exemplary embodiment, the storage modules 6506 and 6716 are implemented as a reactive storage module 6801 in
In an embodiment, the storage modules 6506 and 6716 include one or more capacitive storage elements, illustrated in
The goal of the storage modules 6506 and 6716 is to store non-negligible amounts of energy transferred from the EM signal 1304. Amplitude reproduction of the original, unaffected EM input signal is not necessarily important. In an energy transfer environment, the storage module preferably has the capacity to handle the power being transferred, and to allow it to accept a non-negligible amount of power during a non-negligible aperture period.
A terminal 6806 serves as an output of the capacitive storage module 6802. The capacitive storage module 6802 provides the stored energy at the terminal 6806.
In an alternative embodiment, the storage modules 6506 and 6716 include one or more inductive storage elements, illustrated in
In an alternative embodiment, the storage modules 6506 and 6716 include a combination of one or more capacitive storage elements and one or more inductive storage elements, illustrated in
In an embodiment, the optional energy transfer signal module 6902 includes an aperture generator, an example of which is illustrated in
The width or aperture of the pulses 6826 is determined by delay through the branch 6822 of the aperture generator 6820. Generally, as the desired pulse width increases, the difficulty in meeting the requirements of the aperture generator 6820 decrease. In other words, to generate non-negligible aperture pulses for a given EM input frequency, the components utilized in the example aperture generator 6820 do not require as fast reaction times as those that are required in an under-sampling system operating with the same EM input frequency.
The example logic and implementation shown in the aperture generator 6820 are provided for illustrative purposes only, and are not limiting. The actual logic employed can take many forms. The example aperture generator 6820 includes an optional inverter 6828, which is shown for polarity consistency with other examples provided herein.
An example implementation of the aperture generator 6820 is illustrated in
In an embodiment, the input signal 6824 is generated externally of the energy transfer signal module 6902, as illustrated in
The type of down-conversion performed by the energy transfer system 6901 depends upon the aliasing rate of the energy transfer signal 6306, which is determined by the frequency of the pulses 6826. The frequency of the pulses 6826 is determined by the frequency of the input signal 6824. For example, when the frequency of the input signal 6824 is substantially equal to a harmonic or a sub-harmonic of the EM signal 1304, the EM signal 1304 is directly down-converted to baseband (e.g. when the EM signal is an AM signal or a PM signal), or converted from FM to a non-FM signal. When the frequency of the input signal 6824 is substantially equal to a harmonic or a sub-harmonic of a difference frequency, the EM signal 1304 is down-converted to an intermediate signal.
The optional energy transfer signal module 6902 can be implemented in hardware, software, firmware, or any combination thereof.
4.2 The Energy Transfer System as an Inverted Gated Transfer System
The switch module 7404 can be implemented as described above with reference to
In the illustrated embodiment, the storage module 7206 includes one or more capacitors 7408. The capacitor(s) 7408 are selected to pass higher frequency components of the EM signal 1304 through to a terminal 7410, regardless of the state of the switch module 7404. The capacitor 7408 stores non-negligible amounts of energy from the EM signal 1304. Thereafter, the signal at the terminal 7410 is off-set by an amount related to the energy stored in the capacitor 7408.
Operation of the inverted gated transfer system 7401 is illustrated in
The inverted gated transfer system 7401 can be used to down-convert any type of EM signal, including modulated carrier signals and unmodulated carrier signals.
FET 11006 receives the EM signal 11002 and aliasing signal 11014. In one embodiment, aliasing signal 11014 includes a train of pulses having non-negligible apertures that repeat at an aliasing rate. The aliasing rate may be harmonic or sub-harmonic of the EM signal 11002. FET 11006 samples EM signal 11002 at the aliasing rate of aliasing signal 11014 to generate down-converted signal 11012. In one embodiment, aliasing signal 11014 controls the gate of FET 11006 so that FET 11006 conducts (or turns on) when the FET gate-to-source voltage (VGS) exceeds a threshold voltage (VT). When the FET 11006 conducts, a channel is created from source to drain of FET 11006 so that charge is transferred from the EM signal 11002 to the capacitor 11010. More specifically, the FET 11006 conductance (1/R) vs VGS is a continuous function that reaches an acceptable level at VT, as illustrated in
As stated above, n-channel FET 11006 conducts when VGS exceeds the threshold voltage VT. As shown in
For example,
As stated earlier, the conductance of FET 11006 vs VGS is mathematically continuous and is not a hard cutoff. In other words, FET 11006 will marginally conduct when controlled by pulse 11110, even though pulse 11110 is below VT 11112. However, the insertion loss of FET 11006 will be increased when compared with a VGS pulse 11111, which is greater than VT 11112. The performance reduction caused by a large amplitude input signal is often referred to as clipping or compression. Clipping causes distortion in the down-converted signal 11012, which adversely affects the faithful down-conversion of input EM signal 11102. Dynamic range is a figure of merit associated with the range of input signals that can be faithfully down-converted without introducing distortion in the down-converted signal. The higher the dynamic range of a down-conversion circuit, the larger the input signals that can down-converted without introducing distortion in the down-converted signal.
As stated, aliasing module 11200 operates two complementary FETs to extend the dynamic range and reduce any distortion effects. This requires that two complementary aliasing signals 11224, 11226 be generated from aliasing signal 11220 to control the sampling by FETs 11218, 11204, respectively. To do so, inverter 11222 receives and inverts aliasing signal 11220 to generate aliasing signal 11224 that controls p-channel FET 11218. Delay 11202 delays aliasing signal 11220 to generate aliasing signal 11226, where the amount of time delay is approximately equivalent to that associated with inverter 11222. As such, aliasing signals 11224 and 11226 are approximately complementary in amplitude.
Node 11210 receives EM signal 11208, and couples EM signals 11227, 11228 to the sources of n-channel FET 11204 and p-channel FET 11218, respectively, where EM signals 11227, 11228 are substantially replicas of EM signal 11208. N-channel FET 11204 samples EM signal 11227 as controlled by aliasing signal 11226, and produces samples 11236 at the drain of FET 11204. Likewise, p-channel FET 11218 samples EM signal 11228 as controlled by aliasing signal 11224, and produces samples 11238 at the drain of FET 11218. Node 11212 combines the resulting charge samples into charge samples 11240, which are stored by capacitor 11230. The charge stored by capacitor 11230 during successive samples forms down-converted signal 11214. Aliasing module 11200 offers improved dynamic range over aliasing module 11000 because re-channel FET 11204 and p-channel FET 11214 are complementary devices. Therefore, if one device is cutoff because of a large input EM signal 11208, the other device will conduct and sample the input signal, as long as the input signal is between the power supply voltages V+ 11232 and V− 11234. This is often referred to as rail-to-rail operation as will be understood by those skilled in the arts.
For example,
As stated, n-channel FET 11204 conducts when VGS 11308 exceeds VT 11309, and p-channel FET 11218 conducts when VGS 11316 drops below VT 11317. As illustrated by
As illustrated above, aliasing module 11200 offers an improvement in dynamic range over aliasing module 11000 because of the complimentary FET structure. Any input signal that is within the power supply voltages V+ 11232 and V− 11234 will cause either FET 11204 or FET 11218 to conduct, or cause both FETs to conduct, as is demonstrated by
As stated, an aliasing module with a complementary FET structure offers improved dynamic range when compared with a single (or unipolar) FET configuration. This is further illustrated by comparing the signal waveforms associated aliasing module 11602 (of
Aliasing module 11602 (
EM signals 11608, 11704 are relatively large input signals that approach the power supply voltages of ±1.65 volts, as is shown in
Similarly in
In summary, down-converted signal 11706 reflects distortion introduced by a relatively large EM signal that is pinching-off the single FET 11712 in aliasing module 11702. Down-converted signal 11610 that is produced by aliasing module 11602 is relatively distortion free. This occurs because the complementary FET configuration in aliasing module 11602 is able to handle input signals with large amplitudes without introducing distortion in the down-converted signal 11610. Therefore, the complementary FET configuration in the aliasing module 11602 offers improved dynamic range when compared with the single FET configuration of the aliasing module 11702.
I-channel flip-flop 12412 inputs inverted LO signal 12418. Q-channel flip-flop 12414 inputs non-inverted LO signal 12420. In the current embodiment, I-channel flip-flop 12412 and Q-channel flip-flop 12414 are edge-triggered flip-flops. When either flip-flop receives a rising edge on its input, the flip-flop output changes state. Hence, I-channel flip-flop 12412 and Q-channel flip-flop 12414 each output signals that are approximately half of the input signal frequency. Additionally, as would be recognized by persons skilled in the relevant art(s), because the inputs to I-channel flip-flop 12412 and Q-channel flip-flop 12414 are approximately 180° out of phase, their resulting outputs are signals that are approximately 90° out of phase. I-channel flip-flop 12412 outputs I-channel oscillating signal 12422, as shown in
It should be understood that the illustration of the splitter circuit 12400 in
In an embodiment, the switch modules discussed herein can be implemented as a series of switches operating in parallel as a single switch. The series of switches can be transistors, such as, for example, field effect transistors (FET), bi-polar transistors, or any other suitable circuit switching devices. The series of switches can be comprised of one type of switching device, or a combination of different switching devices.
For example,
In an embodiment, FETs 12502a-n have similar characteristics. In another embodiment, one or more of FETs 12502a-n have different characteristics than the other FETs. For example, FETs 12502a-n may be of different sizes. In CMOS, generally, the larger size a switch is (meaning the larger the area under the gate between the source and drain regions), the longer it takes for the switch to turn on. The longer turn on time is due in part to a higher gate to channel capacitance that exists in larger switches. Smaller CMOS switches turn on in less time, but have a higher channel resistance. Larger CMOS switches have lower channel resistance relative to smaller CMOS switches. Different turn on characteristics for different size switches provides flexibility in designing an overall switch module structure. By combining smaller switches with larger switches, the channel conductance of the overall switch structure can be tailored to satisfy given requirements.
In an embodiment, FETs 12502a-n are CMOS switches of different relative sizes. For example, FET 12502a may be a switch with a smaller size relative to FETs 12502b-n. FET 12502b may be a switch with a larger size relative to FET 12502a, but smaller size relative to FETs 12502c-n. The sizes of FETs 12502c-n also may be varied relative to each other. For instance, progressively larger switch sizes may be used. By varying the sizes of FETs 12502a-n relative to each other, the turn on characteristic curve of the switch module can be correspondingly varied. For instance, the turn on characteristic of the switch module can be tailored such that it more closely approaches that of an ideal switch. Alternately, the switch module could be tailored to produce a shaped conductive curve.
By configuring FETs 12502a-n such that one or more of them are of a relatively smaller size, their faster turn on characteristic can improve the overall switch module turn on characteristic curve. Because smaller switches have a lower gate to channel capacitance, they can turn on more rapidly than larger switches.
By configuring FETs 12502a-n such that one or more of them are of a relatively larger size, their lower channel resistance also can improve the overall switch module turn on characteristics. Because larger switches have a lower channel resistance, they can provide the overall switch structure with a lower channel resistance, even when combined with smaller switches. This improves the overall switch structure's ability to drive a wider range of loads. Accordingly, the ability to tailor switch sizes relative to each other in the overall switch structure allows for overall switch structure operation to more nearly approach ideal, or to achieve application specific requirements, or to balance trade-offs to achieve specific goals, as will be understood by persons skilled in the relevant arts(s) from the teachings herein.
It should be understood that the illustration of the switch module as a series of FETs 12502a-n in
Circuit performance also can be improved by reducing overall switch area. As discussed above, smaller switches (i.e., smaller area under the gate between the source and drain regions) have a lower gate to channel capacitance relative to larger switches. The lower gate to channel capacitance allows for lower circuit sensitivity to noise spikes.
It should be understood that the illustration of the switches above as FETs in
In embodiments wherein the switch modules discussed herein are comprised of a series of switches in parallel, in some instances it may be desirable to minimize the effects of charge injection. Minimizing charge injection is generally desirable in order to reduce the unwanted circuit radiation resulting therefrom. In an embodiment, unwanted charge injection effects can be reduced through the use of complementary n-channel MOSFETs and p-channel MOSFETs. N-channel MOSFETs and p-channel MOSFETs both suffer from charge injection. However, because signals of opposite polarity are applied to their respective gates to turn the switches on and off, the resulting charge injection is of opposite polarity. Resultingly, n-channel MOSFETs and p-channel MOSFETs may be paired to cancel their corresponding charge injection. Hence, in an embodiment, the switch module may be comprised of n-channel MOSFETs and p-channel MOSFETS, wherein the members of each are sized to minimize the undesired effects of charge injection.
It should be understood that the use of FETs in
The processes involved in fabricating semiconductor circuits, such as MOSFETs, have limitations. In some instances, these process limitations may lead to circuits that do not function as ideally as desired. For instance, a non-ideally fabricated MOSFET may suffer from parasitic capacitances, which in some cases may cause the surrounding circuit to radiate noise. By fabricating circuits with structure layouts as close to ideal as possible, problems of non-ideal circuit operation can be minimized.
Operation of MOSFET 12800 shall now be described. When a positive voltage is applied to gate 12802, electrons in the p-type material of channel region 12804 are attracted to the surface below insulator 12814, forming a connecting near-surface region of n-type material between the source and the drain, called a channel. The larger or more positive the voltage between the gate contact 12806 and source region 12808, the lower the resistance across the region between.
In
As shown in
It should be understood that the illustration of the n-channel enhancement-mode MOSFET is for example purposes only. The present invention is applicable to depletion mode MOSFETs, and other transistor types, as will be apparent to persons skilled in the relevant art(s) based on the discussion contained herein.
The implementations described above are provided for purposes of illustration. These implementations are not intended to limit the invention. Alternate implementations, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
The methods and systems described in sections above can be optimized with one or more of the optimization methods or systems described below.
In an embodiment, the optional energy transfer signal module 6902 in
In the example of
F
AR=2·Fosc EQ. (9)
The aperture width of the aliasing pulses is determined by the delay through a first inverter 7108 of
The invention can be implemented in a variety of differential configurations. Differential configurations are useful for reducing common mode noise. This can be very useful in receiver systems where common mode interference can be caused by intentional or unintentional radiators such as cellular phones, CB radios, electrical appliances etc. Differential configurations are also useful in reducing any common mode noise due to charge injection of the switch in the switch module or due to the design and layout of the system in which the invention is used. Any spurious signal that is induced in equal magnitude and equal phase in both input leads of the invention will be substantially reduced or eliminated. Some differential configurations, including some of the configurations below, are also useful for increasing the voltage and/or for increasing the power of the down-converted signal 1308B.
Differential systems are most effective when used with a differential front end (inputs) and a differential back end (outputs). They can also be utilized in the following configurations, for example:
a) A single-input front end and a differential back end; and
b) A differential front end and a single-output back end.
Examples of these system are provided below, with a first example illustrating a specific method by which energy is transferred from the input to the output differentially.
While an example of a differential energy transfer module is shown below, the example is shown for the purpose of illustration, not limitation. Alternate embodiments (including equivalents, extensions, variations, deviations etc.) of the embodiment described herein will be apparent to those skilled in the relevant art based on the teachings contained herein. The invention is intended and adapted to include such alternate embodiments.
One or both of the inputs 7604 and 7606 are coupled to an EM signal source. For example, the inputs can be coupled to an EM signal source, wherein the input voltages at the inputs 7604 and 7606 are substantially equal in amplitude but 180 degrees out of phase with one another. Alternatively, where dual inputs are unavailable, one of the inputs 7604 and 7606 can be coupled to ground.
In operation, when the switch module 7616 is closed, the storage modules 7614 and 7620 are in series and, provided they have similar capacitive values, accumulate charge of equal magnitude but opposite polarities. When the switch module 7616 is open, the voltage at the output 7608 is relative to the input 7604, and the voltage at the output 7610 is relative to the voltage at the input 7606.
Portions of the signals at the outputs 7608 and 7610 include signals resulting from energy stored in the storage modules 7614 and 7620, respectively, when the switch module 7616 was closed. The portions of the signals at the outputs 7608 and 7610 resulting from the stored charge are generally equal in amplitude to one another but 180 degrees out of phase.
Portions of the signals at the outputs 7608 and 7610 also include ripple voltage or noise resulting from the switching action of the switch module 7616. But because the switch module is positioned between the two outputs 7608 and 7610, the noise introduced by the switch module appears at the outputs as substantially equal and in-phase with one another. As a result, the ripple voltage can be substantially canceled out by inverting the signal at one of the outputs 7608 or 7610 and adding it to the other remaining output. Additionally, any noise that is impressed with equal amplitude and equal phase onto the input terminals 7604 and 7606 by any other noise sources will tend to be canceled in the same way.
5.2.1.1 Differential Input-to-Differential Output
5.2.1.2 Single Input-to-Differential Output
The outputs 7608 and 7610 are coupled to a differential circuit 7644 such as a filter, which preferably inverts one of the outputs 7608 or 7610 and adds it to the other output 7608 or 7610. This substantially cancels common mode noise generated by the switch module 7616. The differential circuit 7644 preferably filters the higher frequency components of the EM signal 1304 that pass through the storage modules 7614 and 7620. The resultant filtered signal is output as the down-converted signal 1308B.
5.2.1.3 Differential Input-to-Single Output
In specific alternative embodiments, the present invention is implemented using a plurality of gated transfer modules controlled by a common energy transfer signal with a storage module coupled between the outputs of the plurality of gated transfer modules. For example,
As with the first implementation described above in section 5.5.1 and its sub-sections, the gated transfer differential system 9902 can be implemented with a single input, differential inputs, a single output, differential outputs, and combinations thereof. For example,
Where common-mode rejection is desired to protect the input from various common-mode effects, and where common mode rejection to protect the output is not necessary, a differential input-to-single output implementation can be utilized.
Typically, in a balanced-to-unbalanced system, where a single output is taken from a differential system without the use of a balun, (i.e., where one of the output signals is grounded), a loss of about 6 db is observed. In the configuration of
Gated transfer systems and inverted gated transfer systems can be implemented with any of the various optimizations and configurations disclosed through the specification, such as, for example, impedance matching, tanks and resonant structures, bypass networks, etc. For example, the differential system 10002 in
The down-converted signal 1308B may be smoothed by filtering as desired. The differential circuit 7644 implemented as a filter in
The energy transfer module has input and output impedances generally defined by (1) the duty cycle of the switch module, and (2) the impedance of the storage module, at the frequencies of interest (e.g. at the EM input, and intermediate/baseband frequencies).
Starting with an aperture width of approximately ½ the period of the EM signal being down-converted as a preferred embodiment, this aperture width (e.g. the “closed time”) can be decreased. As the aperture width is decreased, the characteristic impedance at the input and the output of the energy transfer module increases. Alternatively, as the aperture width increases from ½ the period of the EM signal being down-converted, the impedance of the energy transfer module decreases.
One of the steps in determining the characteristic input impedance of the energy transfer module could be to measure its value. In an embodiment, the energy transfer module's characteristic input impedance is 300 ohms. An impedance matching circuit can be utilized to efficiently couple an input EM signal that has a source impedance of, for example, 50 ohms, with the energy transfer module's impedance of, for example, 300 ohms. Matching these impedances can be accomplished in various manners, including providing the necessary impedance directly or the use of an impedance match circuit as described below.
Referring to
The output characteristic impedance can be impedance matched to take into consideration the desired output frequencies. One of the steps in determining the characteristic output impedance of the energy transfer module could be to measure its value. Balancing the very low impedance of the storage module at the input EM frequency, the storage module should have an impedance at the desired output frequencies that is preferably greater than or equal to the load that is intended to be driven (for example, in an embodiment, storage module impedance at a desired 1 MHz output frequency is 2K ohm and the desired load to be driven is 50 ohms). An additional benefit of impedance matching is that filtering of unwanted signals can also be accomplished with the same components.
In an embodiment, the energy transfer module's characteristic output impedance is 2K ohms. An impedance matching circuit can be utilized to efficiently couple the down-converted signal with an output impedance of, for example, 2K ohms, to a load of, for example, 50 ohms. Matching these impedances can be accomplished in various manners, including providing the necessary load impedance directly or the use of an impedance match circuit as described below.
When matching from a high impedance to a low impedance, a capacitor 7314 and an inductor 7316 can be configured as shown in
The configuration of the input impedance match module 7006 and the output impedance match module 7008 are considered to be initial starting points for impedance matching, in accordance with the present invention. In some situations, the initial designs may be suitable without further optimization. In other situations, the initial designs can be optimized in accordance with other various design criteria and considerations.
As other optional optimizing structures and/or components are utilized, their affect on the characteristic impedance of the energy transfer module should be taken into account in the match along with their own original criteria.
Resonant tank and other resonant structures can be used to further optimize the energy transfer characteristics of the invention. For example, resonant structures, resonant about the input frequency, can be used to store energy from the input signal when the switch is open, a period during which one may conclude that the architecture would otherwise be limited in its maximum possible efficiency. Resonant tank and other resonant structures can include, but are not limited to, surface acoustic wave (SAW) filters, dielectric resonators, diplexers, capacitors, inductors, etc.
An example embodiment is shown in
As is apparent to one skilled in the relevant art(s), parallel tank circuits provide:
In the illustrated example of
An energy transfer signal 9442 controls a switch 9414. When the energy transfer signal 9442 controls the switch 9414 to open and close, high frequency signal components are not allowed to pass through tank1 or tank2 However, the lower signal components (50 Mhz in this embodiment) generated by the system are allowed to pass through tank1 and tank2 with little attenuation. The effect of tank1 and tank2 is to further separate the input and output signals from the same node thereby producing a more stable input and output impedance. Capacitors 9418 and 9440 act to store the 50 Mhz output signal energy between energy transfer pulses.
Further energy transfer optimization is provided by placing an inductor 9410 in series with a storage capacitor 9412 as shown. In the illustrated example, the series resonant frequency of this circuit arrangement is approximately 1 GHz. This circuit increases the energy transfer characteristic of the system. The ratio of the impedance of inductor 9410 and the impedance of the storage capacitor 9412 is preferably kept relatively small so that the majority of the energy available will be transferred to storage capacitor 9412 during operation. Exemplary output signals A and B are illustrated in
In
The example tank and resonant structures described above are for illustrative purposes and are not limiting. Alternate configurations can be utilized. The various resonant tanks and structures discussed can be combined or utilized independently as is now apparent.
Concepts of charge transfer are now described with reference to
In
Where the voltage V is represented by Equation 11, Equation 10 can be rewritten as Equation 12. The change in charge Δq over time t is illustrated as in Equation 13 as Δq(t), which can be rewritten as Equation 14. Using the sum-to-product trigonometric identity of Equation 15, Equation 14 can be rewritten as Equation 16, which can be rewritten as equation 17.
Note that the sin term in Equation 11 is a function of the aperture T only. Thus, Δq(t) is at a maximum when T is equal to an odd multiple of π (i.e., π, 3π, 5π, . . . ). Therefore, the capacitor 10906 experiences the greatest change in charge when the aperture T has a value of it or a time interval representative of 180 degrees of the input sinusoid. Conversely, when T is equal to 2π, 4π, 6π, . . . , minimal charge is transferred.
Equations 18, 19, and 20 solve for q(t) by integrating Equation 10, allowing the charge on the capacitor 10906 with respect to time to be graphed on the same axis as the input sinusoid sin(t), as illustrated in the graph of
Power/charge relationships are illustrated in Equations 21-26 of
Concepts of insertion loss are illustrated in
From the above discussion, it is observed that as the aperture T increases, more charge is transferred from the input to the capacitor 10906, which increases power transfer from the input to the output. It has been observed that it is not necessary to accurately reproduce the input voltage at the output because relative modulated amplitude and phase information is retained in the transferred power.
In an embodiment of the invention, the energy transfer signal 6306 of
In
An example method of altering the energy transfer signal 6306 of
It can be shown that by varying the delay of the signal propagated by the inverter 7108, the width of the pulses in the doubler output signal 7104 can be varied. Increasing the delay of the signal propagated by inverter 7108, increases the width of the pulses. The signal propagated by inverter 7108 can be delayed by introducing a R/C low pass network in the output of inverter 7108. Other means of altering the delay of the signal propagated by inverter 7108 will be well known to those skilled in the art.
In an embodiment, the aperture width/duration is adjusted in real time. For example, referring to the timing diagrams in
In an alternative implementation, the inverted clock signal 9822 is delayed relative to the original clock signal 9814, and then ANDed with the original clock signal 9814. Alternatively, the original clock signal 9814 is delayed then inverted, and the result ANDed with the original clock signal 9814.
Operation of the real time aperture control circuit is described with reference to the timing diagrams of
The delayed clock signal 9824 is optionally amplified by the optional amplifier 9828, before being presented to the AND gate 9808. Amplification is desired, for example, where the RC constant of the RC circuit 9804 attenuates the signal below the threshold of the AND gate 9808.
The AND gate 9808 ANDs the delayed clock signal 9824, the inverted clock signal 9822, and the optional Enable signal 9810, to generate the energy transfer signal 9816. The apertures 9820 are adjusted in real time by varying the voltage to the voltage variable capacitor 9812.
In an embodiment, the apertures 9820 are controlled to optimize power transfer. For example, in an embodiment, the apertures 9820 are controlled to maximize power transfer. Alternatively, the apertures 9820 are controlled for variable gain control (e.g. automatic gain control—AGC). In this embodiment, power transfer is reduced by reducing the apertures 9820.
As can now be readily seen from this disclosure, many of the aperture circuits presented, and others, can be modified in the manner described above (e.g. circuits in
In an embodiment of the invention, a bypass network is added to improve the efficiency of the energy transfer module. Such a bypass network can be viewed as a means of synthetic aperture widening. Components for a bypass network are selected so that the bypass network appears substantially lower impedance to transients of the switch module (i.e., frequencies greater than the received EM signal) and appears as a moderate to high impedance to the input EM signal (e.g., greater that 100 Ohms at the RF frequency).
The time that the input signal is now connected to the opposite side of the switch module is lengthened due to the shaping caused by this network, which in simple realizations may be a capacitor or series resonant inductor-capacitor. A network that is series resonant above the input frequency would be a typical implementation. This shaping improves the conversion efficiency of an input signal that would otherwise, if one considered the aperture of the energy transfer signal only, be relatively low in frequency to be optimal.
For example, referring to
The following discussion will demonstrate the effects of a minimized aperture and the benefit provided by a bypassing network. Beginning with an initial circuit having a 550 ps aperture in
Generally, the amplitude of the down-converted signal 1308B varies as a function of the frequency and phase differences between the EM signal 1304 and the energy transfer signal 6306. In an embodiment, the down-converted signal 1308B is used as the feedback 6906 to control the frequency and phase relationship between the EM signal 1304 and the energy transfer signal 6306. This can be accomplished using the example logic in
In the example of
The DAC 8506 controls an input to a voltage controlled oscillator, VCO 8508. VCO 8508 controls a frequency input of a pulse generator 8510, which, in an embodiment, is substantially similar to the pulse generator shown in
In an embodiment, the state machine 8504 operates in accordance with a state machine flowchart 8519 in
The amplitude of the down-converted signal 1308B can be made to vary with the amplitude of the energy transfer signal 6306. In an embodiment where the switch module 6502 is a FET as shown in
The implementations described above are provided for purposes of illustration. These implementations are not intended to limit the invention. Alternate implementations, differing slightly or substantially from those described herein, will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Such alternate implementations fall within the scope and spirit of the present invention.
Example implementations are described below for illustrative purposes. The invention is not limited to these examples.
As described and illustrated in the preceding sections and sub-sections, embodiments of the present invention down-convert an electromagnetic signal by repeatedly transferring energy from portions of the electromagnetic signal. This section describes the operation of the present invention mathematically using matched filter theory, sampling theory, and frequency domain techniques. The concepts and principles of these theories are used to describe the present invention's waveform processing and would be known to persons skilled in the relevant arts.
As will be apparent to persons skilled in the relevant arts based on the teachings contained herein, the description of the present invention contained herein is a unique and specific application of matched filter theory, sampling theory, and frequency domain techniques. It is not taught or suggested in the present literature. Therefore, a new transform has been developed, based on matched filter theory, sampling theory, and frequency domain techniques, to describe the present invention. This new transform is referred to as the UFT transform, and it is described in Section 8, below.
It is noted that the following describes embodiments of the invention, and it is provided for illustrative purposes. The invention is not limited to the descriptions and embodiments described below. It is also noted that characterizations such as “optimal,” “sub-optimal,” “maximum,” “minimum,” “ideal,” “non-ideal,” and the like, contained herein, denote relative relationships.
Embodiments of the present invention down-convert an electromagnetic signal by repeatedly performing a matched filtering or correlating operation on a received carrier signal. Embodiments of the invention operate on or near approximate half cycles (e.g., ½, 1½, 2½, etc.) of the received signal. The results of each matched filtering/correlating process are accumulated, for example using a capacitive storage device, and used to form a down-converted version of the electromagnetic signal. In accordance with embodiments of the invention, the matched filtering/correlating process can be performed at a sub-harmonic or fundamental rate.
Operating on an electromagnetic signal with a matched filtering/correlating process or processor produces enhanced (and in some cases the best possible) signal-to-noise ration (SNR) for the processed waveform. A matched filtering/correlating process also preserves the energy of the electromagnetic signal and transfers it through the processor.
Since it is not always practical to design a matched filtering/correlating processor with passive networks, the sub-sections that follow also describe how to implement the present invention using a finite time integrating operation and an RC processing operation. These embodiments of the present invention are very practical and can be implemented using existing technologies, for example but not limited to CMOS technology.
In order to understand how embodiments of the present invention operate, it is useful to keep in mind the fact that such embodiments do not operate by trying to emulate an ideal impulse sampler. Rather, the present invention operates by accumulating the energy of a carrier signal and using the accumulated energy to produce the same or substantially the same result that would be obtained by an ideal impulse sampler, if such a device could be built. Stated more simply, embodiments of the present invention recursively determine a voltage or current value for approximate half cycles (e.g., ½, 1½, 2½, etc.) of a carrier signal, typically at a sub-harmonic rate, and use the determined voltage or current values to form a down-converted version of an electromagnetic signal. The quality of the down-converted electromagnetic signal is a function of how efficiently the various embodiments of the present invention are able to accumulate the energy of the approximate half cycles of the carrier signal.
Ideally, some embodiments of the present invention accumulate all of the available energy contained in each approximate half cycle of the carrier signal operated upon. This embodiment is generally referred to herein as a matched filtering/correlating process or processor. As described in detail below, a matched filtering/correlating processor is able to transfer substantially all of the energy contained in a half cycle of the carrier signal through the processor for use in determining, for example, a peak or an average voltage value of the carrier signal. This embodiment of the present invention produces enhanced (and in some cases the best possible) signal-to-noise ration (SNR), as described in the sub-sections below.
In step 14810, a matched filtering/correlating operation is performed on a portion of a carrier signal. For example, a match filtering/correlating operation can be performed on a 900 MHz RF signal, which typically comprises a 900 MHz sinusoid having noise signals and information signals superimposed on it. Many different types of signals can be operated upon in step 14810, however, and the invention is not limited to operating on a 900 MHz RF signal. In embodiments, Method 14800 operates on approximate half cycles of the carrier signal.
In an embodiment of the invention, step 14810 comprises the step of convolving an approximate half cycle of the carrier signal with a representation of itself in order to efficiently acquire the energy of the approximate half cycle of the carrier signal. As described elsewhere herein, other embodiments use other means for efficiently acquiring the energy of the approximate half cycle of the carrier signal. The matched filtering/correlating operation can be performed on any approximate half cycle of the carrier signal (although the invention is not limited to this), as described in detail in the sub-sections below.
In step 14820, the result of the matched filtering/correlating operation in step 14810 is accumulated, preferably in an energy storage device. In an embodiment of the present invention, a capacitive storage devise is used to store a portion of the energy of an approximate half cycle of the carrier signal.
Steps 14810 and 14820 are repeated for additional half cycles of the carrier signal. In an embodiment of the present invention, steps 14810 and 14820 are normally performed at a sub-harmonic rate of the carrier signal, for example at a third sub-harmonic rate. In another embodiment, steps 14810 and 14820 are repeated at an off-set of a sub-harmonic rate of the carrier signal.
In step 14830, a down-converted signal is output. In embodiments, the results of steps 14810 and 14820 are passed on to a reconstruction filter or an interpolation filter.
System 14900 can be thought of as a convolution processor. System 14900 multiplies the modulated carrier signal, Si(t), by a representation of itself, Si(t−T), using multiplication model 14902. The output of multiplication module 14902 is then gated by switching module 14904 to integrating module 14906. As can be seen in
As will be apparent to persons skilled in the relevant arts given the discussion herein, the present invention is not a traditional realization of a matched filter/correlator.
As described herein, in some embodiments, a matched filter/correlator embodiment according to the present invention provides maximum energy transfer and maximum SNR. A matched filter/correlator embodiment, however, might not always provide an optimum solution for all applications. For example, a matched filter/correlator embodiment might be too expensive or too complicated to implement for some applications. In such instances, other embodiments according to the present invention may provide acceptable results at a substantially lower cost, using less complex circuitry. The invention is directed to those embodiments as well.
As described herein in subsequent sub-sections, a gated matched filter/correlator processor can be approximated by a processor whose impulse response is a step function having a duration substantially equal to the time interval defined for the waveform, typically a half cycle of the electromagnetic signal, and an integrator. Such an approximation of a gated matched filter/correlator is generally referred to as a finite time integrator. A finite time integrator in accordance with an embodiment of the present invention can be implemented with, for example, a switching device controlled by a train of pulses having apertures substantially equal to the time interval defined for the waveform. The energy transfer and SNR of a finite time integrator implemented in accordance with an embodiment of the present invention is nearly that of a gated matched filter/correlator, but without having to tailor the matched filter/correlator for a particular type of electromagnetic signal. As described in sub-section 6, a finite time integrator embodiment according to the present invention can provide a SNR result that differs from the result of matched filter/correlator embodiment by only 0.91 dB.
In step 15010, a matched filtering/correlating operation is performed on a portion of a carrier signal. For example, a match filtering/correlating operation can be performed on a 900 MHz RF signal, which typically comprises a 900 MHz sinusoid having noise signals and information signals superimposed on it. Many different types of signals can be operated upon in step 15010, however, and the invention is not limited to operating on a 900 MHz RF signal. In embodiments, Method 15000 operates on approximate half cycles of the carrier signal.
In an embodiment of the invention, step 15010 comprises the step of convolving an approximate half cycle of the carrier signal with a representation of itself in order to efficiently acquire the energy of the approximate half cycle of the carrier signal. As described elsewhere herein, other embodiments use other means for efficiently acquiring the energy of the approximate half cycle of the carrier signal. The matched filtering/correlating operation can be performed on any approximate half cycle of the carrier signal (although the invention is not limited to this), as described in detail in the sub-sections below.
In step 15020, the result of the matched filtering/correlating operation in step 15010 is accumulated, preferably in an energy storage device. In an embodiment of the present invention, a capacitive storage devise is used to store a portion of the energy of an approximate half cycle of the carrier signal.
Steps 15010 and 15020 are repeated for additional half cycles of the carrier signal. In one embodiment of the present invention, steps 15010 and 15020 are performed at a sub-harmonic rate of the carrier signal. In another embodiment, steps 15010 and 15020 are repeated at an off-set of a sub-harmonic rate of the carrier signal.
In step 15030, a down-converted signal is output. In embodiments, the results of steps 15010 and 15020 are passed on to a reconstruction filter or an interpolation filter.
Switching module 15102 is controlled by a windowing function, u(t)−u(t−TA). The length of the windowing function aperture is TA, which is equal to an approximate half cycle of the received carrier signal, Si(t). Switching module 15102 ensures that approximate half cycles of the carrier signal can be operated upon at a sub-harmonic rate. In an embodiment of system 15100, the received carrier signal is operated on at an off-set of a sub-harmonic rate of the carrier signal.
Integration module 15104 integrates the output of switching module 15102 and passes on its result, S0(t). This embodiment of the present invention is described in more detail in sub-section 4 below.
The prior sub-section describes how a gated matched filter/correlator can be approximated with a finite time integrator. This sub-section describes how the integrator portion of the finite time integrator can be approximated with a resistor/capacitor (RC) processor. This embodiment of the present invention is generally referred to herein as an RC processor, and it can be very inexpensive to implement. Additionally, the RC processor embodiment according to the present invention can be implemented using only passive circuit devices, and it can be implemented, for example, using existing CMOS technology. This RC processor embodiment, shown in
In step 15210, a matched filtering/correlating operation is performed on a portion of a carrier signal. For example, a match filtering/correlating operation can be performed on a 900 MHz RF signal, which typically comprises a 900 MHz sinusoid having noise signals and information signals superimposed on it. Many different types of signals can be operated upon in step 15210, however, and the invention is not limited to operating on a 900 MHz RF signal. In embodiments, Method 15200 operates on approximate half cycles of the carrier signal.
In an embodiment of the invention, step 15210 comprises the step of convolving an approximate half cycle of the carrier signal with a representation of itself in order to efficiently acquire the energy of the approximate half cycle of the carrier signal. As described elsewhere herein, other embodiments use other means for efficiently acquiring the energy of the approximate half cycle of the carrier signal. The matched filtering/correlating operation can be performed on any approximate half cycle of the carrier signal (although the invention is not limited to this), as described in detail in the sub-sections below.
In step 15220, the result of the matched filtering/correlating operation in step 15210 is accumulated, preferably in an energy storage device. In an embodiment of the present invention, a capacitive storage devise is used to store a portion of the energy of an approximate half cycle of the carrier signal.
Steps 15210 and 15220 are repeated for additional half cycles of the carrier signal. In an embodiment of the present invention, steps 15210 and 15220 are normally performed at a sub-harmonic rate of the carrier signal, for example at a third sub-harmonic rate. In another embodiment, steps 15210 and 15220 are repeated at an off-set of a sub-harmonic rate of the carrier signal.
In step 15230, a down-converted signal is output. In embodiments, the results of steps 15210 and 15220 are passed on to a reconstruction filter or an interpolation filter.
Switching module 15304 is controlled by a windowing function, u(t)−u(t−TA). The length of the windowing function aperture is TA, which is equal to an approximate half cycle of the received carrier signal, Si(t). Switching module 15304 ensures that approximate half cycles of the carrier signal are normally processed at a sub-harmonic rate. In an embodiment of system 15300, the received carrier signal is processed on at an off-set of a sub-harmonic rate of the carrier signal.
Capacitor 15306 integrates the output of switching module 15304 and accumulates the energy of the processed portions of the received carrier signal. RC processor 15300 also passes on its result, S0(t), to subsequent circuitry for further processing. This embodiment of the present invention is described in more detail in subsequent sub-sections.
It is noted that the implementations of the invention presented above are provided for illustrative purposes. Other implementations will be apparent to persons skilled in the art based on the herein teachings, and the invention is directed to such implementations.
This sub-section describes how a power signal can be represented as a sum of energy signals. The detailed mathematical descriptions in the sub-sections below use both Fourier transform analysis and Fourier series analysis to describe embodiments of the present invention. Fourier transform analysis typically is used to describe energy signals while Fourier series analysis is used to describe power signals. In a strict mathematical sense, Fourier transforms do not exist for power signals. It is occasionally mathematically convenient, however, to analyze certain repeating or periodic power signals using Fourier transform analysis.
Both Fourier series analysis and Fourier transform analysis can be used to describe periodic waveforms with pulse like structure. For example, consider the ideal impulse sampling train in EQ. (10).
Suppose that this sampling train is convolved (in the time domain) with a particular waveform s(t), which is of finite duration TA. Hence s(t) is an energy waveform. Then:
The above equation is a well known form of the sampler equation for arbitrary pulse shapes which may be of finite time duration rather than impulse-like. The sampler equation possesses a Fourier transform on a term-by-term basis because each separate is an energy waveform.
Applying the convolution theorem and a term-by-term Fourier transform yields:
where fs=Ts−1. In this manner the Fourier transform may be derived for a train of pulses of arbitrary time domain definition provided that each pulse is of finite time duration and each pulse in the train is identical to the next. If the pulses are not deterministic then techniques viable for stochastic signal analysis may be required. It is therefore possible to represent the periodic signal, which is a power signal, by an infinite linear sum of finite duration energy signals. If the power signal is of infinite time duration, an infinite number of energy waveforms are required to create the desired representation.
The method of
The heuristic discussion presented in the previous section can be applied to the piecewise linear reconstruction of a sine wave function or carrier.
Using the previously developed equations, the waveform y(t) can be represented by:
and y(t) can be rewritten as:
In general, Ts is usually integrally related to Tc. That is, the sampling interval Ts divided by Tc usually results in an integer, which further reduces the above equation. The unit step functions are employed to carve out the portion of a sine function applicable for positive pulses and negative pulse, respectively. The point is a power signal may be viewed as an infinite linear sum of energy signals.
Embodiments of the present invention are interpreted as a specific implementation of a matched filter and a restricted Fourier sine or cosine transform. The matched filter of such embodiments is not a traditional realization of a matched filter designed to extract information at the data bandwidth. Rather, the correlation properties of the filter of the embodiments exploit specific attributes of bandpass waveforms to efficiently down convert signals from RF. A controlled aperture specifically designed to the bandpass waveform is used. In addition, the matched filter operation of embodiments of the present invention is applied recursively to the bandpass signal at a rate sub-harmonically related to the carrier frequency. Each matched filtered result or correlation of embodiments of the present invention is retained and accumulated to provide an initial condition for subsequent recursions of the correlator. This accumulation is approximated as a zero order data hold filter.
An attribute of bandpass waveforms is that they inherently possess time domain structure, which can be compared to sampling processes. For example,
Sampled systems attempt to extract information in the envelope, at the black sample dots 15806, if possible. The sample times illustrated by the black sample dots 15806 are shown here at optimum sampling times.
Difficulties arise when the bandpass waveform is at RF. Then sampling is difficult because of sample rate, sample aperture, and aperture uncertainty. When the traditional sampler acquires, the aperture and aperture uncertainty must be minimized such that the number associated with the acquired waveform value possesses great accuracy at a particular instant in time with minimum variance. Sample rate can be reduced by sampling sub-harmonically. However, precisely controlling a minimized aperture makes the process very difficult, if not impossible, at RF.
In
Historically, an optimization figure of merit is signal-to-noise ration (SNR) at the system output.
Although an RF carrier with modulated information is typically a power signal, the analysis which follows considers the power signal to be a piece-wise construct of sequential energy signals where each energy waveform is a half sine pulse (single aperture) or multiple sine pulses (see sub-section 2 above). Hence, theorems related to finite time observations, Fourier transforms, etc., may be applied throughout.
Analysis begins with the assumption that a filtering process can improve SNR. No other assumptions are necessary except that the system is casual and linear. The analysis determines the optimum processor for SNR enhancement and maximum energy transfer.
The output of the system is given by the convolution integral illustrated in EQ. (17):
S
0(t)=∫0∞h(τ)Si(t−τ)dτ EQ. (17)
where h(τ) is the unknown impulse response of the optimum processor.
The output noise variance is found from EQ. (18):
π02=N0∫0∞h2(τ)dτ EQ. (18)
The signal to noise ratio at time t0 is given by EQ. (19):
The Schwarz inequality theorem may be used to maximize the above ratio by recognizing, in EQ. (20), that:
The maximum SNR occurs for the case of equality in EQ. 20, which yields EQ. (21):
In general therefore:
h(τ)=kSi(t0−τ)u(τ) EQ. (22)
where u(τ) is added as a statement of causality and k is an arbitrary gain constant. Since, in general, the original waveform Si(t) can be considered as an energy signal (single half sine for the present case), it is important to add the consideration of t0, a specific observation time. That is, an impulse response for an optimum processor may not be optimal for all time. This is due to the fact that an impulse response for realizable systems operating on energy signals will typically die out over time. Hence, the signal at t0 is said to possess the maximum SNR.
This can be verified by maximizing EQ. (21) in general.
It is of some interest to rewrite EQ. (21) by a change of variable, substituting
t=t
0−τ.
This yields:
k∫
2
∞
S
i
2(t0−τ)dτ=k∫−∞t
This is the energy of the waveform up to time to. After to, the energy falls off again due to the finite impulse response nature of the processor. EQ. (24) is of great importance because it reveals an often useful form of a matched filter known as a correlator. That is, the matched filter may be implemented by multiplying the subject waveform by itself over the time interval defined for the waveform, and then integrated. In this realization the maximum output occurs when the waveform and its optimal processor aperture are exactly overlapped for t0=Ta. It should also be evident from the matched filter equivalency stated in EQ. (24) that the maximum SNR solution also preserves the maximum energy transfer of the desired waveform through the processor. This may be proven using the Parseval and/or Rayliegh energy theorems. EQ. (24) relates directly to Parseval's theorem.
The previous sub-section derived an optimal processor from the time domain point-of-view according to embodiments of the invention. Alternately, Fourier transforms may be applied to obtain a frequency domain representation for h(t). This result is shown below.
H(f)=kSi(f)e−j2πft
Letting jω=j2Bf and t0=TA, we can write the following EQ. (26) for
The frequency domain representation in
Another simple but useful observation is gleaned from EQ. (24) and Rayleigh's Energy Theorem for Fourier transforms:
E=∫
−∞
∞
|S
i(t)|2dt=∫−∞∞|H(f)|2df EQ. (27)
EQ. (27) verifies that the transform of the optimal filter of various embodiments should substantially match the transform of the specific pulse, which is being processed, for efficient energy transfer.
It is not always practical to design the matched filter with passive networks. Sometimes the waveform correlation of Si(t) is also cumbersome to generate exactly. However, a single aperture realization of embodiments of the present invention is practical, even in CMOS, with certain concessions.
Consider
Applying EQ. (26) for both the matched filter and non-matched filter embodiments yields:
It turns out in practice that realizable apertures are not perfectly rectangular and do possess a finite rise and fall time. In particular, they become triangular or nearly sinusoidal for very high frequency implementations. Thus, the finite time integrating processor result tends toward the matched filtering/correlating processor result when the aperture becomes sine-like, if the processor possesses constant impedance across the aperture duration. Even though the matched filter/correlator response produces a lower output value at TA, it yields a higher SNR by a factor of 0.9 dB, as further illustrated below in sub-section 6.
Sometimes a precise matched filter is difficult to construct, particularly if the pulse shape is complex. Often, such complexities are avoided in favor of suitable approximations, which preserve the essential features. The single aperture realization of embodiments of the present invention is usually implemented conceptually as a first order approximation to a matched filter where the pulse shape being matched is a half-sine pulse. As shown in above, in embodiments, the matched filter is applied recursively to a carrier waveform. The time varying matched filter output correlation contains information modulated onto the carrier. If many such matched filter correlation samples are extracted, the original information modulated onto the carrier is recovered.
A baseband filter, matched or otherwise, may be applied to the recovered information to optimally process the signal at baseband. The present invention should not be confused with this optimal baseband processing. Rather embodiments of the present invention are applied on a time microscopic basis on the order of the time scale of a carrier cycle.
The switch 16504 functions as a sampler, which possesses multiplier attributes. Heviside's operator is used to model the switch function. The operator is multiplied in the impulse response, thus rendering it essential to the matched filtering/correlating process.
In the analysis that follows, only one aperture event is considered. That is, the impulse response of the circuit is considered to be isolated aperture-to-aperture, except for the initial value inherited from the previous aperture.
For circuit 16502, shown in
EQ. (31) represents the integro-differential equation for circuit 16502. The right hand side of EQ. (31) represents the correlation between the input waveform Vi(t) and a rectangular window over the period TA.
The Laplace transform of EQ. (31) is:
Consider that the initial condition equal to zero, then:
Suppose that
as illustrated in
By a change of variables;
Notice that the differential equation solution provides for carrier phase skew, φ. It is not necessary to calculate the convolution beyond TA since the gating function restricts the impulse response length.
Solving the differential equation for V0(t) permits an optimization of β=(RC)−1 for maximization of V0.
In embodiments, one might be tempted to increase β and cutoff earlier (i.e., arbitrarily reduce TA). However, this does not necessarily always lead to enhanced SNR, and it reduces charge transfer in the process. It can also create impedance matching concerns, and possibly make it necessary to have a high-speed buffer. That is, reducing TA and C is shown below to decrease SNR. Nevertheless, some gain might be achieved by reducing TA to 0.75 for β=2.6, if maximum voltage is the goal.
In embodiments, in order to maximize SNR, consider the following. The power in white noise can be found from:
Notice that σ2 is a function of RC.
The signal power is calculated from:
Hence, the SNR at TA is given by:
Maximizing the SNR requires solving:
Solving the SNRmax numerically yields β values that are ever decreasing but with a diminishing rate of return.
As can be seen in
In certain embodiments, it turns out that for an ideal matched filter the optimum sampling point corresponding to correlator peak is precisely TA. However, in embodiments, for the RC processor, the peak output of occurs at approximately 0.75 TA for large β (i.e., β=2.6). That is because the impulse response is not perfectly matched to the carrier signal. However, as β is reduced significantly, the RC processor response approaches the efficiency of the finite time integrating processor response in terms of SNR performance. As β is lowered, the optimal SNR point occurs closer to TA, which simplifies design greatly. Embodiments of the present invention provides excellent energy accumulation over TA for low β, particularly when simplicity is valued.
The basic equation for charge transfer is:
Similarly the energy u stored by a capacitor can be found from:
Thus, the charge stored by a capacitor is proportional to the voltage across the capacitor, and the energy stored by the capacitor is proportional to the square of the charge or the voltage. Hence, by transferring charge, voltage and energy are also transferred. If little charge is transferred, little energy is transferred, and a proportionally small voltage results unless C is lowered.
The law of conversation of charge is an extension of the law of the conservation of energy. EQ. (45) illustrates that if a finite amount of charge must be transferred in an infinitesimally short amount of time then the voltage, and hence voltage squared, tends toward infinity. The situation becomes even more troubling when resistance is added to the equation. Furthermore,
This implies an infinite amount of current must be supplied to create the infinite voltage if TA is infinitesimally small. Clearly, such a situation is impractical, especially for a device without gain.
In most radio systems, the antenna produces a small amount of power available for the first conversion, even with amplification from an LNA. Hence, if a finite voltage and current restriction do apply to the front end of a radio then a conversion device, which is an impulse sampler, must by definition possess infinite gain. This would not be practical for a switch. What is usually approximated in practice is a fast sample time, charging a small capacitor, then holding the value acquired by a hold amplifier, which preserves the voltage from sample to sample.
The analysis that follows shows that given a finite amount of time for energy transfer through a conversion device, the impulse response of the ideal processor, which transfers energy to a capacitor when the input voltage source is a sinusoidal carrier and possesses a finite source impedance, is represented by embodiments of the present invention. If a significant amount of energy can be transferred in the sampling process then the tolerance on the charging capacitor can be reduced, and the requirement for a hold amplifier is significantly reduced or even eliminated.
In embodiments, the maximum amount of energy available over a half sine pulse can be found from:
This points to a correlation processor or matched filter processor. If energy is of interest then a useful processor, which transfers all of the half sine energy, is revealed in EQ. (48), where TA is an aperture equivalent to the half sine pulse. In embodiments, EQ. (49) provides the clue to an optimal processor.
Consider the following equation sequence.
where h(θ)=Si(TA−θ) and t=TA−θ.
This is the matched filter equation with the far most right hand side revealing a correlator implementation, which is obtained by a change of variables as indicated. The matched filter proof for h(τ)=Si(TA−τ) is provided in sub-section 8.4 below. Note that the correlator form of the matched filter is exactly a statement of the desired signal energy. Therefore a matched filter/correlator accomplishes acquisition of all the energy available across a finite duration aperture. Such a matched filter/correlator can be implemented as shown in
In embodiments, when optimally configured, the example matched filter/correlator of
A matched filter/correlator embodiment according to the present invention might be too expensive and complicated to build for some applications. In such cases, however, other processes and processors according to embodiments of the invention can be used. The approximation to the matched filter/correlator embodiment shown in
Another very low cost and easy to build embodiment of the present invention is the RC processor. This embodiment, shown in
When maximum charge is transferred, the voltage across the capacitor 17304 in
Using EQs. (45) and (48) yields:
If it is accepted that an infinite amplitude impulse with zero time duration is not available or practical, due to physical parameters of capacitors like ESR, inductance and breakdown voltages, as well as currents, then EQ. (51) reveals the following important considerations for embodiments of the invention:
The impulse response for the RC processing network was found in sub-section 5.2 below to be;
Suppose that TA is constrained to be less than or equal to ½ cycle of the carrier period. Then, for a synchronous forcing function, the voltage across a capacitor is given by EQ. (54).
Maximizing the charge, q, requires maximizing EQ. (37) with respect to t and β.
It is easier, however, to set R=1, TA=1, A=1, fA=TA−1 and then calculate q=cV0 from the previous equations by recognizing that
which produces a normalized response.
In embodiments, EQ. (49) establishes TA as the entire half sine for an optimal processor. However, in embodiments, optimizing jointly for t and β reveals that the RC processor response creates an output across the energy storage capacitor that peaks for tmax≅0.75TA, and βmax≅2.6, when the forcing function to the network is a half sine pulse.
In embodiments, if the capacitor of the RC processor embodiment is replaced by an ideal integrator then tmax+TA.
βTA≅1.95 EQ. (56)
where ∃=(RC)−1
For example, for a 2.45 GHz signal and a source impedance of 50Ω, EQ. (56) above suggests the use of a capacitor of ≅2 pf. This is the value of capacitor for the aperture selected, which permits the optimum voltage peak for a single pulse accumulation For practical realization of the present invention, the capacitance calculated by EQ. (56) is a minimum capacitance. SNR is not considered optimized at βTA≅1.95. As shown earlier, a smaller β yields better SNR and better charge transfer. In embodiments, as discussed below, it turns out that charge can also be optimized if multiple apertures are used for collecting the charge.
In embodiments, for the ideal matched filter/correlator approximation, βTA is constant and equivalent for both consideration of optimum SNR and optimum charge transfer, and charge is accumulated over many apertures for most practical designs. Consider the following example, β=0.25, and TA=1. Thus βTA=0.25. At 2.45 GHz, with R=50Ω, C can be calculated from:
The charge accumulates over several apertures, and SNR is simultaneously optimized melding the best of two features of the present invention. Checking CV for βTA≅1.95 vs. βTA=0.25 confirms that charge is optimized for the latter.
The general forms of the differential equation and transfer function, described above, for embodiments of the present invention are the same as for a case involving a load resistor, RL, applied across capacitor, C.
Consider RC processing embodiment 17502 (without initial conditions). EQ. (33) becomes:
It should be clear that RL 17504, and therefore k, accelerate the exponential decay cycle.
This result is valid only over the acquisition aperture. After the switch is opened, the final voltage that occurred at the sampling instance t≅TA becomes an initial condition for a discharge cycle across RL 17504. The discharge cycle possesses the following response:
VA is defined as V0(t≅TA). Of course, if the capacitor 17506 does not completely discharge, there is an initial condition present for the next acquisition cycle.
Equations 63.1 through 63.15 derive a relationship between the capacitance of the capacitor CS (CS(R)), the resistance of the resistor R, the duration of the aperture A (aperture width), and the frequency of the energy transfer pulses (freq LO). Equation 63.11 illustrates that optimum energy transfer occurs when x=0.841. Based on the disclosure herein, one skilled in the relevant art(s) will realize that values other that 0.841 can be utilized.
Maximum power transfer occurs when:
Let VC
The prior sub-sections described the basic SNR definition and the SNR of an optimal matched filter/correlator processor according to embodiments of the present invention. This sub-section section describes the SNR of additional processor embodiments of the present invention and compares their SNR with the SNR of a optimal matched filter/correlator embodiment. The description in this sub-section is based on calculations relating to single apertures and not accumulations of multiple aperture averages. Since SNR is a relative metric, this method is useful for comparing different embodiments of the present invention.
EQ. (65), which can be obtained from EQ. (64), represents the output SNR for a single aperture embodiment assuming a constant envelope sine wave input. The results could modify according to the auto-correlation function of the input process, however, over a single carrier half cycle, this relationship is exact.
The description that follows illustrates the SNR for three processor embodiments of the present invention for a given input waveform. These embodiments are:
Consider an example finite time integrator processor, such as the one illustrated in
h(t)=k,0≦t≦TA EQ. (66)
where k is defined as an arbitrary constant.
The output of the finite time integrator processor, y(t), is found from the input, x(t), using:
y(t)=∫t-T
A change of variables yields EQ. (68):
y(t−τ)=∫t-τ-T
The output auto correlation then becomes that shown in EQ. (69):
R
v(τ)=∫t-T
which leads to:
This Fourier transform may be substituted into the expression for Ry(τ), in EQ. (71), which becomes:
Sy (ω) is the power spectral density at the output of the example finite time integrator, whose integration aperture is TA and whose input power spectrum is defined by Sx(ω). For the case of wide band noise:
The total noise power across the band can be found from EQ. (75):
This result can be verified by EQ. (76):
Y
2
The signal power over a single aperture is obtained by EQ. (77):
y(t)2=(2A∫0T
Choosing A=1, the finite time integrator output SNR becomes:
An example RC filter can also be used to model an embodiment of the present invention. The mean squared output of a linear system may be found from EQ. (79):
Y
2
For the case of input AWGN:
R
xn(τ)=N0δ(τ) EQ. (80)
Y
2
Y
n
2
=N
0∫0∞h2(τ)dτ EQ. (82)
This leads to the result in EQ. (83):
R is the resistor associated with processor source, and C is the energy storage capacitor.
And finally:
The detailed derivation for the signal voltage at the output to the RC filter is provided in sub-section 5 above. The use of the 13 parameter is also described in sub-section 5. Hence, the SNRRC is given by:
Illustrative SNR performance values of the three example processor embodiments of the present invention are summarized in the table below:
Notice that as the capacitor becomes larger, the RC processor behaves like a finite time integrator and approximates its performance. As described above in sub-section 5, with a β of 0.25, a carrier signal of 2450 MHz, and R=5052, the value for C becomes C≧16.3 pf.
The second waveform 17760 illustrates the same rect function envelope at passband (RF) and it's matched filter impulse response. Notice the sine function phase reversal corresponding to the required time axis flip.
The fact that a non-coherent processor is used or a differentially coherent BB processor used in lieu of a coherent Costas Loop in no way diminishes the contribution of the UFT correlator effect obtained by selecting the optimal aperture TA based on matched filter theory.
Consider
Moreover, Section IV, part 5.1 above illustrates that a complex UFT downconverter which utilizes a bandpass filter actually resembles the optimal matched filter/correlator kernel in complex form with the in phase result scaled by cos φ and the quadrature phase component scaled by sin φ. This process preserves all the energy of the downconverter signal envelope (minus system loses) with a part of the energy in I and the remainder in Q.
The above sub-sections describe single aperture embodiments of the present invention. That is, the above sub-sections describe the acquisition of single half sine waves according to embodiments of the invention. Other embodiments of the present invention are also possible, however, and the present invention can be extended to other waveform partitions that capture multiple half sine waves. For example, capturing two half sine waves provides twice the energy compared to capturing only a single half sine. Capturing n half sines provides n times the energy, et cetera, until sub harmonic sampling is no longer applicable. The invention is directed to other embodiments as well. Of course, the matched filter waveform requires a different correlating aperture for each new n. This aspect of the present invention is illustrated in
In the example of
Fourier transforming the components for the example processor yields the results shown in
The transform of the periodic, sampled, signal is first given a Fourier series representation (since the Fourier transform of a power signal does not exist in strict mathematical sense) and each term in the series is transformed sequentially to produce the result illustrated. Notice that outside of the desired main lobe aperture response that certain harmonics are nulled by the (sin x)/x response. Even those harmonics, which are not completely nulled, are reduced by the side lobe attenuation. Some sub-harmonics and super-harmonics are eliminated or attenuated by the frequency domain nulls and side lobes of the bipolar matched filter/correlator processor, which is a remarkable result.
Theoretically, arbitrary impulse responses may be constructed in the manner above, particularly if weighting is applied across the aperture or if multiple apertures are utilized to create a specific Fourier response. FIR filters and convolvers may be constructed by extending the aperture and utilizing the appropriate weighting factors. Likewise, disjoint or staggered apertures may be constructed to provide a particular desired impulse response. These apertures can be rearranged and tuned ‘on the fly’.
The operation of the present invention represents a new signal-processing paradigm. Embodiments of the invention can be shown to be related to particular Fourier sine and cosine transforms. Hence, the new term UFT transform is utilized to refer to the process. As already stated, in embodiments of the present invention can be viewed as a matched filter or correlator operation, which in embodiments is normally applied recursively to the carrier signal at a sub-harmonic rate. A system equation may be written to describe this operation, assuming a rectangular sample aperture and integrators as operators, as shown in
where:
Dn represents the UFT transform applicable to embodiments of the invention. The first term defines integration over a rectangular segment of the carrier signal of TA time duration. k pulses are summed to form a memory of the recursively applied kernel. The second term in the equation provides for the fact that practical implementations possess finite memory. Hence, embodiments of the present invention are permitted to leak after a fashion by selecting α and l. This phenomena is reflected in the time variant differential equation, EQ. (31), derived in sub-section 5. In embodiments, for a perfect zero order data hold function, α=0.
The UFT kernel applicable to embodiments of the invention is given by EQ. (89):
D
1=∫0T
EQ. 89 accounts for the integration over a single aperture of the carrier signal with arbitrary phase, φ, and amplitude, A. Although A and φ are shown as constants in this equation, they actually may vary over many (often hundreds or thousands) of carrier cycles. Actually, φ(t) and A(t) may contain the modulated information of interest at baseband. Nevertheless, over the duration of a pulse, they may be considered as constant.
Ever since Nyquist developed general theories concerning waveform sampling and information extraction, researchers and developers have pursued optimum sampling techniques and technologies. In recent years, many radio architectures have embraced these technologies as a means to an end for ever more ‘digital like’ radios. Sub sampling, IF sampling, syncopated sampling, etc., are all techniques employed for operating on the carrier to extract the information of interest. All of these techniques share a common theory and common technology theme, i.e., Nyquist's theory and ideal impulse samplers. Clearly, Nyquist's theory is truly ideal, from a theoretical perspective, while ideal impulse samplers are pursued but never achieved.
Consider the method of developing an impulse sample using functions with shrinking apertures, as illustrated in
As would be apparent to persons skilled in the relevant arts given the discussion herein, an arbitrary capacitance, c, cannot be charged in an infinitesimally short time period without an infinite amount of energy. Even approximations to an ideal impulse therefore can place unrealistic demands on analog sample acquisition interface circuits in terms of parasitic capacitance vs. pulse width, amplitude, power source, etc. Therefore, a trade-off is typically made concerning some portion of the mix.
The job of a sample and hold circuit is to approximate an ideal impulse sampler followed by a memory. There are limitations in practice, however. A hold capacitor of significant value must be selected in order to store the sample without droop between samples. This requires a healthy charging current and a buffer, which isolates the capacitor in between samples, not to mention a capacitor, which is not ‘leaky,’ and a buffer without input leakage currents. In general, ideal impulse samplers are very difficult to approximate when they must operate on RF waveforms, particularly if IC implementations and low power consumption are required.
The ideal sample extraction process is mathematically represented in EQ. (92) by the sifting function.
where:
Δ Sample Time; x(t)Δ Sampled Function; and δ(t)Δ Impulse Sample Function.
Suppose now that:
x(t)=A sin(t+φ) EQ. (93)
then:
This represents the sample value acquired by an impulse sampler operating on a carrier signal with arbitrary phase shift φ. EQ. (95) illustrates that the equivalence of representing the output of the sampler operating on a signal, {tilde over (X)}(t), without phase shift, φ, weighted by cos φ, and the original sampled X(t), which does have a phase shift. The additional requirement is that a time aperture of TA corresponds to π radians.
Next, consider the UFT kernel:
D
1
Δ∫−∞∞(u(t)−u(t−TA))sin(t+φ)dt EQ. (96)
Using trigonometric identities yields:
D
1
ΔA cos(φ)∫−∞∞(u(t)−u(t−TA))sin(t)dt EQ. (97)
Now the kernel does not possess a phase term, and it is clear that the aperture straddles the sine half cycle depicted in
Consider the ideal aperture of embodiments of the invention shown in
D
1
ΔA cos(φ)[∫−∞∞u(t)−u(TA/2))sin(t)dt+∫−∞∞(u(t−TA/2)−u(t−TA))sin(t)dt] EQ. (98)
It should also be apparent to those skilled in the relevant arts given the discussion herein that the first integral is equivalent to the second, so that;
D
1=2A cos(φ)∫−∞∞[∫−∞tδ(t′)dt′−∫−∞rδ(t′−TA/2)dt′] sin(t)dt EQ. (99)
As illustrated in
D
1=2A cos(φ)∫−∞∞[∫−∞tδ(t′)dt′−∫−∞tδ(t′−TA/2)dt′] sin(t)dt EQ. (100)
Using the principle of integration by parts yields EQ. (101).
This is a remarkable result because it reveals the equivalence of the output of embodiments of the present invention with the result presented earlier for the arbitrarily phased ideal impulse sampler, derived by time sifting. That is, in embodiments, the UFT transform calculates the numerical result obtained by an ideal sampler. It accomplishes this by averaging over a specially constructed aperture. Hence, the impulse sampler value expected at TA/2 is implicitly derived by the UFT transform operating over an interval, TA. This leads to the following very important implications for embodiments of the invention:
The following analysis utilizes concepts of the convolution property for the sampling waveform and properties of the Fourier transform to analyze the complex clock waveform for the UFT as well as the down conversion correlation process.
In addition r(t) is considered filtered, by a bandpass filter. In one exemplary embodiment, sub-optimal correlators approximate the UFT. This analysis illustrates that some performance is regained when the front-end bandpass filter is used, such that the derived correlator kernel resembles the optimal form obtained from matched filter theory. Furthermore, the analysis illustrates that the arbitrary phase shift of a carrier on which the UFT operates, does not alter the optimality of the correlator structure which can always be modeled as a constant times the optimal kernel. This is due to the fact that UFT is by definition matched to a pulse shape resembling the carrier half cycle which permits phase skew to be viewed as carrier offset rather than pulse shape distortion.
Using the pulse techniques described above, describing pulse trains, the clock signal for UFT may be written as equation 18802 of
In
Although the approximation is used, ideal carrier tracking for coherent demodulation will yield an equal sign after lock. However, this is not required to attain the excellent benefit from UFT processing. Other sections herein provide embodiments that develop expressions for CI and CQ from Fourier series analysis to illustrate the components of the gating waveforms at the Carrier frequency which are harmonically related to Ts.
By the methods described above, the Fourier transform of the clock is found from:
C
Q(f)=CI(f)e−jnπfT
radians phase skew for CQ relative to CI.
S
0(t)=∫0T
S
0=∫0T
This analysis assumes that r(t), the input carrier plus noise, is band limited by a filter. In this case therefore the delta function comb evident in the transform of CI and CQ are ignored except for the components at the carrier. Embodiments in other sections break CI and CQ into a Fourier series. In this series, only the harmonic of interest would be retained when the input waveform r(t) is bandpass limited because all other cross correlations tend to zero. Hence,
S
0(t)≅K∫0T
S
0(t)≅K∫0T
The clock waveforms have been replaced by the single sine and cosine components from the Fourier transform and Fourier series, which produce the desired result due to the fact that a front-end filter filters all other spectral components. This produces a myriad of cross correlations for the complex UFT processor. K is included as a scaling factor evident in the transform.
A and φ are the original components of the complex modulation envelope (amplitude and phase) for the carrier and are assumed to vary imperceptibly over the duration for TA. What is very interesting to note is that the above equations are exactly the optimum form for the complex correlator whose pulse shape is a half sine with components weighted by cosine for I, and sine for Q. Furthermore, when an input bandpass filter is considered as a part of the system then the approximate kernels used throughout various analyses based on the gating function become replaced by the ideal matched filter analogy. Hence, the approximation in CMOS using rectangular gating functions, which are known to cause only a 0.91 dB hit in performance if C is selected correctly, probably can be considered pessimistic if the receiver front end is filtered.
As illustrated in
0DH Portion of Present Invention Impulse Response
The embodiment in
The ultimate output includes the hold phase of the operation and is written as:
This embodiment considers the aperture operation as implemented with an ideal integrator and the hold operation as implemented with the ideal integrator. As shown elsewhere herein, this can be approximated by energy storage in a capacitor under certain circumstances.
The acquisition portion of the operation possesses a Fourier transform given by:
Si(ω)=ℑ{r(t)} (Modulated Information Spectrum)
S0(ω) can be found in a similar manner.
The example of
The acquisition portion of the Fourier transform yields the following an important insight:
As should be apparent to persons skilled in the relevant arts given the discussion herein, down conversion occurs whenever kωs=ωc. It is useful to find TA, which maximizes the component of the spectrum at ωc, which is subject to down conversion and is the desired signal. This is accomplished simply by examining the kernel.
For ω=ωc,
The kernel is maximized for values of
Advocates of impulse samplers might be quick to point out that letting TA→0 maximizes the sinc function. This is true, but the sinc function is multiplied by TA in the acquisition phase. Hence, a delta function that does not have infinite amplitude will not acquire any energy during the acquisition phase of the sampler process. It must possess infinite amplitude to cancel the effect of TA→0 so that the multiplier of the sinc function possesses unity weighting. Clearly, this is not possible for practical circuits.
On the other hand, embodiments of the present invention with
does pass significant calculable energy during the acquisition phase. This energy is directly used to drive the energy storage element of Z0DH filter or other interpolation filter, resulting in practical RF impedance circuits. The cases for TA/Tc other than ½ can be represented by multiple correlators, for example, operating on multiple half sine basis.
Moreover, it has been shown that the specific gating aperture, C(t), does not destroy the information. Quite the contrary, the aperture design for embodiments of the present invention produces the result of the impulse sampler, scaled by a gain constant, and possessing less variance. Hence, the delta sifting criteria, above trigonometric optimization, and correlator principles all point to an aperture of
nominal.
If other impulse responses are added around the present invention (i.e., energy storage networks, matching networks, etc.) or if the present invention is implemented by simple circuits (such as the RC processor) then in embodiments the optimal aperture can be adjusted slightly to reflect the peaking of these other embodiments. It is also of interest to note that the Fourier analysis above predicts greater DC offsets for increasing ratios of
Therefore, for various embodiments,
is probably the best design parameter for a low DC offset system.
The sine and cosine transforms are defined as follows:
F
c(ω)Δ∫0∞f(t)sin ωtdtω≧0 (sine transform) EQ. (116)
F
s(ω)Δ∫0∞f(t)cos ωtdtω≧0 (cosine transform) EQ. (117)
Notice that when f (0 is defined by EQ. (118):
f(t)=u(t)−u(u−TA) EQ. (118)
the UFT transform kernel appears as a sine or cosine transform depending on φ. Hence, many of the Fourier sine and cosine transform properties may be used in conjunction with embodiments of the present invention to solve signal processing problems.
The following sine and cosine transform properties predict the following results of embodiments of the invention:
Of course many other properties are applicable as well. The subtle point presented here is that for embodiments the UFT transform does in fact implement the transform, and therefore inherently possesses these properties.
Consider the following specific example: let f (0=u(t)−u(t−TA) and let ω=2πf=πfA=1.
This is precisely the result for D1c and D1s. Time shifting yields:
ℑs[f0(t+Ts)+f0(t−Ts)]=2Fs(ω)cos(Tsω) (Time Shift Property)
Let the time shift to be denoted by Ts.
Notice that f0(t) has been formed due to the single sided nature of the sine and cosine transforms. Nevertheless, the amplitude is adjusted by ½ to accommodate the fact that the energy must be normalized to reflect the odd function extension. Then finally:
which is the same solution for phase offset obtained earlier by other means.
The implications of this transform may be far reaching when it is considered that the discrete Fourier sine and cosine transforms are originally based on the continuous transforms as follows:
That is, the original kernel cos(ωt) and function ƒ(t) are sampled such that:
f(n)Δ Sampled Version of f(t)
ωm=2πmΔf
tn=nΔt
ΔfΔ Frequency Sample Interval
ΔtΔ Time Sample Interval
Hence the new discrete cosine transform kernel is:
k
c(m,n)=cos(2πmnΔfΔt)=cos(πmn/n)ΔfΔt=½N EQ. (126)
N is the total number of accumulated samples for m, n, or the total record length.
In recent years, the discrete cosine transform (DCT) and discrete sine transform (DST) have gained much recognition due to their efficiency for waveform coding compression, spectrum analysis, etc. In fact, it can be shown that these transforms can approach the efficiency of Karhunen-Loeve transforms (KLT), with minimal computational complexity. The implication is that the sifted values from D1 could be used as DCT sample values f(n). Then the DCT and DST properties will apply along with their processing architectures. In this manner, communications signals, like OFDM, could be demodulated in a computationally efficient manner. Many other signal processing applications are possible using the present invention, and the possibilities are rich and varied.
The previous sub-sections described how embodiments of the present invention involve gating functions of controlled duration over which integration can occur. This section now addresses some consideration for the controlling waveform of the gating functions.
For sub harmonic sampling:
f
s
=f
c
/M
The subsequent equations illustrate the sampling concept, with an analysis base on approximations that ignore some circuit phenomena. A more rigorous analysis requires explicit transformation of the circuit impulse response. This problem can be solved by convolving in the time domain as well, as will be apparent to persons skilled in the relevant arts given the discussion herein. The results will be the same. The analysis presented herein is an abbreviated version of one provided above. As in the subsection 8, the acquisition portion of the present invention response is analyzed separately from the hold portion of the response to provide some insight into each. The following sub-section uses a shorthand notation for convenience.
X0(t)Δ Output of Sample
Si[t]Δ Waveform being Sampled
kΔ Sampling Index
TsΔ Sampling Interval=fs−1
{tilde over (C)}(t−kTs)Δ Quasi-Matched Filter/Correlator Sampling Aperture, which includes averaging over the Aperture.
EQ. (127) can be rewritten a:
If {tilde over (C)}(t) possesses a very small aperture with respect to the inverse information bandwidth, TA<<BWi−1 then the sampling aperture will weight the frequency domain harmonics of L. The Fourier transform and the modulation property may be applied to EQ. (128) to obtain EQ. (129) (note this problem was solved above by convolving in the time domain).
X
0(ω)=(Si(ω)c{tilde over (C)}(ω)) EQ. (129)
KΔ Arbitrary Gain Constant, which includes a ½π factor
ωΔ 2πf
Essentially, on the macroscopic frequency scale, there is a harmonic sample comb generated, which possesses components at every Nfs for N=1, 2, 3 . . . ∞, with nulls at every Z·fA where fA is defined as TA−1.
The thickness of each spike in
Notice that each harmonic including baseband possesses a replica of Si(ω) which is in fact the original desired signal. {Si(ω) is the original information spectrum and is shown to survive the acquisition response of the present invention (i.e., independent integration over each aperture)}. Lathi and many others pointed out that {tilde over (C)}(ω) could be virtually any harmonic function and that conversion to baseband or passband will result from such operations on Si(t).
Each discrete harmonic spectrum provides a potential down conversion source to baseband (at DC). Of course, theoretically, there cannot be a conversion of Z·fa because of the spectral nulls.
It should also be noted that in all practical cases, fs>>2·BW; so that Nyquist criteria are more than satisfied. The lowpass response of embodiments of the present invention can be ideally modeled as a zero order data hold filter, with a finite time integrator impulse response duration of T=Ts−TA. The ultimate output Fourier transform is given by EQ. (131).
The Z0DH is a type of lowpass filter or sample interpolator which provides a memory in between acquisitions. Each acquisition is accomplished by a correlation over TA, and the result becomes an accumulated initial condition for the next acquisition.
Typically, processor embodiments of the present invention sample at a sub-harmonic rate. Hence the carrier frequency and associated bandpass signal are down converted by a M·fs harmonic. The harmonic generation operation can be represented with a complex phasor.
S
amp(t)Δ(e−jω
Samp(t) can be rewritten as:
S
amp(t)=e−jMω
φ(t)Δ Phase Noise on the Conversion Clock
As EQ. (133) indicates, not only is the frequency content of the phasor multiplied by M but the phase noise is also multiplied by M. This results in an M-tuple convolution of the phase noise spectrum around the harmonic. The total phase noise power increase is approximated by EQ. (134).
φ=Δ20 log10 M (Phase Noise) EQ. (134)
That is, whatever the phase jitter component, φ(t), existing on the original sample clock at Mfs, it possesses a phase noise floor degraded according to EQ. (134).
This section describes what the conversion constant and the output noise is for AM to PM conversion according to embodiments of the present invention, considering the noise frequency of the threshold operation. As illustrated in
The slope at the zero crossings of a pure sine wave, s(t)=A sin ωt, can be calculated. Differentiating s(t) with respect to t yields s(t)′=ωA cos ωt. For ω A≠0, the zero crossings occur at ωt=π/2, 3π/2, 5π/2 . . . .
These zero crossings represent the points of minimum slope or crests of the original s(t). The maximum slope is found at the zero crossings of s(t) at ωt=0, π, 2π, . . . etc. Plugging those arguments into s(t)′ give slopes of: Slope=ωA, −ωA, ωA, −ωA . . . etc. The time at which these zero crossings occur is given by:
It stands to reason that for the low noise power assumption, which implies one zero crossing per carrier cycle, the slope at the zero crossing will be modified randomly if a Gaussian process (n(t)) is summed to the signal. Of course, if the change in slope of the signal is detectable, the delta time of the zero crossing is detectable, and hence phase noise is produced. The addition of noise to the signal has the effect of moving the signal up and down on the amplitude axis while maintaining a zero mean. This can be written more formally as:
If A is replaced by A−Δa, where Δa represents the noise deviation, then one will not always observe a zero crossing at the point of maximum slope ωA. Sometimes the zero crossing will occur at ω(A−Δa). This leads to the low noise approximation:
The low noise assumption implies that the low noise power prohibits the arcos function from transforming the Gaussian pdf of the noise. That is, ±Δa occurs over minute ranges for the argument of the arcos and hence the relationship is essentially linear. Secondly, since A is a peak deviation in the sine wave Δa will be considered as a peak deviation of the additive noise process. This is traditionally accepted as being 4σ where σ is the standard deviation of the process and σ2 is the variance. Therefore we write K arcos (1−4σ/A)=t±∈, where ∈ represents a peak time deviation in the zero crossing excursion, K=1/ω, and t is the mean zero crossing time given previously as: t=1/sf, 1/f, 3/2f, . . . . If only the deviation contribution to the above equation is retained, the equation reduces to:
Since for 4σ/A<<0.01, the above function is quasi-linear, one can write the final approximation as:
An appropriate conversion to degrees becomes,
fc=frequency of carrier
σx=phase noise in degrees rms
σ=standard deviation of equivalent input comparator noise
Now a typical threshold operator may have a noise figure, NF, of approximately 15 dB. Hence, one can calculate σx (assume σφ2=2.4×10−8 rad2 source phase noise):
−174 dBm/Hz+15+10 log10 100×106=−79 dBm EQ. (143)
where 100 MHz of input bandwidth is assumed.
σθ2=phase noise of source before threshold device
Therefore, the threshold device has little to no impact on the total phase noise modulation on this particular source because the original source phase noise dominates. A more general result can be obtained for arbitrarily shaped waveforms (other than simple sine waves) by using a Fourier series expansion and weighting each component of the series according to the previously described approximation. For simple waveforms like a triangle pulse, the slope is simply the amplitude divided by the time period so that in the approximation:
k; an arbitrary scaling constant
Tr; time period for the ramping edge of the triangle
Hence, the ratio of (σTr/Ar) is important and should be minimized. As an example, suppose that the triangle pulse rise time is 500 nsec. Furthermore, suppose that the amplitude, AT, is 35 milli volts. Then, with a 15 dB NF, the Δt becomes:
This is all normalized to a 1Ω system. If a 50Ω system were assumed then: σ˜358.5 ps (50Ω)
In addition, it is straight forward to extend these results to the case of DC offset added to the input of the threshold device along with the sine wave. Essentially the zero crossing slope is modified due to the virtual phase shift of the input sine function at the threshold. DC offset will increase the phase noise component on the present invention clock, and it could cause significant degradation for certain link budgets and modulation types.
Examples and derivations presented in previous sub-sections illustrate that in embodiments single aperture acquisitions recover energies proportional to:
AnΔ as the carrier envelope weighting of the nth sample.
In addition, sub-section 8 above, describes a complete UFT transform over many pulses applicable to embodiments of the invention. The following description therefore is an abbreviated description used to illustrate a long-term time constant consideration for the system.
As described elsewhere herein, the sample rate is much greater than the information bandwidth of interest for most if not all practical applications.
f
s
>>BW
i EQ. (148)
Hence, many samples may be accumulated as indicated in previous sub-sections, provided that the following general rule applies:
where l represents the total number of accumulated samples. EQ. (149) requires careful consideration of the desired information at baseband, which must be extracted. For instance, if the baseband waveform consists of sharp features such as square waves then several harmonics would necessarily be required to reconstruct the square wave which could require BWi of up to seven times the square wave rate. In many applications however the base band waveform has been optimally prefiltered or bandwidth limited apriori (in a transmitter), thus permitting significant accumulation. In such circumstances, fs/l will approach BWi.
This operation is well known in signal processing and historically has been used to mimic an average. In fact it is a means of averaging scaled by a gain constant. The following equation relates to EQ. (127).
Notice that the nth index has been removed from the sample weighting. In fact, the bandwidth criteria defined in EQ. (149) permits the approximation because the information is contained by the pulse amplitude. A more accurate description is given by the complete UFT transform, which does permit variation in A. A cannot significantly vary from pulse to pulse over an l pulse interval of accumulation, however. If A does vary significantly, l is not properly selected. A must be permitted to vary naturally, however, according to the information envelope at a rate proportional to BWi. This means that l cannot be permitted to be too great because information would be lost due to filtering. This shorthand approximation illustrates that there is a long term system time constant that should be considered in addition to the short-term aperture integration interval.
In embodiments, usually the long term time constant is controlled by the integration capacitor value, the present invention source impedance, the present invention output impedance, and the load. The detailed models presented elsewhere herein consider all these affects. The analysis in this section does not include a leakage term that was presented in previous sub-sections.
EQs. (149) and (150) can be considered a specification for slew rate. For instance, suppose that the bandwidth requirement can be specified in terms of a slew rate as follows:
The number of samples per pec is given by:
l
s
=f
s×1×10−6
(fs is derived from the present invention clock rate)
If each sample produces a voltage proportional to A2 TA/2 then the total voltage accumulated per microsecond is:
The previous sub-sections illustrates how the present invention output can accumulate voltage (proportional to energy) to acquire the information modulated onto a carrier. For down conversion, this whole process is akin to lowpass filtering, which is consistent with embodiments of the present invention that utilize a capacitor as a storage device or means for integration.
The previous sub-sections introduced the idea that in embodiments information bandwidth is much less than the bandwidth associated with the present invention's impulse response for practical applications. The concept of single aperture energy accumulation was used above to describe the central ideas of the present invention. As shown in
The staircase output of the example in
Consider the following equation for a window correlator aperture:
E
ASO=∫0TAA·Si(t)dt EQ. (153)
In EQ. (153), the rectangular aperture correlation function is weighted by A. For convenience, it is now assumed to be weighted such that:
E
ASO=∫0TAkA·Si(t)dt=2A (normalized) EQ. (154)
Since embodiments of the present invention typically operate at a sub-harmonic rate, not all of the energy is directly available due to the sub-harmonic sampling process. For the case of single aperture acquisition, the energy transferred versus the energy available is given by:
NΔ harmonic of operation
The power loss due to harmonic operation is:
E
LN=10 log10(2N) EQ. (156)
There is an additional loss due to the finite aperture, TA, which induces (sin x/x) like weighting onto the harmonic of interest. This energy loss is proportional to:
N·fsΔ operating carrier frequency
fsΔ sampling rate (directly related to the clock rate)
EQ. (157) indicates that the harmonic spectrum attenuates rapidly as N·fs approaches TA−1. Of course there is some attenuation even if that scenario is avoided. EQ. (157) also reveals, however, that in embodiments for single aperture operation the conversion loss due to ELSINC will always be near 3.92 dB. This is because:
(2·Nfs)−1=TA (−3.92 dB condition) EQ. (158)
Another way of stating the condition is that TA is always ½ the carrier period.
Consider an ideal implementation of an embodiment of the present invention, without any circuit losses, operating on a 5th harmonic basis. Without any other considerations, the energy loss through the device is at minimum:
E
L
=E
LN
+E
LSINC=10 dB+3.92∞14 dB (for up conversion) EQ. (159)
Down conversion does not possess the 3.92 dB loss so that the baseline loss for down conversion is that represented by EQ. (156). Parasitics will also affect the losses for practical systems. These parasitics must be examined in detail for the particular technology of interest.
Next suppose that a number of pulses may be accumulated using the multi-aperture strategy and diversity means of an embodiment of the present invention, as described above. In this case, some of the energy loss calculated by EQ. (159) can be regained. For example, if four apertures are used then the pulse energy accumulation gain is 6 dB. For the previous example, this results in an overall gain of 6 dB-14 dB, or −8 dB (instead of −14 dB). This energy gain is significant and will translate to system level specification improvements in the areas of noise frequency, intercept point, power consumption, size, etc. It should be recognized, however, that a diversity system with active split or separate amplifier chains would use more power and become more costly. In addition, in embodiments, energy storage networks coupled to the circuitry of the present invention may be used to accumulate energy between apertures so that each aperture delivers some significant portion of the stored energy from the network. In this manner, some inefficiencies of the sub harmonic sampling process can be removed by trading impedance matching vs. complexity, etc., as further described below.
Embodiments of the present invention have been shown to be a type of correlator, which is applied to the carrier on a sub harmonic basis. It is also been shown herein that certain architectures according to embodiments of the invention benefit significantly from the addition of passive networks, particular when coupled to the front end of a processor according to the present invention used as a receiver. This result can be explained using linear systems theory.
To understand this, it is useful to consider the following. Embodiments of the present invention can be modeled as a linear, time-variant (LTV) device. Therefore, the following concepts apply:
These are powerful concepts because they permit the application of the maximum bilateral power transfer theorem to embodiments of the present invention. As a result, in embodiments, energy storage devices/circuits which fly wheel between apertures to pump up the inter sample power can be viewed on the many sample basis (long time average) as providing optimum power transfer through matching properties. The between sample model on the time microscopic scale is best viewed on a differential equation basis while the time macroscopic view can utilize simpler analysis techniques such as the maximum power transfer equations for networks, correlator theory, etc. The fact that the differential equations can be written for all time unifies the theory between the short time (between sample) view and long time (many sample accumulation) view. Fortunately, the concepts for information extraction from the output of the present invention are easily formulated without differential equation analysis.
Network theory can be used to explain why certain networks according to the present invention provide optimum power gain. For example, network theory explains embodiments of the present invention when energy storage networks or matching networks are utilized to ‘fly wheel’ between apertures, thereby, on the average, providing a good impedance match. Network theory does not explain, however, why TA is optimal. For instance, in some embodiments, one may deliberately utilize an aperture that is much less than a carrier half cycle. For such an aperture, there is an optimal matching network nonetheless. That is, a processor according to an embodiment of the present invention utilizing an improper aperture can be optimized, although it will not perform as well as a processor according to an embodiment of the present invention that utilizes an optimal aperture accompanied by an optimal matching network.
The idea behind selecting an optimal aperture is matched filter theory, which provides a general guideline for obtaining the best correlation properties between the incoming waveform and the selected aperture. Any practical correlator or matched filter is constrained by the same physical laws, however, which spawned the maximum power transfer theorems for networks. It does not do any good to design the optimum correlator aperture if the device possesses extraordinary impedance mismatches with its source and load. The circuit theorems do predict the optimal impedance match while matched filter theory does not. The two work hand in hand to permit a practical explanation for:
The following sub-section analyzes the present invention on a macroscopic scale using the notions of average impedance and power transfer.
When a processor embodiment according to the present invention is ‘off,’ there is one impedance, and when a processor embodiment according to the present invention is ‘on,’ there is another impedance due to the architecture of the present invention and its load. In practice, the aperture will affect the ‘on’ impedance. Hence, on the average, the input impedance looking into the circuitry of an embodiment of the present invention (i.e., its ports) is modified according to the present invention clock and TA. Impedance matching networks must take this into account.
EQ. (160) illustrates that the average impedance, av, is related to the voltage, V, divided by the average current flow, Iav, into a device, for example a processor according to an embodiment of the present invention. EQ. (160) indicates that for a processor according to an embodiment of the present invention the narrower TA and the less frequent a sample is acquired, the greater
av becomes.
To understand this, consider the fact that a 10th harmonic system according to an embodiment of the present invention operates with half as many samples as a 5th harmonic sample according to the present invention. Thus, according to EQ. (160), a 5th harmonic sample according to an embodiment of the present invention would typically possess a higher input/output impedance than that a 10th harmonic system according to the present invention. Of course, practical board and circuit parasitics will place limits on how much the impedance scaling properties of the present invention processor clock signals control the processor's overall input/output impedance.
As will be apparent to persons skilled in the relevant arts given the discussion herein, in embodiments, matching networks should be included at the ports of a processor according to the present invention to accommodate av, as measured by a typical network analyzer.
All signals can be represented by vectors in the complex signal plane. Previous sub-sections derived the result for down converting (or up converting) Si(t) in the transform domain via Si(ω). An I/Q modem embodiment of the present invention, however, was developed using a time domain analysis. This time domain analysis is repeated here and provides a complementary view to the previous sub-sections.
where Si(tk) is defined as the kth sample from the UFT transform such that Si(tk) is filtered over the kth interval, n(tk) is defined as the noise sample at the output of the kth present invention kernel interval such that it has been averaged by the present invention process over the interval, CIk is defined as the kth in phase gating waveform (the present invention clock), and CQk is defined as the kth quadrature phase gating waveform (the present invention clock).
The ‘goodness’ of Si(tk) and ni(tk) has been shown previously herein as related to the type of present invention processor used (e.g., matched filtering/correlating processor, finite time integrating processor, or RC processor). Each tk instant is the time tick corresponding to the averaging of input waveform energy over a TA (aperture) duration. It has been assumed that CIk and CQk are constant envelope and phase for the current analysis, although in general this is not required. Many different, interesting processors according to embodiments of the present invention can be constructed by manipulating the amplitudes and phases of the present invention clock.
CIk and CQk can be expanded as follows:
The above treatment is a Fourier series expansion of the present invention clocks where:
KΔ Arbitrary Gain Constant
TsΔ Aperture Time=fs−1
nΔ Harmonic Spectrum Harmonic Order
φΔ As phase shift angle usually selected as 90° (π/2) for orthogonal signaling Each term from CIk, CQk will down convert (or up convert). However, only the odd terms in the above formulation (for φ=π/2) will convert in quadrature. φ could be selected otherwise to utilize the even harmonics, but this is typically not done in practice.
For the case of down conversion, r(t) can be written as:
r(tk)=√{square root over (2)}A({tilde over (S)}iI(tk)cos(m·2πftkΘ)−{tilde over (S)}iQ(tk)sin(m·2πftk+Θ)+n(t)) EQ. (162)
After applying (CIk, CQk) and lowpass filtering, which in embodiments is inherent to the present invention process, the down converted components become:
S
0(tk)I=ASiI(tk)+ñIk EQ. (163)
S
0(tk)Q=ASiQ(tk)+ñQk EQ. (164)
where:
If the carrier is not perfectly coherent, a phase shift occurs as described in previous sub-section. The result presented above would modify to:
S
0(t)=(S0(t)I+jS0(t)Q)ejφ EQ. (165)
where φ is the phase shift. This is the same phase shift affect derived earlier as cos φ in the present invention transform. When there is a slight carrier offset then φ can be written as φ(t) and the I and Q outputs represent orthogonal, harmonically oscillating vectors super imposed on the desired signal output with a beat frequency proportional to:
f
error
Δnf
s
±m(fs±fΔ)=fs(n−m)+mfΔ EQ. (166)
fΔΔ as a slight frequency offset between the carrier and the present invention clock
This entire analysis could have been accomplished in the frequency domain as described herein, or it could have been formulated from the present invention kernel as:
S
0(t)=DIQ(Si(t)+n(t)) EQ. (167)
The recursive kernel DIQ is defined in sub-section 8 and the I/Q version is completed by superposition and phase shifting the quadrature kernel.
The previous equation for r(t) could be replaced with:
BB(t)={tilde over (S)}iI±{tilde over (S)}iQ where f=0 and Θ=π/4 and n(t)=0 EQ. (168)
BB(t) could be up converted by applying CI, CQ. The desired carrier then is the appropriate harmonic of CI, CQ whose energy is optimally extracted by a network matched to the desired carrier.
This sub-section introduces the concept of using a present invention core to modulate signals at RF according to embodiments of the invention. Although many specific modulator architectures are possible, which target individual signaling schemes such as AM, FM, PM, etc., the example architecture presented here is a vector signal modulator. Such a modulator can be used to create virtually every known useful waveform to encompass the whole of analog and digital communications applications, for “wired” or “wireless,” at radio frequency or intermediate frequency. In essence, a receiver process, which utilizes the present invention, may be reversed to create signals of interest at passband. Using I/Q waveforms at baseband, all points within the two dimensional complex signaling constellation may be synthesized when cores according to the present invention are excited by orthogonal sub-harmonic clocks and connected at their outputs with particular combining networks. A basic architecture that can be used is shown in
In
To illustrate this, if a passband waveform must be created at five times the frequency of the sub-harmonic clock then a baseline power for that harmonic extraction can be calculated for n=5. For the case of n=5, it is found that the 5th harmonic yields:
This component can be extracted from the Fourier series via a bandpass filter centered around fs. This component is a carrier at 5 times the sampling frequency.
This illustration can be extended to show the following:
This equation illustrates that a message signal may have been superposed on I and Ī such that both amplitude and phase are modulated, i.e., m(t) for amplitude and φ(t) for phase. In such cases, it should be noted that φ(t) is augmented modulo n while the amplitude modulation m(t) is scaled. The point of this illustration is that complex waveforms may be reconstructed from their Fourier series with multi-aperture processor combinations, according to the present invention.
In a practical system according to an embodiment of the present invention, parasitics, filtering, etc., may modify Ic(t). In many applications according to the present invention, charge injection properties of processors play a significant role. However, if the processors and the clock drive circuits according to embodiments of the present invention are matched then even the parasitics can be managed, particularly since unwanted distortions are removed by the final bandpass filter, which tends to completely reconstruct the waveform at passband.
Like the receiver embodiments of the present invention, which possess a lowpass information extraction and energy extraction impulse response, various transmitter embodiments of the present invention use a network to create a bandpass impulse response suitable for energy transfer and waveform reconstruction. In embodiments, the simplest reconstruction network is an L-C tank, which resonates at the desired carrier frequency N·fs=fc.
I/Q modulation receiver 19700 receives, down-converts, and demodulates a I/Q modulated RF input signal 19782 to an I baseband output signal 19784, and a Q baseband output signal 19786. I/Q modulated RF input signal comprises a first information signal and a second information signal that are I/Q modulated onto an RF carrier signal. I baseband output signal 19784 comprises the first baseband information signal. Q baseband output signal 19786 comprises the second baseband information signal.
Antenna 19772 receives I/Q modulated RF input signal 19782. I/Q modulated RF input signal 19782 is output by antenna 19772 and received by optional LNA 19718. When present, LNA 19718 amplifies I/Q modulated RF input signal 19782, and outputs amplified I/Q signal 19788.
First Processing module 19702 receives amplified I/Q signal 19788. First Processing module 19702 down-converts the I-phase signal portion of amplified input I/Q signal 19788 according to an I control signal 19790. First Processing module 19702 outputs an I output signal 19798.
In an embodiment, first Processing module 19702 comprises a first storage module 19724, a first UFT module 19726, and a first voltage reference 19728. In an embodiment, a switch contained within first UFT module 19726 opens and closes as a function of I control signal 19790. As a result of the opening and closing of this switch, which respectively couples and de-couples first storage module 19724 to and from first voltage reference 19728, a down-converted signal, referred to as I output signal 19798, results. First voltage reference 19728 may be any reference voltage, and is ground in some embodiments. I output signal 19798 is stored by first storage module 19724.
In an embodiment, first storage module 19724 comprises a first capacitor 19774. In addition to storing I output signal 19798, first capacitor 19774 reduces or prevents a DC offset voltage resulting from charge injection from appearing on I output signal 19798
I output signal 19798 is received by optional first filter 19704. When present, first filter 19704 is a high pass filter to at least filter I output signal 19798 to remove any carrier signal “bleed through”. In an embodiment, when present, first filter 19704 comprises a first resistor 19730, a first filter capacitor 19732, and a first filter voltage reference 19734. Preferably, first resistor 19730 is coupled between I output signal 19798 and a filtered I output signal 19707, and first filter capacitor 19732 is coupled between filtered I output signal 19707 and first filter voltage reference 19734. Alternately, first filter 19704 may comprise any other applicable filter configuration as would be understood by persons skilled in the relevant arts. First filter 19704 outputs filtered I output signal 19707.
Second Processing module 19706 receives amplified I/Q signal 19788. Second Processing module 19706 down-converts the inverted I-phase signal portion of amplified input I/Q signal 19788 according to an inverted I control signal 19792. Second Processing module 19706 outputs an inverted I output signal 19701.
In an embodiment, second Processing module 19706 comprises a second storage module 19736, a second UFT module 19738, and a second voltage reference 19740. In an embodiment, a switch contained within second UFT module 19738 opens and closes as a function of inverted I control signal 19792. As a result of the opening and closing of this switch, which respectively couples and de-couples second storage module 19736 to and from second voltage reference 19740, a down-converted signal, referred to as inverted I output signal 19701, results. Second voltage reference 19740 may be any reference voltage, and is preferably ground. Inverted I output signal 19701 is stored by second storage module 19736.
In an embodiment, second storage module 19736 comprises a second capacitor 19776. In addition to storing inverted I output signal 19701, second capacitor 19776 reduces or prevents a DC offset voltage resulting from above described charge injection from appearing on inverted I output signal 19701.
Inverted I output signal 19701 is received by optional second filter 19708. When present, second filter 19708 is a high pass filter to at least filter inverted I output signal 19701 to remove any carrier signal “bleed through”. In an embodiment, when present, second filter 19708 comprises a second resistor 19742, a second filter capacitor 19744, and a second filter voltage reference 19746. In an embodiment, second resistor 19742 is coupled between inverted I output signal 19701 and a filtered inverted I output signal 19709, and second filter capacitor 19744 is coupled between filtered inverted I output signal 19709 and second filter voltage reference 19746. Alternately, second filter 19708 may comprise any other applicable filter configuration as would be understood by persons skilled in the relevant arts. Second filter 19708 outputs filtered inverted I output signal 19709.
First differential amplifier 19720 receives filtered I output signal 19707 at its non-inverting input and receives filtered inverted I output signal 19709 at its inverting input. First differential amplifier 19720 subtracts filtered inverted I output signal 19709 from filtered I output signal 19707, amplifies the result, and outputs I baseband output signal 19784. Other suitable subtractor modules may be substituted for first differential amplifier 19720, and second differential amplifier 19722, as would be understood by persons skilled in the relevant arts from the teachings herein. Because filtered inverted I output signal 19709 is substantially equal to an inverted version of filtered I output signal 19707, I baseband output signal 19784 is substantially equal to filtered I output signal 19709, with its amplitude doubled. Furthermore, filtered I output signal 19707 and filtered inverted I output signal 19709 may comprise substantially equal noise and DC offset contributions of the same polarity from prior down-conversion circuitry, including first Processing module 19702 and second Processing module 19706, respectively. When first differential amplifier 19720 subtracts filtered inverted I output signal 19709 from filtered I output signal 19707, these noise and DC offset contributions substantially cancel each other.
Third Processing module 19710 receives amplified I/Q signal 19788. Third Processing module 19710 down-converts the Q-phase signal portion of amplified input I/Q signal 19788 according to an Q control signal 19794. Third Processing module 19710 outputs an Q output signal 19703.
In an embodiment, third Processing module 19710 comprises a third storage module 19748, a third UFT module 19750, and a third voltage reference 19752. In an embodiment, a switch contained within third UFT module 19750 opens and closes as a function of Q control signal 19794. As a result of the opening and closing of this switch, which respectively couples and de-couples third storage module 19748 to and from third voltage reference 19752, a down-converted signal, referred to as Q output signal 19703, results. Third voltage reference 19752 may be any reference voltage, and is preferably ground. Q output signal 19703 is stored by third storage module 19748.
In an embodiment, third storage module 19748 comprises a third capacitor 19778. In addition to storing Q output signal 19703, third capacitor 19778 reduces or prevents a DC offset voltage resulting from above described charge injection from appearing on Q output signal 19703.
Q output signal 19703 is received by optional third filter 19716. When present, third filter 19716 is a high pass filter to at least filter Q output signal 19703 to remove any carrier signal “bleed through”. In an embodiment, when present, third filter 19712 comprises a third resistor 19754, a third filter capacitor 19758, and a third filter voltage reference 19758. In an embodiment, third resistor 19754 is coupled between Q output signal 19703 and a filtered Q output signal 19711, and third filter capacitor 19756 is coupled between filtered Q output signal 19711 and third filter voltage reference 19758. Alternately, third filter 19712 may comprise any other applicable filter configuration as would be understood by persons skilled in the relevant arts. Third filter 19712 outputs filtered Q output signal 19711.
Fourth Processing module 19714 receives amplified I/Q signal 19788. Fourth Processing module 19714 down-converts the inverted Q-phase signal portion of amplified input I/Q signal 19788 according to an inverted Q control signal 19796. Fourth Processing module 19714 outputs an inverted Q output signal 19705.
In an embodiment, fourth Processing module 19714 comprises a fourth storage module 19760, a fourth UFT module 19762, and a fourth voltage reference 19764. In an embodiment, a switch contained within fourth UFT module 19762 opens and closes as a function of inverted Q control signal 19796. As a result of the opening and closing of this switch, which respectively couples and de-couples fourth storage module 19760 to and from fourth voltage reference 19764, a down-converted signal, referred to as inverted Q output signal 19705, results. Fourth voltage reference 19764 may be any reference voltage, and is preferably ground. Inverted Q output signal 19705 is stored by fourth storage module 19760.
In an embodiment, fourth storage module 19760 comprises a fourth capacitor 19780. In addition to storing inverted Q output signal 19705, fourth capacitor 19780 reduces or prevents a DC offset voltage resulting from above described charge injection from appearing on inverted Q output signal 19705.
Inverted Q output signal 19705 is received by optional fourth filter 19716. When present, fourth filter 19716 is a high pass filter to at least filter inverted Q output signal 19705 to remove any carrier signal “bleed through”. In an embodiment, when present, fourth filter 19716 comprises a fourth resistor 19766, a fourth filter capacitor 19768, and a fourth filter voltage reference 19770. In an embodiment, fourth resistor 19766 is coupled between inverted Q output signal 19705 and a filtered inverted Q output signal 19713, and fourth filter capacitor 19768 is coupled between filtered inverted Q output signal 19713 and fourth filter voltage reference 19770. Alternately, fourth filter 19716 may comprise any other applicable filter configuration as would be understood by persons skilled in the relevant arts. Fourth filter 19716 outputs filtered inverted Q output signal 19713.
Second differential amplifier 19722 receives filtered Q output signal 19711 at its non-inverting input and receives filtered inverted Q output signal 19713 at its inverting input. Second differential amplifier 19722 subtracts filtered inverted Q output signal 19713 from filtered Q output signal 19711, amplifies the result, and outputs Q baseband output signal 19786. Because filtered inverted Q output signal 19713 is substantially equal to an inverted version of filtered Q output signal 19711, Q baseband output signal 19786 is substantially equal to filtered Q output signal 19713, with its amplitude doubled. Furthermore, filtered Q output signal 19711 and filtered inverted Q output signal 19713 may comprise substantially equal noise and DC offset contributions of the same polarity from prior down-conversion circuitry, including third Processing module 19710 and fourth Processing module 19714, respectively. When second differential amplifier 19722 subtracts filtered inverted Q output signal 19713 from filtered Q output signal 19711, these noise and DC offset contributions substantially cancel each other.
I/Q modulation control signal generator 19800 comprises a local oscillator 19802, a first divide-by-two module 19804, a 180 degree phase shifter 19806, a second divide-by-two module 19808, a first pulse generator 19810, a second pulse generator 19812, a third pulse generator 19814, and a fourth pulse generator 19816.
Local oscillator 19802 outputs an oscillating signal 19818.
First divide-by-two module 19804 receives oscillating signal 19818, divides oscillating signal 19818 by two, and outputs a half frequency LO signal 19820 and a half frequency inverted LO signal 19826.
180 degree phase shifter 19806 receives oscillating signal 19818, shifts the phase of oscillating signal 19818 by 180 degrees, and outputs phase shifted LO signal 19822. 180 degree phase shifter 19806 may be implemented in circuit logic, hardware, software, or any combination thereof, as would be known by persons skilled in the relevant arts. In alternative embodiments, other amounts of phase shift may be used.
Second divide-by two module 19808 receives phase shifted LO signal 19822, divides phase shifted LO signal 19822 by two, and outputs a half frequency phase shifted LO signal 19824 and a half frequency inverted phase shifted LO signal 19828.
First pulse generator 19810 receives half frequency LO signal 19820, generates an output pulse whenever a rising edge is received on half frequency LO signal 19820, and outputs I control signal 19790.
Second pulse generator 19812 receives half frequency inverted LO signal 19826, generates an output pulse whenever a rising edge is received on half frequency inverted LO signal 19826, and outputs inverted I control signal 19792.
Third pulse generator 19814 receives half frequency phase shifted LO signal 19824, generates an output pulse whenever a rising edge is received on half frequency phase shifted LO signal 19824, and outputs Q control signal 19794.
Fourth pulse generator 19816 receives half frequency inverted phase shifted LO signal 19828, generates an output pulse whenever a rising edge is received on half frequency inverted phase shifted LO signal 19828, and outputs inverted Q control signal 19796.
In an embodiment, control signals 19790, 19792, 19794 and 19796 output pulses having a width equal to one-half of a period of I/Q modulated RF input signal 19782. The invention, however, is not limited to these pulse widths, and control signals 19790, 19792, 19794, and 19796 may comprise pulse widths of any fraction of, or multiple and fraction of, a period of I/Q modulated RF input signal 19782. Also, other circuits for generating control signals 19790, 19792, 19794, and 19796 will be apparent to persons skilled in the relevant arts based on the herein teachings.
First, second, third, and fourth pulse generators 19810, 19812, 19814, and 19816 may be implemented in circuit logic, hardware, software, or any combination thereof, as would be known by persons skilled in the relevant arts.
As shown in
For example,
As
It should be understood that the above control signal generator circuit example is provided for illustrative purposes only. The invention is not limited to these embodiments. Alternative embodiments (including equivalents, extensions, variations, deviations, etc., of the embodiments described herein) for I/Q modulation control signal generator 19800 will be apparent to persons skilled in the relevant arts from the teachings herein, and are within the scope of the present invention.
According to embodiments of the invention, the amplitude level of the down-converted signal can be controlled by modifying the aperture of the control signal that controls the switch module. Consider
Some embodiments may include a control mechanism to enable manual control of aperture T, and thus manual control of the amplitude level of the down-converted signal. Other embodiments may include automatic or semi-automatic control modules to enable automatic or semi-automatic control of aperture T, and thus automatic or semi-automatic control of the amplitude level of the down-converted signal. Such embodiments are herein referred to (without limitation) as automatic gain control (AGC) embodiments. Other embodiments include a combination of manual and automatic control of aperture T.
Additional aspects/embodiments of the invention are considered in this section.
In one embodiment of the present invention there is provided a method of transmitting information between a transmitter and a receiver comprising the steps of transmitting a first series of signals each having a known period from the transmitter at a known first repetition rate; sampling by the receiver each signal in the first series of signals a single time and for a known time interval the sampling of the first series of signals being at a second repetition rate that is a rate different from the first repetition rate by a known amount; and generating by the receiver an output signal indicative of the signal levels sampled in step B and having a period longer than the known period of a transmitted signal.
In another embodiment of the invention there is provided a communication system comprising a transmitter means for transmitting a first series of signals of known period at a known first repetition rate, a receiver means for receiving the first series of signals, the receiver means including sampling means for sampling the signal level of each signal first series of signals for a known time interval at a known second repetition rate, the second repetition rate being different from the first repetition rate by a known amount as established by the receiver means. The receiver means includes first circuit means for generating a first receiver output signal indicative of the signal levels sampled and having a period longer than one signal of the first series of signals. The transmitter means includes an oscillator for generating an oscillator output signal at the first repetition rate, switch means for receiving the oscillator output signal and for selectively passing the oscillator output signal, waveform generating means for receiving the oscillator output signal for generating a waveform generator output signal having a time domain and frequency domain established by the waveform generating means.
The embodiment of the invention described herein involves a single or multi-user communications system that utilizes coherent signals to enhance the system performance over conventional radio frequency schemes while reducing cost and complexity. The design allows direct conversion of radio frequencies into baseband components for processing and provides a high level of rejection for signals that are not related to a known or controlled slew rate between the transmitter and receiver timing oscillators. The system can be designed to take advantage of broadband techniques that further increase its reliability and permit a high user density within a given area. The technique employed allows the system to be configured as a separate transmitter-receiver pair or a transceiver.
An objective of the present system is to provide a new communication technique that can be applied to both narrow and wide band systems. In its most robust form, all of the advantages of wide band communications are an inherent part of the system and the invention does not require complicated and costly circuitry as found in conventional wide band designs. The communications system utilizes coherent signals to send and receive information and consists of a transmitter and a receiver in its simplest form. The receiver contains circuitry to turn its radio frequency input on and off in a known relationship in time to the transmitted signal. This is accomplished by allowing the transmitter timing oscillator and the receiver timing oscillator to operate at different but known frequencies to create a known slew rate between the oscillators. If the slew rate is small compared to the timing oscillator frequencies, the transmitted waveform will appear stable in time, i.e., coherent (moving at the known slew rate) to the receiver's switched input. The transmitted waveform is the only waveform that will appear stable in time to the receiver and thus the receiver's input can be averaged to achieve the desired level filtering of unwanted signals. This methodology makes the system extremely selective without complicated filters and complex encoding and decoding schemes and allows the direct conversion of radio frequency energy from an antenna or cable to baseband frequencies with a minimum number of standard components further reducing cost and complexity. The transmitted waveform can be a constant carrier (narrowband), a controlled pulse (wideband and ultra-wideband) or a combination of both such as a dampened sinusoidal wave and or any arbitrary periodic waveform thus the system can be designed to meet virtually any bandwidth requirement. Simple standard modulation and demodulation techniques such as AM and Pulse Width Modulation can be easily applied to the system.
Depending on the system requirements such as the rate of information transfer, the process gain, and the intended use, there are multiple preferred embodiments of the invention. The embodiment discussed herein will be the amplitude and pulse width modulated system. It is one of the simplest implementations of the technology and has many common components with the subsequent systems. A amplitude modulated transmitter consists of a Transmitter Timing Oscillator, a Multiplier, a Waveform Generator, and an Optional Amplifier. The Transmitter Timing Oscillator frequency can be determined by a number of resonate circuits including an inductor and capacitor, a ceramic resonator, a SAW resonator, or a crystal. The output waveform is sinusoidal, although a squarewave oscillator would produce identical system performance.
The Multiplier component multiplies the Transmitter Timing Oscillator output signal by 0 or 1 or other constants, K1 and K2, to switch the oscillator output on and off to the Waveform Generator. In this embodiment, the information input can be digital data or analog data in the form of pulse width modulation. The Multiplier allows the Transmitter Timing Oscillator output to be present at the Waveform Generator input when the information input is above a predetermined value. In this state the transmitter will produce an output waveform. When the information input is below a predetermined value, there is no input to the Waveform Generator and thus there will be no transmitter output waveform. The output of the Waveform Generator determines the system's bandwidth in the frequency domain and consequently the number of users, process gain immunity to interference and overall reliability), the level of emissions on any given frequency, and the antenna or cable requirements. The Waveform Generator in this example creates a one cycle pulse output which produces an ultra-wideband signal in the frequency domain. An optional power Amplifier stage boosts the output of the Waveform Generator to a desired power level.
With reference now to the drawings, the amplitude and pulse width modulated transmitter in accord with the present invention is depicted at numeral 13000 in
The Receiver Timing Oscillator 13510 is connected to the Waveform Generator 13508 which shapes the oscillator signal into the appropriate output to control the amount of the time that the RF switch 13506 is on and off. The on-time of the RF switch 13506 should be less than ½ of a cycle ( 1/10 of a cycle is preferred) or in the case of a single pulse, no wider than the pulse width of the transmitted waveform or the signal gain of the system will be reduced. Examples are illustrated in Table A1. Therefore the output of the Waveform Generator 13508 is a pulse of the appropriate width that occurs once per cycle of the receiver timing oscillator 13510. The output 13604 of the Waveform Generator is shown in
The RF Switch/Integrator 13506 samples the RF signal 13606 shown in
In an embodiment of the present invention, the gating or sampling rate of the receiver 13500 is 300 Hz higher than the 25 MHZ transmission rate from the transmitter 13000. Alternatively, the sampling rate could be less than the transmission rate. The difference in repetition rates between the transmitter 13000 and receiver 13500, the “slew rate,” is 300 Hz and results in a controlled drift of the sampling pulses over the transmitted pulse which thus appears “stable” in time to the receiver 13500. With reference now to
Decode Circuitry 13514 extracts the information contained in the transmitted signal and includes a Rectifier that rectifies signal 13608 or 13610 to provide signal 13612 shown in
In the illustrated embodiment, the signal 13616 at “F” has a period of 83.33 usec, a frequency of 12 KHz and it is produced once every 3.3 msec for a 300 Hz slew rate. Stated another way, the system is converting a 1 gigahertz transmitted signal into an 83.33 microsecond signal.
Accordingly, the series of RF pulses 13210 that are transmitted during the presence of an “on” signal at the information input gate 13102 are used to reconstruct the information input signal 13204 by sampling the series of pulses at the receiver 13500. The system is designed to provide an adequate number of RF inputs 13606 to allow for signal reconstruction.
An optional Amplifier/Filter stage or stages 13504 and 13512 may be included to provide additional receiver sensitivity, bandwidth control or signal conditioning for the Decode Circuitry 13514. Choosing an appropriate time base multiplier will result in a signal at the output of the Integrator 13506 that can be amplified and filtered with operational amplifiers rather than RF amplifiers with a resultant simplification of the design process. The signal 13610 shown in
The present invention provides, among other things, the following architectural features:
optimal baseband signal to noise ratio regardless of modulation (programmable RF matched filter);
exceptional linearity per milliwatt consumed;
easily integrated into bulk C-MOS (small size/low cost, high level of integration);
fundamental or sub-harmonic operation (does not change conversion efficiency);
transmit function provides frequency multiplication and signal gain; and
optimal power transfer into a scalable output impedance (independent of device voltage or current);
The present invention provides simultaneous solutions for two domains: power sampling and matched filtering. A conventional sampler is a voltage sampling device, and does not substantially affect the input signal. A power sampler according to the present invention attempts to take as much power from the input to construct the output, and does not necessarily preserve the input signal.
The present invention out-performs a theoretically perfect impulse sampler. The performance of a practical implementation of the present invention exceeds the performance of a practical implementation of an impulse sampler. The present invention is easily implemented (does not require impulse circuitry).
The present invention provides exceptional linearity per milliwatt. For example, rail to rail dynamic range is possible with minimal increase in power. In an example integrated circuit embodiment, the present invention provides +55 dmb IP2, +15 dbm IP3, @ 3.3V, 4.4 ma, −15 dmb LO. GSM system requirements are +22 dbm IP2, −10.5 dmb IP3. CDMA system requirements are +50 dmb IP2, +10 dbm IP3.
In an embodiment of the present invention, output impedance is scalable to facilitate a low system noise figure. In an embodiment, changes in output impedance do not affect power consumption.
In an embodiment, the present invention enables a high level of integration in bulk C-MOS. Other features include:
small footprint;
no multiplier circuits (no device matching or balancing transistors);
transmit and receive filters at baseband;
low frequency synthesizers;
DC offset solutions;
Referring to
In an embodiment, LO signal 21806 runs at a sub-harmonic. Gilbert cells lose efficiency when run at a sub-harmonic, as compared to the receiver of the present invention.
Single-switch, differential input, differential output receiver embodiments according to the present invention, are discussed in further detail elsewhere herein.
architecturally reduces re-radiation;
Referring to
Referring to
Receiver embodiments, according to the present invention, for reducing or eliminating circuit re-radiation, such as receiver 21814, are discussed in further detail elsewhere herein.
inherent noise rejection; and
lower cost.
Sub-harmonic operation is preferred for many direct down-conversion implementations because it tends to avoid oscillators and/or signals near the desired operating frequency.
Conversion efficiency is generally constant regardless of the sub-harmonic. Sub-harmonic operation enables micro power receiver designs.
A transmit function in accordance with the present invention provides frequency multiplication and signal gain. For example, a 900 MHz design example (0.35μ CMOS) embodiment features −15 dbm 180 MHz LO, 0 dbm 900 MHz I/O output, 5 VDC, 5 ma. A 2400 MHz design example (0.35μ CMOS) embodiment features −15 dbm 800 MHz LO, −6 dbm 2.4 GHz I/O output, 5 VDC, 16 ma.
A transmit function in accordance with the present invention also provides direct up-conversion (true zero IF).
A non-negligible aperture, as taught herein, substantially preserves amplitude and phase information, but not necessarily the carrier signal. A general concept is to under-sample the carrier while over sampling the information.
The present invention transfers optimum energy. Example embodiments have been presented herein, including DC examples and carrier half cycle examples.
With regard to input bandwidth, optimum energy transfer generally occurs every n+½ cycle. Output bandwidth is generally a function of the LO.
A universal frequency down-converter (UDF), in accordance with the invention, can be designed to provides, among other things, the following features:
Complimentary FET switch implementations of the invention provide, among other things, increased dynamic range (lower Rdson—increased conversion efficiency, higher IIP2, IIP3, minimal current increase (+CMOS inverter), and lower re-radiation (charge cancellation). For example, refer to
Differential configuration implementations of the invention provide, among other things, DC off-set advantages, lower re-radiation, input and output common mode rejection, and minimal current increase. For example, refer to
Clock spreading aspects of the invention provide, among other things, lower re-radiation, DC off-set advantages, and flicker noise advantages. For example, refer to
The invention provides, among other things, optimization of signal to noise ratio subject to maximum energy transfer given a controlled aperture, and maximum energy transfer while preserving information. The invention also provides bandpass wave form auto sampling and pulse energy accumulation
Pulse width can be optimized for a frequency of interest. Generally, pulse width is n plus ½ cycles of a desired input frequency. Generally, in CMOS implementations of the invention, pulse width variation across process variations and temperature of interest is less than +/−16 percent.
Conventional heterodyne systems, in contrast to the present invention, are relatively complex, require multiple RF synthesizers, require management of various electromagnetic modes (shield, etc.), require significant inter-modulation management, and require a myriad of technologies that do not easily integrate onto integrated circuits.
High quality mobile wireless devices have not been implemented via zero IF because of the high power requirements for the first conversion in order to obtain necessary dynamic range, the high level of LO required (LO re-radiation), adjacent channel interference rejection filtering, transmitter modulation filtering, transmitter LO leakage, and limitations on RF synthesizer performance and technology.
The complex phasor notation of a harmonic signal is known from Euler's equation, shown here as equation 172.
S(t)=e−j(ω
Suppose that φ is also some function of time φ(t). φ(t) represents phase noise or some other phase perturbation of the waveform. Furthermore, suppose that φ(t) and −φ(t) can be derived and manipulated. Then if follows that the multiplication of S1(t) and S2(t) will yield equation 173.
S(t)=S1(t)·S2(t)=e−j(ω
Thus, the phase noise φ(t) can be canceled. Trigonometric identities verify the same result except for an additional term at DC. This can be implemented with, for example, a four-quadrant version of the invention.
In an embodiment two clocks are utilized for phase noise cancellation of odd and even order harmonics by cascading stages. A four quadrant implementation of the invention can be utilized to eliminate the multiplier illustrated in
In an embodiment, parallel receivers and transmitters are implemented using single pole, double throw, triple throw, etc., implementations of the invention.
A multiple throw implementation of the invention can also be utilized. In this embodiment, many frequency conversion options at multiple rates can be performed in parallel or serial. This can be implemented for multiple receive functions, multi-band radios, multi-rate filters, etc.
Multiple apertures can be utilized to accomplish a variety of effects. For example,
Similarly, the number of apertures can be extended with associated bipolar weighting to form a variety of impulse responses and to perform filtering at RF.
The present invention can be utilized to implement maximal ratio post detection combiners, equal gain post detection combiners, and selectors.
The present invention can serve as a quadrature down converter and as a unit delay function. In an example of such an implementation, the unit delay function is implemented with a decimated clock at baseband.
Example embodiments of the methods, systems, and components of the present invention have been described herein. As noted elsewhere, these example embodiments have been described for illustrative purposes only, and are not limiting. Other embodiments are possible and are covered by the invention. Such other embodiments include but are not limited to hardware, software, and software/hardware implementations of the methods, systems, and components of the invention. Such other embodiments will be apparent to persons skilled in the relevant art(s) based on the teachings contained herein. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.
The present application is a continuation of pending U.S. application “Method and System for Down-Converting an Electromagnetic Signal, and Transforms for Same, and Aperture Relationships”, Ser. No. 14/172,392, filed Feb. 4, 2014, which is a continuation of U.S. application “Method and System for Down-Converting an Electromagnetic Signal, and Transforms for Same, and Aperture Relationships”, Ser. No. 13/549,213, filed Jul. 13, 2012, (now U.S. Pat. No. 8,660,513) which is a continuation of “Method and System for Down-Converting an Electromagnetic Signal and Transforms for the Same, and Aperture Relationships”, Ser. No. 12/976,839, filed Dec. 22, 2010 (now U.S. Pat. No. 8,340,618), which is a continuation of U.S. application “Method and System for Down-Converting an Electromagnetic Signal, and Transforms for Same, and Aperture Relationships,” Ser. No. 12/349,802, filed Jan. 7, 2009 (now U.S. Pat. No. 7,865,177), which is a divisional application of U.S. application “Method and System for Down-Converting an Electromagnetic Signal, and Transforms for Same, and Aperture Relationships,” Ser. No. 09/550,644, filed Apr. 14, 2000 (now U.S. Pat. No. 7,515,896), which is a continuation-in-part application of U.S. application “Method and System for Down-Converting an Electromagnetic Signal Including Resonant Structures for Enhanced Energy Transfer,” Ser. No. 09/293,342, filed Apr. 16, 1999 (now U.S. Pat. No. 6,687,493), which is a continuation-in-part application of U.S. application “Method and System for Down-Converting Electromagnetic Signals,” Ser. No. 09/176,022, filed Oct. 21, 1998 (now U.S. Pat. No. 6,061,551), each of which is herein incorporated by reference in their entireties. The following applications of common assignee are related to the present application, and are herein incorporated by reference in their entireties: “Method and System for Frequency Up-Conversion,” Ser. No. 09/176,154, filed Oct. 21, 1998 (now U.S. Pat. No. 6,091,940); “Method and System for Ensuring Reception of a Communications Signal,” Ser. No. 09/176,415, filed Oct. 21, 1998 (now U.S. Pat. No. 6,061,555); “Integrated Frequency Translation and Selectivity,” Ser. No. 09/175,966, filed Oct. 21, 1998 (now U.S. Pat. No. 6,049,706); “Universal Frequency Translation, and Applications of Same,” Ser. No. 09/176,027, filed Oct. 21, 1998 (now abandoned); “Method and System for Down-Converting Electromagnetic Signals Having Optimized Switch Structures,” Ser. No. 09/293,095, filed Apr. 16, 1999 (now U.S. Pat. No. 6,580,902); “Method and System for Frequency Up-Conversion with a Variety of Transmitter Configurations,” Ser. No. 09/293,580, filed Apr. 16, 1999 (U.S. Pat. No. 6,542,722); and “Integrated Frequency Translation and Selectivity with a Variety of Filter Embodiments,” Ser. No. 09/293,283, filed Apr. 16, 1999 (now U.S. Pat. No. 6,560,301).
| Number | Date | Country | |
|---|---|---|---|
| Parent | 09550644 | Apr 2000 | US |
| Child | 12349802 | US |
| Number | Date | Country | |
|---|---|---|---|
| Parent | 14172392 | Feb 2014 | US |
| Child | 14639296 | US | |
| Parent | 13549213 | Jul 2012 | US |
| Child | 14172392 | US | |
| Parent | 12976839 | Dec 2010 | US |
| Child | 13549213 | US | |
| Parent | 12349802 | Jan 2009 | US |
| Child | 12976839 | US |
| Number | Date | Country | |
|---|---|---|---|
| Parent | 09293342 | Apr 1999 | US |
| Child | 09550644 | US | |
| Parent | 09176022 | Oct 1998 | US |
| Child | 09293342 | US |
