Patent Application
20150303577
1. Field of the Invention
The present invention relates to a MIMO (Multiple Input Multiple Output) antenna including a plurality of antenna elements, and a wireless device. 2. Description of the Related Art
In the field of communication devices such as mobile terminals where an adequate distance cannot be secured between antenna elements, there is a demand for a MIMO antenna with a high antenna gain and a low correlation coefficient between antenna elements in order to ensure good MIMO effects. The MIMO antenna is a multi-antenna that is capable of multiple-input and multiple-output operations at a predetermined frequency using a plurality of antenna elements. For example, Japanese Laid-Open Patent Publication No. 2010-130115 discloses a MIMO antenna including a plurality of monopole antenna elements that utilize a ground plane as a MIMO antenna including a plurality of antennal elements.
In MIMO antennas, the correlation coefficient between antenna elements has to be lowered. However, in MIMO antennas that use monopole antenna elements, the correlation coefficient cannot be lowered unless the monopole antenna elements are released from the ground plane. When the monopole antenna elements are released from the ground plane, the space required for installing the antenna elements is expanded, and as such, it is difficult to reduce the installation space of the antenna elements and lower the correlation coefficient between the antenna elements at the same time.
An aspect of the present invention is directed to providing a MIMO antenna and a wireless device that can reduce the installation space of antenna elements and lower the correlation coefficient between the antenna elements at the same time.
According to one embodiment of the present invention, a MIMO antenna is provided that includes a ground plane, and a plurality of dipole antenna elements that are arranged in the vicinity of the ground plane. Each of the plurality of dipole antenna elements includes a radiating element including a conductor portion extending along an outer edge portion of the ground plane, and a feeding portion that feeds the radiating element.
In the following, embodiments of the present invention will be described with reference to the accompanying drawings.
The ground plane 70 is, for example, a ground region including at least one corner portion 73, an outer edge portion 71 linearly extending from the corner portion 73 in the Y-axis direction, and an outer edge portion 72 linearly extending in the X-axis direction from the corner portion 73. Although the extending direction of the outer edge portion 71 and the extending direction of the outer edge portion 72 are preferably arranged to be orthogonal, the intersecting angle of the extending directions may deviate within a range that would not impair the effects of the present invention. For example, the intersecting angle may preferably be greater than or equal to 70° and less than or equal to 110°, and more preferably greater than or equal to 80° and less than for equal to 100°.
The dipole antenna elements 10 and 20 are arranged in the vicinity of the corner portion 73 of the ground plane 70, for example. The dipole antenna element 10 is arranged along the outer edge portion 71, and may be spaced apart from the outer edge portion 71 by a predetermined distance D1 in the X-axis direction and extend parallel to the outer edge portion 71 in the Y-axis direction, for example. The dipole antenna element 20 is arranged along the outer edge 72, and may be spaced apart from the outer edge portion 72 by the predetermined distance D1 in the Y-axis direction and extend parallel to the outer edge portion 72 in the X-axis direction, for example. In
Each of the plurality of dipole antenna elements may include a radiating element having a conductor portion extending in a direction perpendicular to the extending direction of the conductor portion of another dipole antenna element of the plurality of dipole antenna elements, for example. The dipole antenna element 10 includes a radiating element 11, and the dipole antenna element 20 includes a radiating element 21. The radiating element 11 is an antenna conductor that functions as an antenna having a feeding portion 16 as a feeding point, and the radiating element 21 is an antenna conductor that functions as an antenna having a feeding portion 26 as a feeding point.
The radiating element 11 of the dipole antenna element 10 includes a conductor portion 12 and a conductor portion 13 that extend in a direction perpendicular to the extending direction of a conductor portion 22 or a conductor portion 23 of the radiating element 21 of the other dipole antenna element 20 that is different from the dipole antenna element 10. The conductor portions 12 and 13 are linear antenna conductor portions that are arranged along the outer edge portion 71, and may be spaced apart from the outer edge portion 71 by the predetermined distance D1 in the X-axis direction and extend parallel to the outer edge portion 71 in the Y-axis direction, for example. By arranging the radiating element 11 to have the conductor portions 12 and 13 along the outer edge portion 71, the directivity of the MIMO antenna 1 may be easily controlled, for example.
The radiating element 21 of the dipole antenna element 20 includes the conductor portion 22 and the conductor portion 23 that extend in a direction perpendicular to the extending direction of the conductor portion 12 or the conductor portion 13 of the radiating element 11 of the other dipole antenna element 10 that is different from the dipole antenna element 20. The conductor portions 22 and 23 are linear antenna conductor portions that are arranged along the outer edge portion 72, and may be spaced apart from the outer edge portion 72 by the predetermined distance D1 in the X-axis direction and extend parallel to the outer edge portion 72 in the Y-axis direction, for example. By arranging the radiating element 21 to have the conductor portions 22 and 23 along the outer edge portion 72, the directivity of the MIMO antenna 1 may be easily controlled, for example.
The radiating elements 11 and 21 may be mounted to a dielectric substrate 80, and may be placed on a surface of the dielectric substrate 80 or installed inside the dielectric substrate 80, for example. The dielectric substrate 80 may be a resin substrate, for example. However, a dielectric material other than resin such as glass, glass ceramic, or LTCC (Low Temperature Co-Fired Ceramics) may be used as well. The ground plane 70 may be a region formed at the dielectric substrate 80 or a region formed at a separate member from the dielectric substrate 80. In the illustrated case, the radiating elements 11 and 21 are arranged at the same surface of the dielectric substrate 80. However, the radiating elements 11 and 21 may be arranged at different layers in the Z-axis direction. Also, the radiating element 11 or the radiating element 21 may be arranged at the same layer in the Z-axis direction as the ground plane 70, or the radiating elements 11 and 21 may be arranged at different layers from the ground plane 70.
The dipole antenna element 10 includes the feeding portion 16 for feeding the radiating element 11. The feeding portion 16 is a feeding point that is inserted into a conductor portion between one end portion 14 and another end portion 15 of the radiating element 11.
In
To facilitate matching of the dipole antenna element 10, the feeding portion 16 may be a feeding point located at a region between the end portion 14 and the end portion 15 having higher impedance than the central portion 90. The impedance of the radiating element 11 becomes higher as the distance away from the central portion 90 and toward the end portion 14 or the end portion 15 of the radiating element 11 increases, and in
The dipole antenna element 20 includes a feeding portion 26 for feeding the radiating element 21. The feeding portion 26 is a feeding point that is inserted into a conductor portion between one end portion 24 and another end portion 25 of the radiating element 21.
In
To facilitate matching of the dipole antenna element 20, the feeding portion 26 may be a feeding point located at a region between the end portion 24 and the end portion 25 having higher impedance than the central portion 90. The impedance of the radiating element 21 becomes higher as the distance away from the central portion 90 and toward the end portion 24 or the end portion 25 of the radiating element 21 increases, and in
The feeding portion 16 and the feeding portion 26 are located at regions that are shifted from the central portions 90 of the radiating elements 11 and 21 in directions approaching each other. In this way, matching of the dipole antenna elements 10 and 20 may be facilitated, and transmission lines respectively connected to the feeding portions 16 and 26 may be brought closer to each other such that the space required for installing the dipole antenna elements 10 and 20 may be easily reduced.
As a method for feeding the feeding portion 16 and the feeding portion 26, for example, unbalanced lines such as coaxial cables may be directly connected to the radiating elements 11 and 21, or the lines may be converted into balanced lines via baluns and directly connected to the radiating elements 11 and 21, for example. Also, in a case where the radiating elements 11 and 21 are formed on a dielectric substrate having a ground plane, they may be connected by planar transmission lines, for example. Further, metal pins from another dielectric substrate that is different from the dielectric substrate at which the radiating elements 11 and 21 are formed may be connected to the conductor portions of the radiating elements 11 and 21, for example. In this way, a suitable method for feeding the dipole antenna elements 10 and 20 may be selected according to the implementation environment.
The dipole antenna elements 30 and 40 are arranged in the vicinity of the corner portion 73 of the ground plane 70, for example. The dipole antenna element 30 includes a radiating element 31 as a radiating element having a conductor portion extending in a direction perpendicular to the extending direction of a conductor portion of the dipole antenna element 40. The dipole antenna element 40 includes a radiating element 41 as a radiating element having the conductor portion extending perpendicular to the extending direction of the conductor portion of the dipole antenna element 30. Note that the dipole antenna element 40 has a configuration substantially similar to that of the dipole antenna element 30, and as such, the following descriptions of the dipole antenna element 30 apply to the dipole antenna element 40.
The radiating element 31 of the dipole antenna element 30 includes a conductor portion extending perpendicular to the extending direction of the conductor portion of the radiating element 41 of the other dipole antenna element 40. The conductor portion of the radiating element 31 is a linear antenna conductor portion arranged along the outer edge portion 71, and may be spaced apart from the outer edge portion 71 by a predetermined distance D1 in the X-axis direction and extend parallel to the outer edge portion 71 in the Y-axis direction, for example. By arranging the radiating element 31 to have the conductor portion along the outer edge portion 71, the directivity of the MIMO antenna 2 may be easily controlled, for example. Also, in a case where the radiating element 31 and the outer edge portion 71 are spaced apart in both the X-axis direction and a thickness direction (Z-axis direction), the shortest distance D2 between the radiating element 31 and the outer edge portion 71 corresponds to the distance of a straight line connecting sections of the radiating element 31 and the outer edge portion 71 that are closest to each other.
The dipole antenna element 30 includes a feeding portion 36 for feeding the radiating element 31, and a feeding element 37 corresponding to a conductor that is spaced apart from the radiating element 31 by a predetermined distance in the Z-axis direction. Note that in
The feeding element 37 and the radiating element 31 are spaced apart by a distance that enables electromagnetic field coupling of these elements. Non-contact feeding of the radiating element 31 at the feeding portion 36 via the feeding element 37 may be implemented by electromagnetic field coupling. By being fed in the above-described manner, the radiating element 31 may function as a radiating conductor of an antenna. As illustrated in
Electromagnetic field coupling refers to coupling that utilizes a resonance phenomenon of an electromagnetic field as disclosed, for example, in the following non-patent literature: A. Kurs et. al., “Wireless Power Transfer via Strongly Coupled Magnetic Resonances,” Science Express, Vol. 317, No. 5834, pp. 83-86, July 2007. Electromagnetic field coupling, also referred to as “electromagnetic field resonance coupling” or “electromagnetic field resonant coupling,” is a technique in which resonators that resonate at the same frequency are brought close to each other, one of the resonators is caused to resonate to generate a near field (non-radiation field area) between the resonators, and energy is transmitted to another one of the resonators via coupling at the near field. Also, electromagnetic field coupling refers to coupling via an electric field and a magnetic field at a high frequency excluding electrostatic capacitive coupling and electromagnetic induction coupling. Here, “excluding electrostatic capacitive coupling and electromagnetic induction coupling” does not necessarily mean electrostatic capacitive coupling and electromagnetic induction coupling are completely eliminated, but indicates that their influence is negligible. A medium between the feeding element 37 and the radiating element 31 may be air or a dielectric material such as glass or resin. It is preferable to not place a conductor material such as a ground plane or a display between the feeding element 37 and the radiating element 31.
By coupling the feeding element 37 and the radiating element 31 through electromagnetic field coupling, a durable structure that is resistant to impact may be obtained. That is, by utilizing electromagnetic field coupling, feeding of the radiating element 31 may be implemented using the feeding element 37 without requiring physical contact between the radiating element 31 and the feeding element 37, and thus, a durable structure that is resistant to impact may be obtained as compared to a contact type feeding mechanism that requires physical contact between the feeding element and the radiating element.
By coupling the feeding element 37 and the radiating element 31 through electromagnetic field coupling, non-contact feeding may be easily implemented. That is, by utilizing electromagnetic field coupling, feeding of the radiating element 31 may be implemented using the feeding element 37 without requiring physical contact between the radiating element 31 and the feeding element 37, and thus, feeding may be performed with a simpler configuration as compared to a contact-type feeding mechanism requiring physical contact. Also, by utilizing electromagnetic field coupling, feeding of the radiating element 31 using the feeding element 37 may be implemented without requiring extra components such as a capacitor plate, and thus, feeding may be implemented with a simpler configuration as compared to feeding using electrostatic capacitive coupling.
Also, as compared with feeding using electrostatic capacitive coupling, when feeding using electromagnetic field coupling is implemented, the total efficiency (antenna gain) of the radiating element 31 may be less likely to decrease even if the distance between the feeding element 37 and the radiating elements 31 (coupling distance) is increased. Note that the total efficiency is calculated as the radiation efficiency×return loss of the antenna, and the total efficiency is defined as the efficiency of the antenna with respect to the input power. Therefore, by coupling the feeding element 37 and the radiating element 31 through electromagnetic field coupling, a greater degree of freedom for determining the arrangement positions of the feeding element 37 and the radiating element 31 may be obtained and position robustness may be increased. Note that when high position robustness is achieved, this means that the total efficiency of the radiating element 31 may be less likely to be affected even when variations occur in the arrangement positions of the feeding element 37 and the radiating element 31. Also, by obtaining a greater degree of freedom for determining the arrangement positions of the feeding element 37 and the radiating element 31, the space required for installing the dipole antenna elements 30 and 40 may be easily reduced.
Also, in
The impedance of the radiating element 31 becomes higher as the distance from the central portion 90 toward the end portion 34 or the end portion 35 of radiating element 31 increases. In the case of coupling at high impedance by electromagnetic field coupling, even when slight variations occur in the impedance between the feeding element 37 and the radiating element 31, its impact on impedance matching may be relatively small as long as the feeding element 37 and the radiating element 31 are coupled at a sufficiently high impedance of at least a certain level. Thus, to facilitate matching, the feeding portion 36 of the radiating element 31 is preferably positioned at a high impedance portion of the radiating element 31.
For example, to facilitate impedance matching of the dipole antenna element 30, the feeding portion 36 may be positioned at a region spaced apart from the region having the lowest impedance at the resonant frequency of the fundamental mode of the radiating element 31 (the central portion 90 in the present case) by a distance greater than equal to ⅛ of the total length of the radiating element 31 (preferably greater than or equal to ⅙ of the total length, and more preferably greater than or equal to ¼ of the total length). In
The radiating element 41 of the dipole antenna element 40 includes a conductor portion that extends perpendicular to the extending direction of the conductor portion of the radiating element 31 of the dipole antenna elements 30 as described above. The dipole antenna element 40 includes a feeding portion 46 for feeding the radiating element 41, and a feeding element 47 corresponding to a conductor that is spaced apart from the radiating element 41 by a predetermined distance in the Z-axis direction. In
The feeding portion 36 and the feeding portion 46 are located at regions that are shifted from the central portions 90 of the radiating elements 31 and 41 in directions approaching each other. In this way, matching of the dipole antenna elements 30 and 40 may be facilitated, and transmission lines respectively connected to the feeding portions 36 and 46 can be brought closer to each other such that the space required for installing the dipole antenna elements 30 and 40 may be easily reduced.
The feeding element 37 is connected to the feeding point 38, which is connected to a transmission line such as a microstrip line. The feeding element 37 is a linear conductor that feeds the radiating element 31 via the feeding portion 36 without physical contact. In
Note that illustration of the dielectric substrate 110 of
Also, assuming λ0 denotes the radio wave wavelength in vacuum at the resonant frequency of the fundamental mode of the radiating element 31, a shortest distance H4 (≈H2>0) between the feeding element 37 and the radiating element 31 is preferably less than or equal to 0.2 λ0 (more preferably less than or equal to 0.1 λ0, and more preferably less than or equal to 0.05 λ0). By arranging the radiating element 31 and the feeding element 37 to be spaced apart by the shortest distance H4 as described above, the total efficiency of the radiating element 31 may be improved.
Note that the shortest distance H4 refers to the linear distance between sections of the radiating element 31 and the feeding element 37 that are closest to each other. Also, the feeding element 37 and the radiating element 31 may be intersecting or they may not be intersecting when viewed from a given direction, and their intersecting angle may be at any angle as long as the feeding element 37 and the radiating element 31 are coupled by electromagnetic field coupling.
Also, a distance over which the feeding element 37 and the radiating element 31 run parallel to each other at a shortest distance x is preferably less than or equal to ⅜ of the physical length of the radiating element 31. More preferably, the distance is less than or equal to ¼ of the physical length, and more preferably less than or equal to ⅛ of the physical length. The location where the feeding element 37 and the radiating element 31 are at the shortest distance x corresponds to where coupling between the feeding element 37 and the radiating element 31 is strong, and when the distance over which the feeding element 37 and the radiating element 31 run parallel to each other at the shortest distance x is too long, strong coupling may occur at both a high impedance portion and a low impedance portion of the radiating element 31, and as such, impedance matching may become difficult. Thus, to obtain strong coupling only at a region where there is little variation in the impedance of the radiating element 31, the distance over which the feeding element 37 and the radiating element 31 run parallel to each other at the shortest distance x is preferably arranged to be relatively short, and in this way, advantageous effects may be achieved in terms of impedance matching.
Also, assuming Le37 denotes the electrical length that imparts the fundamental mode of resonance to the feeding element 37, Le31 denotes the electrical length that imparts the fundamental mode of resonance to the radiating element 31, and λ denotes a wavelength on the feeding element 37 or the radiating element 31 at a resonant frequency f of the fundamental mode of the radiating element 31, Le37 is preferably less than or equal to (⅜)λ, and Le31 is preferably greater than or equal to (⅜)λ and less than or equal to (⅝)λ.
Also, when the ground plane 70 is formed such that the outer edge portion 71 extends along the radiating element 31, a resonance current (distribution) can be formed on the feeding element 37 and the ground plane 70 as a result of an interaction between the feeding element 37 and the outer edge portion 71, and the feeding element 37 resonates and is coupled with the radiating element 31 by electromagnetic field coupling. For this reason, there is no specific lower limit for the electrical length Le37 of the feeding element 37 as long as the feeding element 37 has a physical length that is sufficient to be coupled to the radiating element 31 by electromagnetic field coupling.
Also, in order to allow a greater degree of freedom in the shape of the feeding element 37, the electrical length Le37 is preferably greater than or equal to (⅛)λ and less than or equal to (⅜)λ, and more preferably greater than or equal to ( 3/16)λ and less than or equal to ( 5/16)λ. By arranging the electrical length Le37 to be within the above ranges, resonance of the feeding element 37 may occur at the design frequency (resonant frequency f) of the radiating element 31, and in this way, the feeding element 37 and the radiating element 31 may resonate without depending on the ground plane 70 and desirable electromagnetic field coupling may be achieved.
Note that when electromagnetic field coupling is achieved this means that impedance matching is achieved. Also, in this case, the feeding element 37 does not have to be designed to have a suitable electrical length according to the resonant frequency of the radiating element 31, and the feeding element 37 may be freely designed as a radiating conductor. In this way, the dipole antenna element 30 may be easily designed to support multiple frequencies. Note that the sum of the length of the outer edge portion 71 of the ground plane 70 extending along the radiating element 31 and the electrical length of the feeding element 37 is preferably greater than or equal to (¼)λ of the design frequency (resonant frequency f). When the feeding element 37 does not include a component such as a matching circuit, a physical length L37 of the feeding element 37 is determined by λg1=λ0k1, where λ0 denotes the radio wave wavelength in vacuum at the resonant frequency of the fundamental mode of the radiating element 31, and k1 denotes a shortening coefficient of a wavelength shortening effect in an actual environment. Here, k1 is calculated based on, for example, a relative permittivity and a relative permeability of a medium (environment) such as an effective relative permittivity (εr1) and an effective relative permeability (μr1) of the dielectric substrate at which the feeding element is arranged, a thickness of the medium (environment), and a resonant frequency. That is, L37 is less than or equal to (⅜) λg1. The shortening coefficient may be calculated based on the physical properties described above, or by actual measurement. For example, a resonant frequency of a target element placed in an environment whose shortening coefficient is to be obtained may be measured, a resonance frequency of the same target element may be measured in an environment whose shortening coefficient for each frequency is known, and the shortening coefficient may be calculated based on a difference between the measured resonance frequencies.
The physical length L37 (corresponding to D1+L31 in
When the fundamental mode of resonance of the radiating element 31 is the dipole mode (i.e., when the radiating element 31 is a linear conductor having open ends), Le31 is preferably greater than or equal to (⅜)λ and less than or equal to (⅝)λ, more preferably greater than or equal to ( 7/16)λ and less than or equal to ( 9/16)λ, and more preferably greater than or equal to ( 15/32)λ and less than or equal to ( 17/32)λ. When a higher-order mode is taken into account, Le31 is preferably greater than or equal to (⅜) λm and less than or equal to (⅝) λm, more preferably greater than or equal to ( 7/16) λm and less than or equal to ( 9/16) λm, and more preferably greater than or equal to ( 15/32) λm and less than or equal to ( 17/32) λm. Here, m denotes a mode number of a higher-order mode and is represented by a natural number. The value of m is preferably an integer between 1 through 5, and more preferably an integer between 1 through 3. In this case, m=1 represents the fundamental mode. When Le31 is within the above ranges, the radiating element 31 may function sufficiently as a radiating conductor, and the efficiency of the dipole antenna element 30 may be desirably high.
A physical length L31 of the radiating element 31 is determined by λg2=λ0k2, where λ0 denotes the radio wave wavelength of in vacuum at the resonant frequency of the fundamental mode of the radiating element 31, and k2 denotes a shortening coefficient of a wavelength shortening effect in an actual environment. Here, k2 is calculated based on, for example, a relative permittivity and a relative permeability of a medium (environment) such as an effective relative permittivity (εr2) and an effective relative permeability (μr2) of the dielectric substrate at which the radiating element 31 is arranged, a thickness of the medium (environment), and a resonant frequency. That is, in an ideal case, the fundamental mode of resonance of the radiating element 31 is the dipole mode and L31 is equal to (½) λg2. The physical length L31 of the radiating element 31 is preferably greater than or equal to (¼) λg2 and less than or equal to (⅝) λg2, and more preferably greater than or equal to (⅜) λg2. The physical length L31 of the radiating element 31 is a physical length that gives Le31. In an ideal case where no other factor is considered, the physical length L31 is equal to Le31. Even when L31 is reduced by using a matching circuit such as an inductor, for example, L31 is preferably greater than zero and less than or equal to Le31, and more preferably greater than or equal to 0.4×Le31 and less than or equal to 1×Le31. By adjusting the length L31 of the radiating element 31 to such a length, the total efficiency of the radiating element 31 may be improved.
For example, when BT resin (registered trademark) CCL-HL870 (M) (Mitsubishi Gas Chemical Company, Inc.) with a relative permittivity εr of 3.4, a loss tangent tan δ of 0.003, and a substrate thickness of 0.8 mm is used as a dielectric substrate, L37 is 20 mm when the design frequency of the feeding element 37 used as a radiating conductor is 3.5 GHz, and L31 is 34 mm when the design frequency of the radiating element 31 is 2.2 GHz.
Note that electromagnetic field coupling of the feeding element 47 and the radiating element 41 and the relationship of their lengths may be similar to those of the feeding element 37 and the radiating element 31 as described above. As such, descriptions thereof will be omitted.
The radiating element 31 is an antenna conductor that functions as an antenna operating in dipole mode by being fed by the feeding element 37 in a non-contact manner at the feeding portion 36 (through electromagnetic field coupling in particular). Similarly, the radiating element 41 is an antenna conductor that functions as an antenna operating in dipole mode by being fed by the feeding element 47 in a non-contact manner at the feeding portion 46 (through electromagnetic field coupling in particular).
In a MIMO antenna according to an embodiment of the present invention, the correlation coefficient between dipole antenna elements may be low, and thus, the distance between the dipole antenna element and the outer edge portion of a ground plane may be freely designed. In particular, as compared to a configuration using monopole antenna elements, in the MIMO antenna according to the present embodiment, the dipole antenna element and the outer edge portion of the ground plane may be arranged closer to each other. That is, assuming λ0 denotes the radio wave wavelength of in vacuum at the design frequency of the fundamental mode of the radiating element of the dipole antenna element, the shortest distance D2 (>0) between the radiating element and the outer edge portion of the ground plane may be arranged to be less than or equal to 0.05 λ0. Further, the distance D2 may be arranged to be less than or equal to 0.043 λ0. Further, the distance D2 may be arranged to be less than or equal to 0.034 λ0. By arranging the distance D2 to be within these ranges, the installation space of the dipole antenna elements may be reduced while maintaining a low correlation coefficient between the dipole antenna elements. For example, in a case where the design frequency is set to 2.5 GHz, the distance D2 is preferably less than or equal to 6 mm, and more preferably less than or equal to 5 mm. Still more preferably, the distance D2 is less than or equal to 4 mm.
In the following, the correlation coefficient between antenna elements is described by comparing a case of using monopole antenna elements with the case of using dipole antenna elements according to an embodiment of the present invention.
In the MIMO antenna 100 that uses the monopole antenna elements 50 and 60, the correlation coefficient increases (the antenna gain decreases) as the radiating elements 51 and 61 come closer to the ground plane 70. That is, in order to improve the antenna gain, the distance D2 has to be increased. As a result, unnecessary space between the radiating elements 51 and 61 and the outer edge portions 71 and 72 of the ground plane 70 have to be secured and the installation space is enlarged.
In contrast, the dipole antenna elements used in the MIMO antennas 1 and 2 according to embodiments of the present invention do not use the ground plane, and thus, even when the radiating elements are brought closer to the ground plane, the correlation coefficient between the dipole antenna elements may be maintained at a low value. That is, the installation space of the dipole antennas may be reduced and the correlation coefficient between the dipole antenna elements may be lowered at the same time.
The plurality of dipole antenna elements according to embodiments of the present invention as described above have radiating elements with conductor portions extending in orthogonal directions (e.g., in the MIMO antenna 1 of
Also, a MIMO antenna according to an embodiment of the present invention has a plurality of dipole antenna elements, and as such, it may be easily implemented in multiband applications supporting a combination of the fundamental mode of the radiating element, and a higher-order mode in which the radiating element resonates at an integer multiple of the resonant frequency of the fundamental mode. In contrast, the MIMO antenna using a plurality of monopole antenna elements may not be suitable for multiband applications because the gap between the resonant frequency of the higher-order mode and the resonant frequency of the fundamental mode is too wide (the resonant frequency of the second order mode is three times that of the fundamental mode).
When the dipole antenna elements and the ground plane are brought too close to each other, the radiation resistance of the radiating elements is reduced due to coupling of the radiating elements and the ground plane such that impedance matching of the MIMO antenna becomes difficult. However, in a MIMO antenna according to an embodiment of the present invention, the feeding portion is arranged at a region other than the central portion of the radiating element (e.g., portion having higher impedance than the central portion), and in this way, impedance matching of the MIMO antenna may be facilitated. Also, the distance D2 between the radiating element of the dipole antenna element and the outer edge portion of the ground plane can be easily reduced such that the installation space of the dipole antenna elements may be reduced and the antenna gain of the MIMO antenna may be improved at the same time.
In particular, when the distance D2 is arranged to be less than or equal to 0.05 λ0 (preferably less than or equal to 0.043 λ0, and more preferably less than or equal to 0.034 λ0), impedance matching of the dipole antenna element may be facilitated by offsetting the feeding portion from the central portion of the radiating element. For example, when the distance D2 is arranged to be less than or equal to 0.05 λ0 (preferably less than or equal to 0.043 λ0, and more preferably less than or equal to 0.034 λ0), the feeding portion is preferably offset from the central portion of the radiating element by a distance greater than or equal to ⅛ of the total length of the radiating element (preferably greater than or equal to ⅙ of the total length, and more preferably greater than or equal to ¼ of the total length).
A MIMO antenna according to an embodiment of the present invention may be implemented in a wireless device (e.g., wireless communication device such as a portable communication terminal). Specific examples of the wireless device include electronic devices such as an information terminal, a mobile phone, a smartphone, a personal computer, a game console, a TV, a music/video player, and the like.
For example, in
If the radiating element 31 is arranged on the surface of the cover glass, the radiating element 31 may be formed by applying a conductive paste such as copper or silver on the surface of the cover glass and firing the applied conductive paste, for example. The conductive paste used in this case is preferably a conductive paste that can be fired at a sufficiently low temperature that would not weaken the strength of the chemically strengthened glass that is used for the cover glass. Also, plating may be performed in order to prevent deterioration of the conductor due to oxidation, for example. Also, the cover glass may be subjected to decorative printing, and a conductor may be formed on the decorative printed portion. Also, in a case where a black concealing layer is formed at the peripheral edges of the cover glass in order to conceal wiring and the like, the radiating element 31 may be formed on the black concealing layer.
Also, the positions of the feeding elements 37, 47, the radiating elements 31, 41, and the ground plane 70 in the height direction parallel to the Z-axis may be different from each other. Alternatively, the positions of the feeding elements 37 and 47, the radiating elements 31 and 41, and the ground plane 70 in the height direction may all be the same or partially the same.
Also, in some embodiments, one feeding element 37 may be configured to feed a plurality of radiating elements. By utilizing a plurality of radiating elements, implementation of multiband operations, wideband operations, and directivity control may be facilitated, for example. Further, a plurality of MIMO antennas may be implemented in a single wireless device.
In the following, S11 characteristics, correlation coefficient characteristics, and total efficiency characteristics (antenna gain characteristics) obtained from the simulation analyses of the MIMO antennas illustrated in
The dimensions of the configuration illustrated in
L11, L21: 4
L12, L22: 34
L13, L23: 3.5
W11, W21: 1.9
The dimensions of the configuration illustrated in
L31, L41: 10.95
L32, L42: 30
L33, L43: 4.05
W31, W41: 1.9
W32, W42: 1.9
W33, W43: 1
The dimensions of the configuration illustrated in
L51, L61: 22.95 (D1=1)
L51, L61: 21.95 (D1=2)
L51, L61: 20.95 (D1=3)
L51, L61: 19.95 (D1=4)
L51, L61: 18.95 (D1=5)
L51, L61: 17.95 (D1=6)
L52, L62: 5
W51, W61: 1.9
W52, W62: 1.9
Also, the thickness (height) in the Z-axis direction of the ground plane 70, the feeding elements, and the radiating elements was set to 0.018 mm. The dielectric substrate 80 was arranged to have a relative permittivity of εr=3.3 and a loss tangent of tan δ=0.003, and the dielectric substrate 110 was arranged to have a relative permittivity of εr=8.6 and a loss tangent of tan δ=0.000326. Also, in
Note that in
The S11 of the MIMO antennas using dipole antenna elements (
Also, it can be appreciated that the correlation coefficients of the MIMO antennas using dipole antenna elements (
Meanwhile, it can be appreciated that the total efficiency of the MIMO antennas using dipole antenna elements (
In this way, the installation space of the antenna elements may be reduced and the correlation coefficient between the antenna elements may be lowered at the same time.
In the following, comparison results of comparing the characteristics of the MIMO antennas 1, 2, and 100 having radiating elements with conductor portions that are orthogonal (
Note that the dimensions of the configurations of
Table 1 indicates the frequencies at which the minimum S11 was obtained (i.e., resonant frequencies at which best matching was obtained) in the MIMO antennas 1, 2, and 100 according to the graphs showing the S11 characteristics of the MIMO antennas 1, 2, and 100 (
Table 2 indicates the correlation coefficients at the frequencies at which the minimum S11 was obtained in the MIMO antennas 1, 2, and 100 according to the graphs showing the correlation coefficient characteristics of the MIMO antennas 1, 2, and 100 (
Table 3 indicates the total efficiencies of the MIMO antennas 1, 2, and 100 at the frequencies at which the minimum S11 was obtained according to the graphs showing the total efficiency characteristics of the MIMO antennas 1, 2, and 100 (
Note that in Table 1 through Table 3, “1 mm,” “2 mm,” “3 mm,” “4 mm,” “5 mm,” and “6 mm” represent the distance D1, and when converted into the shortest distance D2, they would respectively be “3 mm,” “3.4 mm,” “4.1 mm,” “4.9 mm,” “5.7 mm,” and “6.6 mm.”
In the following, comparison results of comparing the characteristics of MIMO antennas 3, 4, and 101 having radiating elements with conductor portions that are parallel (
The dimensions of the configuration illustrated in
L11, L21: 6.5
L12, L22: 31.5
L3: 2.1
W11, W21: 1.9
The dimensions of the configuration illustrated in
L31, L41: 10.95
L32, L42: 30
L4: 2.1
W31, W41: 1.9
W32, W42: 1.9
W33, W43: 1
The dimensions of the configuration illustrated in
L51, L61: 22.95 (D1=1)
L51, L61: 21.95 (D1=2)
L51, L61: 20.95 (D1=3)
L51, L61: 19.95 (D1=4)
L51, L61: 18.95 (D1=5)
L51, L61: 17.95 (D1=6)
L101: 2.1
W51, W61: 1.9
W52, W62: 1.9
Note that the thickness of the ground plane 70, and the feeding/radiating elements, and the dimensions of the dielectric substrate were set up to be the same as those of Application Example 1.
Table 4 indicates the frequencies at which the minimum S11 was obtained (i.e., resonant frequencies at which best matching was obtained) in the MIMO antennas 3, 4, and 101 according to the graphs showing the S11 characteristics of the MIMO antennas 3, 4, and 101 (
Table 5 indicates the correlation coefficients at the frequencies at which the minimum S11 was obtained in the MIMO antennas 3, 4, and 101 according to the graphs showing the correlation coefficient characteristics of the MIMO antennas 3, 4, and 101 (
Table 6 indicates the total efficiencies of the MIMO antennas 3, 4, and 101 at the frequencies at which the minimum S11 was obtained according to the graphs showing the total efficiency characteristics of the MIMO antennas 3, 4, and 101 (
Tables 4 through 6, “1 mm,” “2 mm,” “3 mm,” “4 mm,” “5 mm,” and “6 mm” represent the distance D1, and when converted into the shortest distance D2, they would respectively be “3 mm,” “3.4 mm,” “4.1 mm,” “4.9 mm,” “5.7 mm,” and “6.6 mm.”
In the following, results of measuring the voltage standing wave ratio (VSWR) of the MIMO antenna 1 using dipole antenna elements (
Note that the fundamental mode resonant frequency of the radiating elements 11 and 21 were set in the vicinity of 2.4 GHz, and the dimensions of the configuration illustrated in
Table 7 indicates S11 values calculated from the VSWR that were measured upon changing the distance D2 and the offset distance. Note that the “Distance from Ground” in Table 7 represents a normalized value (D2/125) corresponding to the actual distance D2 normalized by the wavelength in vacuum λ0 of the frequency 2.4 GHz (λ0=125 mm). The “Feeding Position” in Table 7 represents a ratio of a shift amount (=offset distance) of the feeding portions 16 and 26 toward the end portions 14 and 24 from the central portion 90 with respect to the total length (=38 mm) of the radiating elements 11 and 21. When this ratio is 0, this means that the feeding portions 16 and 26 are located at the central portion 90. Also, in Table 7, S11 values that are less than −6.0 are surrounded by dotted lines. It is assumed that good matching of the dipole antenna elements can be achieved when the S11 is less than −6.0.
According to Table 7, if the radiating element is spaced apart from the ground plane such that the distance D2 is greater than 0.046 λ0 and less than 0.053 λ0 (e.g., D2=0.05 λ0), the feeding portion may be located in the vicinity of the central portion of the radiating element.
Also, according to Table 7, when the distance D2 is less than or equal to 0.05 λ0, the feeding portion is preferably offset from the central portion of the radiating element by a distance greater than or equal to ⅛ (=0.125) of the total length of the radiating element (0.11<0.125<0.13). Also, according to Table 7, when the distance D2 is less than or equal to 0.043 λ0, the feeding portion is preferably offset from the central portion of the radiating element by a distance greater than or equal to ⅙ (=0.166) of the total length of the radiating element (0.16<0.166<0.24). Also, according to Table 7, when the distance D2 is less than or equal to 0.034 λ0, the feeding portion is preferably offset from the central portion of the radiating element by a distance greater than or equal to ¼ (=0.25) of the total length of the radiating element (0.24<0.25<0.32).
Although the MIMO antenna according to the present invention has been described above with respect to certain illustrative embodiments, the present invention is not limited to the above embodiments. Note that various modifications and improvements may be made within the scope of the present invention, for example, by combining or substituting the above embodiments with a part or all of other exemplary embodiments.
For example, the MIMO antenna is not limited to having two dipole antenna elements but may have three or more dipole antenna elements.
Also, the plurality of dipole antenna elements is not limited to the configurations illustrated in the drawings. For example, the dipole antenna element 10 of
Also, the dipole antenna element is not limited to those including a linear conductor portion extending linearly, but may also include a curved conductor portion. For example, the dipole antenna element may include an L-shaped conductor portion, a meander-shaped conductor portion, or a conductor portion that branches out from a branch point.
Also, the feeding element may include a stub, or a matching circuit, for example. In this way, an area of a substrate occupied by the feeding element may be reduced.
Also, the transmission line to which the feeding portion is connected is not limited to a microstrip line. For example, the transmission line may be a strip line, or a coplanar waveguide having a ground plane (coplanar waveguide with a ground plane arranged on a surface on the opposite side of a conductor face). The feeding element and the feeding points may be connected via these different types of transmission lines, for example.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2013-002988 | Jan 2013 | JP | national |
The present application is a continuation application filed under 35 U.S.C. 111(a) claiming benefit under 35 U.S.C. 120 and 365(c) of PCT International Application No. PCT/JP2014/050356 filed on Jan. 10, 2014 and designating the U.S., which claims priority to Japanese Patent Application No. 2013-002988 filed on Jan. 10, 2013. The entire contents of the foregoing applications are incorporated herein by reference.
| Number | Date | Country | |
|---|---|---|---|
| Parent | PCT/JP2014/050356 | Jan 2014 | US |
| Child | 14790472 | US |
