Patent Grant
11631927
Electromagnetic compatibility (EMC) testing is widely performed on equipment, such as complete systems, integrated circuits, printed circuit boards (PCBs) and other electronic modules, to determine whether the equipment radiates more radio frequency (RF) energy than either allowed by regulations or acceptable to avoid interference with wireless receivers, or to determine if the equipment is susceptible to electromagnetic (EM) disturbances. An EMC test may involve a number of different EM analyses. As an example, EMC testing may involve radiating electromagnetic waves at the equipment, measuring the emissions from the equipment or testing the immunity to electrostatic discharges (ESD).
Electromagnetic interference (EMI) testing is usually performed according to standards, e.g., the Federal Communications Commission (FCC) normally uses a semi-anechoic chamber or an open area test site to measure the fields in the far field region. Such methodology, however, provides little insight into the root cause of EMI problems. EMI analysis can also be performed by near-field scanning, i.e., measuring local electric or magnetic field around the equipment under test (EUT) to identify areas of strong electric or magnetic field. This near-field information may then assist in identifying the cause of an EMI problem of the EUT based on an implicit assumption that an area of strong field is the cause of the EMI problem.
An immunity or ESD analysis can also be performed by subjecting the EUT to strong electromagnetic fields (immunity) or injecting ESD currents into the EUT at different locations. Such analysis can then include determining whether an error has occurred because of the RF field or ESD current stress injected into the selected location.
The difference between the immunity analysis and the ESD analysis is the type of noise injected. Modulated RF signals are usually injected for the immunity analysis, whereas narrow pulses (having one or sub nanosecond rise time) are injected for an ESD analysis. Another relevant difference is that immunity analysis subjects the EUT to fields, most often in the far-field region of the transmitting antenna, while ESD testing injects currents directly into the EUT. Indirect ESD testing, which subjects the EUT only to the fields of the ESD, is also performed.
A method that provides better insight into the possible root cause of an immunity or susceptibility problem is susceptibility scanning. In this method, a probe is moved above the equipment (e.g., PCB, cables etc.) and a strong local field is caused by injecting pulses or RF signals into the probe. The probe is moved around the equipment and the reaction of the equipment is observed. This way, local areas of higher susceptibility can be identified.
The near-field EMI scanning and the near-field susceptibility scanning both identify local effects, which are difficult to connect to the system level performance of the EUT. Thus, strong local fields might be the cause of strong radiated emissions, and local areas of high susceptibility might be the reason for immunity or ESD problems as they show up if the complete system is tested in accordance to the standards, such as IEC 61000-4-3 (radiated immunity) or IEC 61000-4-2 (ESD).
Phase-resolved near-field scanning (NFS) has been widely used in electromagnetics and antenna research for many years. With the ongoing development of various technologies (e.g., high speed communication systems, cloud computing, autonomous vehicles, etc.), millimeter (mm) wavebands above 20 GHz are being intensively studied, and there is a great need for high frequency probes and a corresponding methodology for calibrating them. In most EMC near-field scanning systems, a probe (or a set of probes) captures a large set of near-field data on a surface plane close to the EUT. For example, an E-field probe or an H-field probe can be used to visualize the E-field or the H-field near-field distribution over an EUT.
Various probe calibration methods suitable for different frequency ranges are well known in the art including, for example, the different calibration methods and their typical frequency ranges disclosed in the Institute of Electrical and Electronics Engineers (IEEE) standards (See e.g., IEEE Standard 1309-2013), as well as methods disclosed by the International Electrotechnical Commission (IEC) (See e.g., IEC 61000-4-20 Annex E which discusses E-field probe calibration in transverse electromagnetic (TEM) waveguides).
Previous work has shown that referring a measured voltage to the known fields of a 50Ω transmission line (TL) is an effective method for calculating the probe factor. If the measurements are done with a Vector Network Analyzer (VNA) (e.g., the electrical analyzing instrument 110 illustrated in
where ref is the normalized near-field strength (E or H) from a simulation at a given input voltage and at a given height above the TL:
Here, it should be appreciated that a “pure” TEM mode is generally desirable for calibration since a pure TEM is frequency-independent and the field components are well defined. However, it should be further appreciated that, although a physical structure can be pure TEM, any given structure will always have frequency limitations since transitions and inhomogeneity cause non-TEM modes (e.g., a transition from connector to transmission line would create some non-TEM mode behavior). Therefore, in the physical world, the desired features of a transmission line for calibration could generally be prioritized as follows:
It should be noted that a simple microstrip can be used up to a few gigahertz (GHz), while a grounded coplanar waveguide (GCPW) generally performs better for higher frequencies. The inhomogeneous medium of a coplanar waveguide (CPW), however, undesirably causes non-TEM behavior, wherein calibration is more difficult with non-TEM modes (e.g., frequency-dependent) and more inaccurate because of the longitudinal field component.
Accordingly, there is a need for a transmission line system and method for probe calibration that comes as close as possible to exhibiting pure TEM line behavior.
The following presents a simplified summary of one or more aspects of the present disclosure, in order to provide a basic understanding of such aspects. This summary is not an extensive overview of all contemplated features of the disclosure, and is intended neither to identify key or critical elements of all aspects of the disclosure nor to delineate the scope of any or all aspects of the disclosure. Its sole purpose is to present some concepts of one or more aspects of the disclosure in a simplified form as a prelude to the more detailed description that is presented later.
Various aspects directed towards a transmission line for probe calibration are disclosed. In a particular example, an integrated transverse electromagnetic (TEM) transmission line structure for probe calibration is disclosed, which includes a printed circuit board (PCB) and an air-dielectric coplanar waveguide (CPW). For this embodiment, the air-dielectric CPW includes an air trace in a cutout slot of the PCB.
In another aspect of the disclosure, a method for probe calibration is disclosed, which comprises forming a first trace on one end of an integrated TEM transmission line structure, and a second trace on an opposite end of the integrated TEM transmission line structure. The method further comprises forming a PCB and forming an air-dielectric CPW on the PCB. For this embodiment, the air-dielectric CPW includes an air trace in a cutout slot of the PCB.
In yet another aspect of the disclosure, a system for probe calibration is disclosed, which includes an air-dielectric CPW with an air trace. For this embodiment, a first connector is electrically coupled to a first end of the air-dielectric CPW, and a second connector is electrically coupled to a second end of the air-dielectric CPW. In a particular aspect of this embodiment, the system further includes a first grounded CPW (GCPW) in between a first end of the air-dielectric CPW and the first connector, and a second GCPW in between a second end of the air-dielectric CPW and the second connector. Within such embodiment, the first GCPW includes a first trace aligned with the air trace, and the second GCPW includes a second trace aligned with the air trace.
Other aspects and advantages of the present invention will become apparent from the following detailed description, taken in conjunction with the accompanying drawings, illustrated by way of example of the principles of the invention.
Overview
With the ongoing development of various technologies (e.g., fifth generation (5G) wireless communication systems, radar systems, cloud computing, Internet-of-Things (IoT), autonomous vehicles, etc.) frequencies as high as 40 gigahertz (GHz) have become relevant for electromagnetic interference (EMI) near-field scanning. Aspects disclosed herein are directed towards a transmission line for probe calibration that includes an air-dielectric coplanar waveguide (CPW). Because air is a homogeneous dielectric, the transmission line structure disclosed herein becomes an almost pure transverse electromagnetic (TEM) transmission line, which is preferable for probe calibration to a coplanar waveguide. Moreover, the air-dielectric CPW design disclosed herein is particularly desirable for high frequency probe calibration since it provides a more pure TEM structure relative to TEM structures that utilize a conventional CPW.
Exemplary Near-Field Scanning System
It should be appreciated that the transverse electromagnetic (TEM) transmission line structure disclosed herein can be used for any of various types of near-field measurements including, for example, emission and immunity near-field scanning purposes (e.g., electromagnetic interference (EMI) testing, electrostatic discharge (ESD) testing, current spreading (CSP), phase measurement (PHM), emission source microscopy (ESM), resonance testing, etc.). With reference to
In some embodiments, the electrical analyzing instrument 110 is a network analyzer 110, in particular, a vector network analyzer. Thus, the electrical analyzing instrument 110 is referred to herein as a network analyzer. In these embodiments, the electrical analyzing instrument 110 can be one of many commercially available vector network analyzers. However, in related embodiments, the electrical analyzing instrument 110 may be a spectrum analyzer with a tracking generator or a spectrum analyzer with a radio frequency (RF) generator, or an RF source and an oscilloscope.
The motor driver 116 of the automatic scanning subsystem 106 is designed to provide driving signals to the probe positioning mechanism 114 so that the probe 108 can be displaced to desired testing locations of the EUT and/or be rotated to desired rotational positions. The motor driver 116 is electrically connected to the motors 130 and 132 of the probe positioning mechanism 114 to provide driving signals to these motors so that the probe 108 can be linearly displaced along the X-axis and the Y-axis. The motor driver 116 is also electrically connected to the motors 138 and 140 of the scan head 122 to provide driving signals to these motors so that the probe 108 can be vertically moved along the Z-axis and be rotated about the Z-axis. In an embodiment, the motor driver 116 is controlled by the processing device 112. Thus, the processing device 112 is able to track the movements of the probe 108 that is being displaced by the automatic scanning subsystem 106.
Exemplary Transverse Electromagnetic (TEM) Transmission Line Embodiment
Referring next to
In a particular aspect disclosed herein, the air trace 210 is plated (e.g., a copper plating) except on each of a first end of the cutout slot and a second end of the cutout slot. Namely, for this embodiment, it is contemplated that each of the first end of the cutout slot and the second end of the cutout slot are un-plated (i.e., un-plated end 212 and un-plated end 214, respectively). As used herein, it should be appreciated that “plating” (e.g., edge-plating) is defined as the process of adding metal to the sides of a printed circuit board (PCB). It should be further appreciated that embodiments are also contemplated in which the air trace 210 is un-plated.
Various other aspects of the air-dielectric CPW 200 are also contemplated. For instance, in order to avoid reflections, it is contemplated that the impedance of the air-dielectric CPW 200 is matched with the impedance of the first GCPW 260 and/or second GCPW 270. Similarly, since at least one connector may be electrically coupled to either the first GCPW 260 or the second GCPW 270 (See e.g.,
In another aspect of the disclosure, the dimensions of the integrated TEM transmission line structure are carefully selected so as to facilitate near-pure TEM behavior. For instance, dimensions may be selected to facilitate maintaining one of an electric near-field or a magnetic near-field having an orthogonal component across the air trace 210 and a minimized longitudinal component across the air trace 210. Similarly, the dimensions may be selected to facilitate maintaining one of an electric near-field or a magnetic near-field having an amplitude along a line across the first and second GCPWs, 260 and 270, wherein the dimensions further facilitate minimizing a frequency dependence of the amplitude.
Referring next to
In general, it is contemplated that the first and second GCPWs, 360 and 370, are not plated, whereas the air trace 310 may or may not be plated. For instance, in a particular aspect disclosed herein, the air trace 310 is plated (e.g., a copper plating) except on each of a first end of the cutout slot and a second end of the cutout slot. Namely, for this embodiment, it is contemplated that each of the first end of the cutout slot and the second end of the cutout slot are un-plated (i.e., un-plated end 312 and un-plated end 314, respectively).
In another aspect of the disclosure, the dimensions of the integrated TEM transmission line structure are again carefully selected so as to facilitate near-pure TEM behavior. For instance, to avoid reflections caused by the transition from the air-dielectric CPW 300 to the first and second GCPWs, 360 and 370, dimensions may be selected to facilitate an impedance match of the air-dielectric CPW 300 and the first and second GCPWs, 360 and 370. Similarly, to avoid reflections caused by the transition from the first and second GCPWs, 360 and 370, to either the first connector 340 or the second connector 350, the dimensions may be selected to facilitate an impedance match of the first connector 340 to the first GCPW 360, and an impedance match of the second connector 350 to the second GCPW 370.
In another aspect of the disclosure, the dimensions of the TEM transmission line structure are carefully selected so as to facilitate maintaining one of an electric near-field or a magnetic near-field having an orthogonal component across the air trace 310 and a minimized longitudinal component across the air trace 310. Similarly, the dimensions may be selected to facilitate maintaining one of an electric near-field or a magnetic near-field having an amplitude along a line across the first and second GCPWs, 360 and 370, wherein the dimensions further facilitate minimizing a frequency dependence of the amplitude.
Referring next to
Process 700 begins at block 710 with the forming of a PCB (e.g., PCB 280), and concludes with the forming of an air-dielectric CPW (e.g., air-dielectric CPW 200) on the PCB at block 720, wherein the air-dielectric CPW includes an air trace (e.g., air trace 210) formed in a cutout slot of the PCB. In a particular aspect disclosed herein, process 700 may further comprise forming a first GCPW (e.g., first GCPW 260) on a first end of the air-dielectric CPW, wherein the first GCPW includes a first trace (e.g., first trace 220) aligned with the air trace, and forming a second GCPW (e.g., second GCPW 270) on a second end of the air-dielectric CPW, wherein the second GCPW includes a second trace (e.g., second trace 230) aligned with the air trace.
It is also contemplated that process 700 may further comprise plating the air trace (e.g., a copper plating). Within such embodiment, it is contemplated that process 700 may further comprise the removal of each of a first plating and a second plating from each of a first end of the cutout slot (e.g., un-plated end 212) and a second end of the cutout slot (e.g., un-plated end 214).
Various other aspects of process 700 are also contemplated. For instance, in order to avoid reflections, it is contemplated that process 700 may further comprise matching the impedance of the air-dielectric CPW with the impedance of the first and second GCPWs. Similarly, since process 700 may also comprise electrically coupling a connector to either end of the air-dielectric CPW (e.g., either directly to either end of the air-dielectric CPW, or via the first and second GCPWs), process 700 may further comprise matching the impedance of the connectors with the impedance of the air-dielectric CPW (i.e., to avoid reflections caused by a transition from the connector to the air-dielectric CPW, if the connectors are directly connected to the air-dielectric CPW), and/or matching the impedance of the connectors with the impedance of the first and second GCPWs (i.e., to avoid reflections caused by a transition from the connector to the first or second GCPW, if the connectors are connected to the air-dielectric CPW via the first and second GCPWs).
In another aspect of the disclosure, it is contemplated that process 700 may comprise selecting the dimensions of the integrated TEM transmission line structure so as to facilitate near-pure TEM behavior. For instance, the selecting of dimensions may facilitate maintaining one of an electric near-field or a magnetic near-field having an orthogonal component across the air trace and a minimized longitudinal component across the air trace. Similarly, the selecting of dimensions may facilitate maintaining one of an electric near-field or a magnetic near-field having an amplitude along a line across the first and second GCPWs to further facilitate minimizing a frequency dependence of the amplitude.
Exemplary Transverse Electromagnetic (TEM) Transmission Line Implementations
An exemplary implementation of aspects disclosed herein is now described with reference to components illustrated in
It should be noted that software was used to simulate the above exemplary implementation of the TEM transmission line structure disclosed herein. In a particular simulation, input power was normalized to 1 watt, and 6 million mesh cells were used. With reference to
Based on the aforementioned desired features of a transmission line for calibration (i.e., well defined field components; frequency-independent near-field amplitude; and impedance matched), the calibration structure for this particular implementation was evaluated with respect to the longitudinal field component (non-TEM mode), S-parameters, and amplitude across and along the air trace. Time domain reflectometry (TDR) was used in measurements to analyze imperfections in the structure. The near-field was evaluated 1 mm above the transmission line, which is a typical scanning height for high frequency (up to EHF band of radio frequencies) applications. Here, it should be noted that, although a 1 mm height was used, other heights may also be used.
Referring next to
Referring next to
Comparisons were also made between the quasi-TEM behavior of the GCPW and the more pure TEM behavior of the air-dielectric CPW. For instance,
The measurements and simulations of the air-dielectric CPW disclosed herein, reveal that the transition between the GCPW and the air-dielectric CPW should be accounted for in order to avoid standing waves and loss. At 40 GHz, the wavelength in free space is 7.5 mm. With a PCB thickness of 1 mm, the length of the detour the return current has to travel in the transition is more than 1/10 of the wavelength. Hence, this distance is comparable with the wavelength. To overcome this problem, it is contemplated that the PCB thickness can be made thinner. For instance, the PCB may be designed with 0.8 mm and 0.6 mm thicknesses, and the air and substrate gaps may be adjusted to the thinner board in order to obtain a 50Ω characteristic impedance.
To demonstrate the effects of varying PCB thickness,
In another aspect disclosed herein, in order to overcome reflections, it is contemplated that attenuators can be included along the transmission line. However, it should be noted that, since there are two transitions at both ends (e.g., on one end, first connector 340 to first GCPW 360, and first GCPW 360 to air trace 310; and on the opposite end, second connector 350 to second GCPW 370, and second GCPW 370 to air trace 310), four attenuators might be needed, which may cause an undesirably large reduction in the dynamic range.
Although specific embodiments of the invention have been described and illustrated, the invention is not to be limited to the specific forms or arrangements of parts so described and illustrated. The scope of the invention is to be defined by the claims appended hereto and their equivalents.
| Number | Name | Date | Kind |
|---|---|---|---|
| 5920240 | Alexanian | Jul 1999 | A |
| 20020180570 | Facer | Dec 2002 | A1 |
| 20070024515 | Suh | Feb 2007 | A1 |
| 20160018393 | Brown | Jan 2016 | A1 |
| Number | Date | Country | |
|---|---|---|---|
| 20200411934 A1 | Dec 2020 | US |
