Patent Application
20200284873
The present invention is directed to radar systems, and in particular to radar systems for vehicles.
The use of radar to determine range, velocity, and angle (elevation or azimuth) of objects in an environment is important in a number of applications including automotive radar and gesture detection. Radar systems typically transmit a radio frequency (RF) signal and listen for the reflection of the radio signal from objects in the environment. A radar system estimates the location of objects, also called targets, in the environment by correlating delayed versions of the received radio signal with the transmitted radio signal. A radar system can also estimate the velocity of the target by Doppler processing. A radar system with multiple transmitters and multiple receivers can also determine the angular position of a target.
A radar system consists of transmitters and receivers. The transmitters generate a baseband signal which is up converted to a radio frequency (RF) signal that propagates according to an antenna pattern. The transmitted signal is reflected off of object or targets in the environment. The received signal at each receiver is the totality of the reflected signal from all targets in the environment. The receiver down converts the received signal to baseband and compares the baseband received signal to the baseband signal at one or more transmitters. This is used to determine the range, velocity and angle of targets in the environment.
Embodiments of the present invention provide methods and an apparatus to improve the performance of a radar system. An exemplary radar system is configured to provide sequences of spreading code chips to digital signal generators of each transmitter and to digital baseband processing sections of each receiver. A programmable code generator unit is communicatively coupled to each of the transmitters and receivers and is configured to provide the sequences of spreading code chips. The programmable code generator provides one spreading sequence consisting of spreading code chips to each transmitter while multiple spreading codes are supplied to each receiver, such that a digital signal generator and a corresponding digital baseband processing section of a first transmitter and each receiver will each be supplied the same spreading code used by the transmitter and each receiver to generate transmitted radio signals and to process received radio signals, respectively.
A radar sensing system for a vehicle in accordance with an embodiment of the present invention includes a plurality of transmitters, a plurality of receivers, and a programmable code generation unit. The plurality of transmitters is configured for installation and use on a vehicle and is configured to transmit radio signals. The plurality of receivers is configured for installation and use on the vehicle, and is configured to receive radio signals that include the transmitted radio signals transmitted by the transmitters and reflected from objects in an environment. The programmable code generation unit is configured to provide sequences of spreading code chips to digital signal generators of the transmitters and to digital baseband processing sections of the receivers. Particular sequences of spreading code chips are provided to individual transmitters. A particular sequence of spreading code chips is used by a digital signal generator as part of the process to generate transmitted radio signals. Multiple sequences of spreading code chips are used by a digital baseband processing section of a receiver to process received radio signals.
A method for controlling a radar sensing system comprising a plurality of transmitters configured for installation and use on a vehicle, a plurality of receivers configured for installation and use on the vehicle, and a programmable code generation unit configured for installation and use on the vehicle. The method includes transmitting, with the transmitters, radio signals and receiving, with the receivers, radio signals that include the transmitted radio signals transmitted by the transmitters and reflected from objects in an environment. A radio frequency (RF) front end of a first receiver is used to process the received radio signals and output processed radio signals. An analog-to-digital converter (ADC) of the first receiver is used to sample the processed radio signals and output data samples. The method further includes generating, with the programmable code generation unit, spreading sequences used by digital signal generators of the transmitters to generate transmitted radio signals and used by digital baseband processing sections of the receivers to process received radio signals.
In an aspect of the present invention, the programmable code generation unit may be configured to generate a plurality of different spreading codes and/or a plurality of maximum length sequences.
In another aspect of the present invention, the programmable code generation unit may be configured to generate a plurality of codes from one of a Hadamard code, a Golay code, a Frank-Zadoff-Chu code, and an APAS code.
In yet a further aspect of the present invention, the programmable code generation unit may include a control processor and a plurality of feedback shift registers. The shift register feedback connections and shift register contents of the feedback shift registers are defined by the control processor. The programmable code generation unit also includes logic that alters the output of the feedback shift registers in order to alter properties of the sequences produced by the feedback shift registers. The plurality of feedback shift registers may be loaded with any seed contents, as defined by the control processor. The plurality of feedback shift registers may also be further configured with any set of feedback connections, as defined by the control processor. The shift register sequences may be altered to provide DC balance within a sequence of a transmitter and/or across the sequences for different transmitters.
These and other objects, advantages, purposes and features of the present invention will become apparent upon review of the following specification in conjunction with the drawings.
The present invention will now be described with reference to the accompanying figures, wherein numbered elements in the following written description correspond to like-numbered elements in the figures. Methods and systems of the present invention provide for a programmable code generation unit configured to generate sequences to be received by both transmitters and receivers of a radar system and used to generate transmitted signals as well as used to process received signals. An exemplary programmable code generation unit comprises a control processor and a plurality of feedback shift registers. The control processor configures shift register feedback connections and selects contents (seeds) of the feedback shift registers in order to alter properties of the sequences that are generated by the feedback shift registers.
An exemplary radar system operates by transmitting one or more signals from one or more transmitters and then listening for reflections of those signals from objects in the environment by one or more receivers. By comparing the transmitted signals and the received signals, estimates of the range, velocity, and angle (azimuth and/or elevation) of the objects can be estimated.
There are several ways to implement a radar system. One way, illustrated in
A radar system with multiple antennas, transmitters and receivers is shown in
The radar system 300 may be connected to a network via an Ethernet connection or other types of network connections 314, such as, for example, CAN-FD and FlexRay. The radar system 300 may also have memory (310, 312) to store software used for processing the signals in order to determine range, velocity and location of objects. Memory 310, 312 may also be used to store information about targets in the environment. There may also be processing capability contained in the ASIC 208 apart from the transmitters 203 and receivers 204.
The description herein includes an exemplary radar system in which there are NT transmitters and NR receivers for NT×NR virtual radars, one for each transmitter-receiver pair. For example, a radar system with eight transmitters and eight receivers will have 64 pairs or 64 virtual radars (with 64 virtual receivers). When three transmitters (Tx1, Tx2, Tx3) generate signals that are being received by three receivers (Rx1, Rx2, Rx3), each of the receivers is receiving the transmission from each of the transmitters reflected by objects in the environment. Each receiver can attempt to determine the range and Doppler of objects by correlating with delayed replicas of the signal from each of the transmitters. The physical receivers may then be “divided” into three separate virtual receivers, each virtual receiver correlating with delay replicas of one of the transmitted signals.
There are several different types of signals that transmitters in radar systems employ. A radar system may transmit a pulsed signal or a continuous signal. In a pulsed radar system, the signal is transmitted for a short time and then no signal is transmitted. This is repeated over and over. When the signal is not being transmitted, the receiver listens for echoes or reflections from objects in the environment. Often a single antenna is used for both the transmitter and receiver and the radar transmits on the antenna and then listens to the received signal on the same antenna. This process is then repeated. In a continuous wave radar system, the signal is continuously transmitted. There may be an antenna for transmitting and a separate antenna for receiving.
Another classification of radar systems is the modulation of signal being transmitted. A first type of continuous wave radar signal is known as a frequency modulated continuous wave (FMCW) radar signal. In an FMCW radar system, the transmitted signal is a sinusoidal signal with a varying frequency. By measuring a time difference between when a certain frequency was transmitted and when the received signal contained that frequency, the range to an object can be determined. By measuring several different time differences between a transmitted signal and a received signal, velocity information can be obtained.
A second type of continuous wave signal used in radar systems is known as a phase modulated continuous wave (PMCW) radar signal. In a PMCW radar system, the transmitted signal from a single transmitter is a sinusoidal signal in which the phase of the sinusoidal signal varies. Typically, the phase during a given time period (called a chip period or chip duration) is one of a finite number of possible phases. A spreading code consisting of a sequence of chips, (e.g., +1, +1, −1, +1, −1 . . . ) is mapped (e.g., +1→0, −1→π) into a sequence of phases (e.g., 0, 0, π, 0, π . . . ) that is used to modulate a carrier to generate the radio frequency (RF) signal. The spreading code could be a periodic sequence or could be a pseudo-random sequence with a very large period so it appears to be a nearly random sequence. The spreading code could be a binary code (e.g., +1 or −1). The resulting signal has a bandwidth that is proportional to the rate at which the phases change, called the chip rate Rc, which is the inverse of the chip duration Tc=1/Rc. By comparing the return signal to the transmitted signal, the receiver can determine the range and the velocity of reflected objects.
In some radar systems, the signal (e.g. a PMCW signal) is transmitted over a short time period (e.g. 1 microsecond) and then turned off for a similar time period. The receiver is only turned on during the time period where the transmitter is turned off. In this approach, reflections of the transmitted signal from very close targets will not be completely available because the receiver is not active during a large fraction of the time when the reflected signals are being received. This is called pulse mode.
The radar sensing system of the present invention may utilize aspects of the radar systems described in U.S. Pat. Nos. 9,846,228; 9,806,914; 9,791,564; 9,791,551; 9,772,397; 9,753,121; 9,599,702; 9,575,160 and/or 9,689,967, and/or U.S. Publication Nos. US-2017-0309997; US-2017-0307728 and/or US-2017-0310758, and/or U.S. patent application Ser. No. 15/496,038, filed Apr. 25, 2017, Ser. No. 15/689,273, filed Aug. 29, 2017, and/or Ser. No. 15/705,627, filed Sep. 15, 2017, and/or U.S. provisional applications, Ser. No. 62/486,732, filed Apr. 18, 2017, Ser. No. 62/528,789, filed Jul. 5, 2017, Ser. No. 62/573,880, filed Oct. 18, 2017, Ser. No. 62/598,563, filed Dec. 14, 2017, and/or Ser. No. 62/623,092, filed Jan. 29, 2018, which are all hereby incorporated by reference herein in their entireties.
The digital baseband processing section 490 of a receiver 450 in a radar system will processes samples from the ADC 480. This processing involves calculating correlations with different delays of the baseband signal for each of the possible transmitted baseband signals. There are a variety of ways that the correlation process can be accomplished, including the use of a matched filter and Fourier transform processing. The processing for calculating correlations can be done for a set of distances. Each delay of the baseband signal corresponds to a range for a target. A range bin is the range corresponding to a certain delay of the baseband signal used in a correlation. The correlation with a particular delay is an indication of whether or not a target is present at the particular range. For example, a spreading code might have a chip rate of Rc=500 Mchips/second which would correspond to a chip duration of Tc=2 nanoseconds (ns). The receiver 450 might perform correlation delays at intervals of 2 ns, starting at a delay of 2 ns up to a maximum delay of 256 ns. That is, 128 different correlations would be performed. A target at a distance of 30 meters would produce a delay of 200 ns. A correlation with a baseband spreading code at a delay of 100 chips (200 ns) would produce a large magnitude output because the reflected signal off the target at 30 meters would produce a large correlation with the baseband spreading signal also delayed by 200 ns. While a target at a distance of 30.3 meters would produce a large correlation with a baseband signal delayed by 101 chips (202 ns). Therefore, targets at a distance between close to 30 meters (e.g. within ±0.15 meters would produce larger outputs when correlated with a baseband signal delay by 100 chips than a baseband signal delayed by either 99 chips or 101 chips. The range bin then corresponding to 30 meters would be of a width of 0.3 meters. A correlation unit might be set up to produce a certain set of correlations (e.g. 128 correlations for range bins starting at 0.3 meters to a distance of 38.4 meters). In other words, a correlation unit might be capable of producing a certain number of correlations or determine the presence of targets in a certain set of range bins. A correlation unit could be set up to consider different sets of range bins with which to perform correlations over.
Spreading codes with good autocorrelation functions are important so that the receiver can distinguish targets from different locations. Sometimes the spreading codes are called sequences and sometimes they are called codes. One type of spreading code is known as a maximal-length sequence or an m-sequence for short. This is also called a linear feedback shift register (LFSR) sequence. This might also be known as a pseudo-noise (PN) sequence. Therefore, such a generator may be known as a pseudo-random sequence generator (PRBS).
The sequence of output of the far-right element (605) is the m-sequence. This sequence has a number of properties. The generated sequence, xn, is a periodic sequence. This is because there are a finite number of possible contents (states) of the shift register 600. Every time the shift register 600 is in a certain state, the sequence of states subsequently is identical. If the shift register 600 is ever in an all-zero state (where every element contains the binary value 0) then the shift register 600 will remain in the all zero state and the output will be a binary value 0 always. If there are m elements in the shift register 600, then it is possible for the state to be one of 2m possible values. However, as indicated above, the all zero state produces a trivial sequence of all zeros. There are 2m−1 possible nonzero states. If the shift register 600 goes through all these states before repeating, then the sequence will repeat with period Lc=2m−1. This sequence is known as a maximal length sequence because it is the maximum length sequence that can be produced by a shift register 600 of length m. The method to generate the sequence, illustrated in
In
Often, the sequence of 1s and 0s is converted to a binary sequence of +1's and −1s. There are two possible mappings, either 0→+1, 1→−1 or 0→−1, 1→+1. If the former is used, then adding modulo 2 in the 0, 1 domain is the same as multiplication in the +1, −1 domain. If the sequence generated consisting of 1s and 0s is converted to +1 s and −1s and called un, n=0, 1, 2, . . . , then the periodic autocorrelation of the sequence un has a two-level property. The periodic autocorrelation of the sequence un is:
where either the sequence is an infinite length periodic sequence with period Lc or the index on un+l is determined modulo Lc. The autocorrelation function is the correlation between the sequence un and the same sequence with offset l, namely un+l. Note that in the case where the sequence contains complex numbers, un* is the complex conjugate of un. The periodic autocorrelation θu(l) of an m-sequence (converted to a +1, −1 sequence) is Lc (mod Lc), if =0 and θu(l)=−1, if l(mod Lc)=1, 2, . . . , N−1. This is illustrated in
Another autocorrelation function is the aperiodic autocorrelation function. This is defined as
This autocorrelation is important when the signal transmitted consists of a single period of a spreading code, as opposed to multiple periods of a spreading code. One important spreading code with good aperiodic autocorrelation is the Barker code. There are a limited number of possible Barker codes of lengths up to Lc=13. These are shown in the Table below. The aperiodic autocorrelation of Barker codes is no larger in absolute value than 1. The aperiodic autocorrelation is illustrated in
Another class of codes is the Frank-Zadoff-Chu codes. Unlike Barker codes or m-sequences, which are binary codes, the Frank-Zadoff-Chu codes are sequences of complex numbers, where each complex number is on the unit circle. That is, a plot of each sequence element on the complex plane (real part is the horizontal axis, imaginary part is the vertical axis) lies on the unit circle. The codes can be defined when Lc is even, as
for when Lc is odd. The value of the periodic autocorrelation function θu(l) is Lc for l mod (Lc)=0, and is 0 for l mod (Lc)≠0. This is the ideal autocorrelation. Note that, in general for complex sequences, the autocorrelation function is a complex number which contains a real part and an imaginary part. Two real correlations are needed to compute the real part of the autocorrelation and two real correlations are needed to compute the imaginary part of the correlation. Generally, the received signal after down-conversion and sampling is a complex number. For real transmitted sequences, two correlations are needed: a correlation of the real transmitted sequence with the real part of the received sequence and a correlation of the real transmitted sequence with the imaginary part of the received sequence. For FZC sequences, the periodic autocorrelation function is (ideally) purely real. That is, the imaginary part is zero. The real part of the periodic autocorrelation is illustrated in
where j=√{square root over ((−1))}.
There are various other codes, including Golay codes, almost perfect autocorrelation sequences (APAS), and Hadamard codes. Codes with good autocorrelation functions are desired in a radar system. Complex codes require more storage than binary codes. Long codes are generally desirable because the autocorrelation function is generally better, the longer the code. Generating codes at a high chip rate is also desirable. Furthermore, codes that are unpredictable are also desirable.
Codes used for different transmitters that have low cross-correlation are desirable. The periodic cross-correlation function between a code sequence ul, l=0, 1, . . . , Lc−1 and a code sequence vl, l=0, 1, . . . , Lc−1 is defined as:
where again either the code sequence repeats or the index for the sequences are determined modulo Lc. One class of codes with interesting cross-correlation properties is known as Gold codes. The Gold codes can be formed from two m-sequences of the same length.
Another set of codes are the Hadamard codes, which can be defined recursively when the length is a power of 2. Namely
The rows in a Hadamard matrix are orthogonal. That is, the periodic cross correlation between two sequences u and v is 0, at offset 0; θu,v(0)=0. This is useful when different rows of a Hadamard matrix are used as the spreading codes. However, the autocorrelation of the codes is not good. A combination of codes is possible that have good autocorrelation function properties, and are orthogonal as well, when the sequences are synchronized at different transmitters.
In one embodiment, an exemplary spreading code is generated from several constituent codes. This is illustrated in
One way to combine different sequences is via a Kronecker product operation. The Kronecker product of a sequence x1, x2, x3 that is of length 3, with a sequence y1, y2, that is of length 2, is the sequence x1y1, x2y1, x3y1, x1y2, x2y2, x3y2, that is of length 6.
While it is possible to use a conventional general-purpose processor to generate any possible code, the processor would need to operate with a clock cycle several times as fast as the chip rate. A chip rate of 1 Gchips/second would require a processor that is difficult or costly to implement. In one embodiment illustrated in
The programmable code generator 1300 contains preferably 48 shift registers of 31 memory elements each, that can be programmed to generate any desired feedback connection, as well as loaded with any seed sequence. The shift registers 1303 can be programmed to generate an m-sequence of length 231−1. The feedback shift registers 1303 can also be shortened in length via feeding back from any set of desired elements in the register which might not include the last element. In this implementation, the last element and potentially other elements, become pure delay elements. The feedback shift register generators can be loaded with any contents (sometimes called the seed) and generators can be configured via software to have any set of feedback connections desired. The structure of the programmable code generator 1300 also has memory 1302 that can store chips used in an arbitrary sequence up to the size of the memory 1302. For example, the memory 1302 could store Hadamard sequences that are read by the processor 1301 and then combined with shift register sequences in the Hadamard processing block 1306. The memory 1302 can store actual chip sequences for different transmitters. One mode of operation, known as range domain MIMO, uses the full memory to provide a first chip sequence for one transmitter. The sequences for the other transmitters are then delayed versions of the first chip sequence. In one embodiment, the memory 1302 has the storage capacity to allow for 12 sequences of length 192K chips.
The memory 1302 can store the seeds and taps to control the feedback shift registers 1303. The processor 1301 can pick a random seed for the shift registers 1303 to choose different sequences instead of deterministically traversing the memory 1302 in a fixed sequence.
The combinatorial logic 1304 can be used for a variety of purposes. In one embodiment, the combinatorial logic 1304 performs exclusive OR (XOR) operations on different streams and can also be used to define which shift register stream goes to which transmitter. Alternatively, one code may be forwarded to all transmitters.
In one embodiment, the DC balancer 1305 first tracks the DC balance. The DC balance is the sum of the quantity of chips that take a value=1 in the past/chips, minus the sum of the quantity of chips that take a value=0 over the same interval. The DC balancer 1305 can be configured to control a maximum absolute value of the DC balance a chip stream may have, called the DC threshold. The DC threshold is configurable but preferably 20. Once the DC balance reaches the threshold the DC balancer will flip the next chip if that chip would cause the DC balance to exceed the DC threshold. Otherwise the DC balancer need not flip the next bit and the DC balance will improve. If the chip sequence is a finite length sequence then the DC balancer can force the DC balance to zero at the end of the sequence by forcing the appropriate last number of bits (e.g. 20) so that the DC balance at the end is zero. Alternatively, the DC balancer can slowly decrease the maximum DC balance threshold when the number of chips left in the finite sequence is smaller than some value. For example, for a sequence of length 512 with a maximum DC balance of 20 when there are 60 chips left in the sequence the DC balancer can reduce the maximum DC balance by 1 every three chips. So that by the time there are only 3 chips left the DC balance is at most 1. At that point, it can reduce the DC balance to zero by forcing the last chip to a value that causes the DC balance to 0 over the whole sequence.
The DC balancer 1305 may also flip an entire sequence of bits, or may take the previous sequence of bits over a range of bits, flip all the bits in the sequence and then used the flipped bits as the spreading code. The DC balancer can also take the flipped bits and do a permutation of the sequence of flipped bits and transmit that sequence. The DC balancer may insert either zeros or ones (0s, 1s) into any of the chip sequences as well.
The DC balancer 1305 may also balance the code across different transmitters over a configured interval. The DC balancer with multiple transmitters checks the DC balance across the different transmitters. If the absolute DC balance across transmitters exceeds a threshold then the DC balancer can change a chip in one of the transmitters. For example, if the DC balance across all transmitters for an individual chip in each of the transmitters exceeds the maximum value then the individual sequence with the worst DC balance in that individual sequence for which changing that chip can improve the DC balance across transmitters and improve the DC balance for the individual transmitter will have that chip flipped in sign. For example, suppose that there are eight transmitters and the individual DC balance for these transmitters is 2, −6, 8, −18, 12, −2, 3, 1 and for a particular chip the DC balance across transmitters is above a threshold, such as the eight chips are 0, 0, 1, 0, 0, 1, 0, 1. Then the DC balancer would need to change one of these chips from a 0 to a 1 in order to have a better balance. One of the transmitters that has a chip of 0 would need to change to 1 in order to improve the DC balance across chips. Transmitters 1, 3, 5, 7, and 8 have more ones than zeros while transmitters 2, 4, and 6 have more zeros than ones. The transmitter that would be changed would flip the bit would be transmitter 5 which would change the balance across transmitters to 2, −6, 8, −18, 10, −2, 3, 1 because what was a one before is removed and an addition zero is added to the balance for transmitter 5. While this will work to keep the DC balance across transmitters when the threshold for balancing an individual transmitter is reduced near the end of a sequence, the balance across transmitters can be eliminated because simultaneous balancing might require look ahead for the chip sequence.
This application is a continuation of U.S. patent application Ser. No. 15/892,865, filed Feb. 9, 2018, now U.S. Pat. No. 10,670,695, which claims the filing benefits of U.S. provisional application, Ser. No. 62/457,394, filed Feb. 10, 2017, which are both hereby incorporated by reference herein in their entireties.
| Number | Date | Country | |
|---|---|---|---|
| 62457394 | Feb 2017 | US |
| Number | Date | Country | |
|---|---|---|---|
| Parent | 15892865 | Feb 2018 | US |
| Child | 16884155 | US |
