UNIVERSAL CMOS CURRENT-MODE ANALOG FUNCTION SYNTHESIZER

Abstract

The universal CMOS current-mode analog function synthesizer is based on approximating the required function using its sixth-order Taylor series expansion. These approximations can be implemented by adding the weighted output currents of a number of basic building blocks built around a basic current squarer, and a constant current. The circuit can simultaneously realize thirty-two different mathematical functions and can be easily expanded to accommodate many others.

Description

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to analog function synthesizers, and particularly to a universal CMOS current mode analog function synthesizer that can realize over thirty-two nonlinear functions using a transistor squaring unit without dedicated current multipliers.

2. Description of the Related Art

The use of analog nonlinear networks, signal processing and medical equipment justifies the large number of analog nonlinear function synthesizers available in the literature. Over the years, different approaches have been used for synthesizing nonlinear functions. Initially, diodes and linear resistors were extensively used. Later on, the nonlinear characteristics of MOSFETs operated in weak or strong inversion regions and JFETs operated in the pinch-off or the triode region have been exploited in realizing various analog functions including the exponential, the squarer and the square rooting functions. Recently, the realization of nonlinear functions using piecewise-linear function approximations and integrated-circuit operational amplifiers, current conveyors, operational transconductance amplifiers and current comparators has been reported.

The exponential characteristic of the bipolar junction transistor has been exploited to advantage in the design of analog nonlinear functions using the translinear principle. Design of analog BiCMOS computational circuits has also been reported.

Over the years analog CMOS circuits have evolved based on the exponential law characteristic of a MOS transistor operating in weak inversion. Moreover, the square-law characteristic of a MOS transistor operating in strong inversion has been reported. Voltage multipliers, linear voltage-to-current and current-to-voltage converters, exponential and pseudo exponential current-to-voltage, voltage-to-current, voltage-to-voltage converters, vector-summation circuits, sin(x) shapers, square-rooters, arc sine function and arc cosine function are several examples of the analog nonlinear CMOS circuit realizations available in the literature. These circuit realizations suffer from many disadvantages. For example, the related art circuit realizations may permit only one function realization at a time. They may require numerical optimization routines to select the device size ratios and the bias voltages. They may use piecewise linear approximations for synthesizing the nonlinear functions. They may require programming of the parameters of several circuits. They may operate in voltage-mode with input and output voltages or mixed-mode with voltages as the input or output and current as output or input.

Current-mode circuits, with currents as input and output variables, are more attractive than their voltage-mode counterparts. This is attributed to wider signal bandwidths and larger dynamic ranges of operation that can be obtained using current-mode circuits rather than voltage-mode circuits.

Although a number of CMOS current-mode analog function synthesizers are available these circuits suffer from disadvantages such as, e.g., they may use piecewise linear approximations for synthesizing nonlinear functions; they may provide only a few functions (mostly the exponential or the pseudo exponential functions); they may extensively use integrated circuits such as operational transconductance amplifiers and current comparators; they may require digital control circuits to select the required function; they may realize only one function at a time.

Recently, a universal CMOS current-mode analog function synthesizer has been proposed. The key idea of the proposed circuit is the fact that numerous nonlinear functions can be approximated, to a high degree of accuracy, using a few terms of their Taylor series expansion.

Although a number of dedicated current multipliers are already available, current multipliers usually suffer from limited bandwidth, complexity leading to high power consumption, the need to trim out the feed-through terms (offset currents) and to adjust the scale factor (the multiplier gain).

It therefore would be desirable to present a new universal CMOS current-mode analog function synthesizer that can realize a wide range of nonlinear functions without recourse to dedicated current multipliers.

Thus, a universal CMOS current-mode analog function synthesizer solving the aforementioned problems is desired.

SUMMARY OF THE INVENTION

The universal CMOS current-mode analog function synthesizer is based on approximating the required function using a sixth-order Taylor series expansion of the function. These approximations can be implemented by adding the weighted output currents of a number of basic building blocks, built around a basic current squarer, and a constant current. The circuit can simultaneously realize thirty-two different mathematical functions and can be easily expanded to accommodate many others. Simulation results have verified the accuracy and performance of the synthesizer circuit.

The synthesizer circuit enjoys the following attractive features: (1) the circuit uses CMOS transistors and is, therefore, compatible with the current digital signal processing CMOS technology; (2) CMOS transistors work in the strong inversion, thus operation at relatively high frequency is feasible; (3) the circuit can simultaneously realize several nonlinear functions; (4) the circuit does not require any programming for its bias voltages or currents and parameter optimization is not required; (5) the circuit avoids the use of piecewise linear approximation of the synthesized nonlinear functions; and (6) the circuit operates in current-mode, thus providing higher frequencies of operation and wider dynamic ranges.

These and other features of the present invention will become readily apparent upon further review of the following specification and drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a class AB current mirror and its modification to provide current squarer and current proportional to the input current of a universal CMOS current-mode analog function synthesizer according to the present invention.

FIG. 2 is a block diagram for realizing the functions

$x,x^2,\frac{1}{2}x^3,\frac{1}{8}x^4,\frac{1}{2}x^5\mathrm{and}\frac{1}{8}x^6.$

FIG. 3 is a block diagram of a squaring circuit.

FIGS. 4A, 4B, 4C, and 4D together form a block diagram of a 32-by-32 function generator circuit.

FIG. 5 is a block diagram of a y=1/(1−x) portion of the 32-by-32 function generator circuit.

FIG. 6 is a block diagram of a y=cos(x) portion of the 32-by-32 function generator circuit.

FIG. 7 is a block diagram of a y=ln(1+x) portion of the 32-by-32 function generator circuit.

FIG. 8 is a plot showing calculated and simulated results obtained for the output current synthesizing y=x.

FIG. 9 is a plot showing calculated and simulated results obtained for the output current synthesizing (y=−x²/2).

FIG. 10 is a plot showing calculated and simulated results obtained for the output current synthesizing (y=−x³/2).

FIG. 11 is a plot showing calculated and simulated results obtained for the output current synthesizing (y=−x⁴/8).

FIG. 12 is a plot showing calculated and simulated results obtained for the output current synthesizing (y=x⁵/2).

FIG. 13 is a plot showing calculated and simulated results obtained for the output current synthesizing (y=x⁶/8).

FIG. 14 is a plot showing calculated and simulated results obtained for the output current synthesizing y=(1−x)⁻¹.

FIG. 15 is a plot showing calculated and simulated results obtained for the output current synthesizing y=cos(x).

FIG. 16 is a plot showing calculated and simulated results obtained for the output current synthesizing function y=ln(1+x).

Similar reference characters denote corresponding features consistently throughout the attached drawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The universal CMOS current-mode analog function synthesizer (shown in FIGS. 4A through 4D as circuitry 400a through 400d) is based on approximating the required function using a sixth-order Taylor series expansion of the function. These approximations can be implemented by adding the weighted output currents of a number of basic building blocks, which are built around a basic current squarer and a constant current. The circuitry 400a through 400d can simultaneously realize thirty-two different mathematical functions and can be easily expanded to accommodate many others.

The universal CMOS current-mode analog function synthesizer circuitry 400a-400d approximates numerous nonlinear functions to a high degree of accuracy by using the first seven terms of a Taylor series expansion of the form:

f(x)≅y=a₀+a₁x+a₂x²+a₃x³+a₄x⁴+a₅x⁵+a₆x⁶; |x|<1  (1)

TABLE I
Node
no	function	a0	a1	a2	a3	a4	a5	a6
201	$\frac{1}{1-x}$	1	1	1	1	1	1	1
202	$\frac{1}{1+x}$	1	−1	1	−1	1	−1	1
203	$\frac{1}{1-x^2}$	1	0	1	0	1	0	1
204	$\frac{1}{1+x^2}$	1	0	−1	0	1	0	−1
205	{square root over (1−x)}	1	$-\frac{1}{2}$	$-\frac{1}{8}$	$-\frac{1}{16}$	$-\frac{5}{128}$	$-\frac{7}{256}$	$-\frac{21}{1024}$
206	{square root over (1+x)}	1	$\frac{1}{2}$	$-\frac{1}{8}$	$\frac{1}{16}$	$-\frac{5}{128}$	$\frac{7}{256}$	$-\frac{21}{1024}$
207	$\frac{1}{\sqrt{1-x}}$	1	$\frac{1}{2}$	$\frac{3}{8}$	$\frac{5}{16}$	$\frac{35}{128}$	$\frac{63}{256}$	$\frac{231}{1024}$
208	$\frac{1}{\sqrt{1+x}}$	1	$-\frac{1}{2}$	$\frac{3}{8}$	$-\frac{5}{16}$	$\frac{35}{128}$	$-\frac{63}{256}$	$\frac{231}{1024}$
209	{square root over (1−x2)}	1	0	$-\frac{1}{2}$	0	$-\frac{1}{8}$	0	$-\frac{19}{240}$
210	{square root over (1+x2)}	1	0	$\frac{1}{2}$	0	$-\frac{1}{12}$	0	$\frac{1}{24}$
211	sin(x)	0	1	0	$-\frac{1}{6}$	0	$\frac{1}{120}$	0
212	tan(x)	0	1	0	$\frac{1}{3}$	0	$\frac{2}{15}$	0
213	cos(x)	1	0	$-\frac{1}{2}$	0	$\frac{1}{24}$	0	$-\frac{1}{720}$
214	sinh(x)	0	1	0	$\frac{1}{6}$	0	$\frac{1}{120}$	0
215	tanh(x)	0	1	0	$-\frac{1}{3}$	0	$\frac{2}{15}$	0
216	cosh(x)	1	0	$\frac{1}{2}$	0	$\frac{1}{24}$	0	$\frac{1}{720}$
217	sin−1(x)	0	1	0	$\frac{1}{6}$	0	$\frac{3}{40}$	0
218	tan−1(x)	0	1	0	$-\frac{1}{3}$	0	$\frac{1}{5}$	0
219	cos−1(x)	$\frac{\pi }{2}$	−1	0	$-\frac{1}{6}$	0	$-\frac{3}{40}$	0
220	sec(x)	1	0	$\frac{1}{2}$	0	$\frac{5}{24}$	0	$\frac{61}{720}$
221	sec(x2)	0	0	1	0	0	0	$-\frac{1}{6}$
222	cos(x2)	1	0	0	0	$-\frac{1}{2}$	0	0
223	J0(x)	1	0	$-\frac{1}{4}$	0	$\frac{1}{64}$	0	$-\frac{1}{2304}$
224	J1(x)	0	$\frac{1}{2}$	0	$\frac{1}{16}$	0	$\frac{1}{384}$	0
252	I0(x)	1	0	$\frac{1}{4}$	0	$\frac{1}{64}$	0	$\frac{1}{2304}$
226	ln(1 − x)	0	−1	$-\frac{1}{2}$	$-\frac{1}{3}$	$-\frac{1}{4}$	$-\frac{1}{5}$	$-\frac{1}{6}$
227	ln(1+ x)	0	1	$-\frac{1}{2}$	$\frac{1}{3}$	$-\frac{1}{4}$	$\frac{1}{5}$	$-\frac{1}{6}$
228	$\ln \left[\frac{1+x}{1-x}\right]$	0	2	0	$\frac{2}{3}$	0	$\frac{2}{5}$	0
229	ex	1	1	$\frac{1}{2}$	$\frac{1}{6}$	$\frac{1}{24}$	$\frac{1}{120}$	$\frac{1}{720}$
230	e−x	1	−1	$\frac{1}{2}$	$-\frac{1}{6}$	$\frac{1}{24}$	$-\frac{1}{120}$	$\frac{1}{720}$
231	ex2	1	0	1	0	$\frac{1}{2}$	0	$\frac{1}{6}$
232	e−x2	1	0	−1	0	$\frac{1}{2}$	0	$-\frac{1}{6}$

Table I shows a number of nonlinear functions with the corresponding values of the parameters a_n, n=0, 1, 2, . . . , 6. In current mode, with the variable x representing the normalized input current, Equation (1) can be implemented by adding the weighted output currents of a number of power-factor raising circuits with power factors=2, 3, . . . 6, current amplifiers (or attenuators) and a constant current. In the function synthesizer circuit shown in FIGS. 4A through 4D, the power-factor raising circuits and the current amplifiers (or attenuators) are constructed via successive use of the traditional class-AB current mirror 100, shown in FIG. 1.

Table II shows the aspect ratios (W/L) of transistors M₁-M₁₀ of current mirror 100.

TABLE II
Transistor W/L
M1         1/1
M2         1/1
M3         1/1
M4         1/1
M5         1/1
M6         1/1
M7         1/1
M8         1/1
M9         1/2
M10        1/1

Assuming that transistors M₁ M₂ as well as transistors M₃ and M₄ are well matched and that all transistors are operating in their saturation region and having the same value of the process transconductance parameter, i.e., β_n=β_p, then applying the translinear principle, we obtain:

2√{square root over (I_q)}=√{square root over (I_D2)}+√{square root over (I_D4)}  (2)

where the currents I_D2 and I_D4 are the drain currents of transistors M₂ and M₄, respectively. Combining Equation (2) with:

I _D2 +I _in =I _D4  (3)

and using simple mathematical manipulations, the currents I_D2 and I_D4 can be expressed as:

$\begin{array}{cc}\frac{I_{D2}}{I_q}=1-\frac{1}{2}\frac{I_{\mathrm{in}}}{I_q}+{\left(\frac{I_{\mathrm{in}}}{4I_q}\right)}^2\mathrm{and}& \left(4\right)\\ \frac{I_{D4}}{I_q}=1+\frac{1}{2}\frac{I_{\mathrm{in}}}{I_q}+{\left(\frac{I_{\mathrm{in}}}{4I_q}\right)}^2.& \left(5\right)\end{array}$

From Equations (4) and (5), the current I_D5 and the output current I_B can be expressed as:

$\begin{array}{cc}\frac{I_{D5}}{I_q}=\frac{I_{D2}+I_{D4}}{I_q}=2+\frac{1}{8}{\left(\frac{I_{\mathrm{in}}}{I_q}\right)}^2\mathrm{and}& \left(6\right)\\ \frac{I_B}{I_q}=\frac{\frac{1}{2}I_{D5}-I_{D4}}{I_q}=-\frac{1}{2}\frac{I_{\mathrm{in}}}{I_q}& \left(7\right)\end{array}$

By subtracting a constant current=2I_q from I_D5, the output current I_A can be expressed as:

$\begin{array}{cc}\frac{I_A}{I_q}=\frac{1}{8}{\left(\frac{I_{\mathrm{in}}}{I_q}\right)}^2& \left(8\right)\end{array}$

From Equations (7) and (8) it appears that the currents I_B and I_A are proportional to the normalized input-current

$x=\frac{I_{\mathrm{in}}}{I_q}$

and the square of the normalized input current

$x^2={\left(\frac{I_{\mathrm{in}}}{I_q}\right)}^2$

respectively.

As shown in FIG. 2, using additional current-mirrors with MOSFETs having different values of the aspect ratio W/L, these two currents can be used to obtain any current of value a₁x or a₂x² where a₁ and a₂ are constants. Moreover, a normalized current proportional to x³ can be obtained by using the square-difference identity └(A+B)²−(A−B)²=4AB┘. First, the sum and difference of the inputs A=x and B=x² are provided for the Squaring Unit (SU) derived from FIG. 1 and shown in FIG. 3.

The aspect ratios (W/L) of transistors M₁-M₈ of FIG. 3 are the same as shown in Table 2. W/L is a geometric property of the transistor defined as the channel width of the Field Effect transistor (W) divided by its channel length (L). The weighting factor, a₃, of the normalized output current proportional to x³ can be adjusted by proper selection of the aspect ratios (W/L) of additional MOSFET current mirrors.

Similarly, as shown in FIG. 2, a normalized current proportional to x⁵ can be obtained using the square-difference identity with normalized input currents proportional to x² and x³. The normalized current proportional to x⁴ can be obtained using a squaring circuit input current proportional to x² and finally a normalized current proportional to x⁶ can be obtained using another squaring circuit with normalized input current proportional to x³.

Normalized currents proportional to higher powers of the normalized input current with different weighting factors can be obtained by successive use of squares and the square-difference identity and appropriate current-mirrors. Thus, higher-order terms of Equation (1) can be obtained using the already available design of the SU derived from FIG. 1 and shown in FIG. 3, and without recourse to a dedicated current multiplier.

Table III shows the aspect ratios (W/L) of the MOSFETs used to obtain the normalized currents x, x³/2, x⁴/8, x⁵/2 and x⁶/8. Using these normalized currents in addition to the appropriate normalized DC current component, any function in Table I can be realized using MOSFET current-mirrors with the appropriate aspect ratios (W/L).

TABLE III
Transistor W/L
M1-M16     1/1
M21-M50    1/1
M53-M75    1/1
M77-M84    1/1
M17-M20    8/1
M51-M52    8/1
M76        2/1

A complete circuit realization for the thirty-two functions of Table I is shown in FIG. 4. To illustrate the described approach, FIG. 5 shows a circuit portion 500 of the sub-circuits used for realizing the function y=(1−x)⁻¹ wherein the current load accepted by R201 represents the Taylor Expansion for said function. Similarly, as shown in FIG. 6, circuit portion 600 realizes the function y=cos(x) wherein the current load accepted by R213 represents the Taylor Expansion for said function. Moreover, as shown in FIG. 7, circuit portion 700 realizes the function y=ln(1+x) wherein the current load accepted by R227 represents the Taylor Expansion for said function. Using Table 1, FIGS. 4-7 show the values of the aspect ratios (W/L) selected according to the values a_n, n=1, 2, . . . , 6 of the desired nonlinear functions.

The circuit of FIG. 2 was simulated with the HSPICE circuit simulation program using the BSIM2 level 39 MOSFET transistor models with L=0.1 μm, bias current I_q=1 μA and supply voltages V_DD=−V_SS=2V. With the input current changing between 0.0 μA and 1 μA, the output currents through load resistances R_L1=R_L2=100 kΩ were monitored. The results are shown in FIGS. 8-13 and Table IV.

TABLE IV
x	Exact Simulated	0	.2	.4	.6	.8	1
		.00008	.200008	.400008	.600008	.800008	1
−x2	Exact Simulated	0	−.04	−.16	−.36	−.64	−1
		−.00034	−.04035	−.16035	−.36036	−.64036	−1.0004
$-\frac{x^3}{2}$	Exact Simulated	0 −.000008	−.004 −.004045	−.032 −.03208	−.108 −.10812	−.256 −.25616	−.5 −.5002
$-\frac{x^4}{8}$	Exact Simulated	0 −.000053	−.0002 −.00026	−.0032 −.00326	−.0162 −.016285	−.0512 −.05131	−.125 −.12514
$\frac{x^5}{2}$	Exact Simulated	0 −.000017	.00016 .000182	.00512 .005198	.03888 .03905	.16384 .16415	.5 .5005
$\frac{x^6}{8}$	Exact Simulated	0 .000036	.000008 .000043	.000512 .00055	.00583 .00588	.03277 .03284	.125 .125

In FIGS. 8-13 all input and output currents are in microamperes. Inspection of FIGS. 8-13 and Table IV clearly shows that the simulated results are in excellent agreement with the calculated ones. For example, as shown in plot 800 of FIG. 8, the version of the function y=x synthesized by the present invention nearly identically matches the exact version of the function. As shown in plot 900 of FIG. 9, similar results were obtained for the function (y=−x²/2). As shown in plot 1000 of FIG. 10, the function (y=−x³/2) is faithfully synthesized by the present invention. Equally high fidelity results were obtained for synthesis of the function (y=−x⁴/8) as shown in plot 1100 of FIG. 11. Plot 1200 of FIG. 12 shows the present invention synthesizing the function (y=x⁵/2). Plot 1300 of FIG. 13 shows the present invention synthesizing the function y=x⁶/8.

The outputs of the circuit of FIG. 2 were used as inputs to the circuits of FIGS. 4-7. Samples of the results are shown in FIGS. 14-16 and a summary of the results is given Table V where the output is the voltage across a load resistance=1 MΩ. In FIGS. 14-16 all input currents are in microampere and all outputs are in volts. Whenever necessary, a DC current source=1 μA was added to the output node. This DC current source represents the constant term, a₀ of equation (1), which equals, according to Table I, either to 1 or zero. Inspection of plot 1400 in FIG. 14 clearly shows that for the function y=1/(1−x) the error is less than 1% for values of x≦0.5 corresponding to I_in≦0.5 μA and is less than 5% for values of x≦0.66 corresponding to I_in≦0.66 μA. Similarly, inspection of plot 1500 in FIG. 15 clearly shows that for the function y=cos(x) the error is less than 1% for values of x corresponding to I_in≦1.0 μA and inspection of plot 1600 in FIG. 16 shows that for the function y=ln(1+x) the error is less than 1% for x≦0.675 corresponding to I_in≦0.675 μA and less than 5% for values of x≦0.875 corresponding to I_in≦0.875μA. Similar results, shown in Table V, were obtained from the simulation of FIG. 4 for the thirty-two functions tabulated in Table I.

The outputs of the circuit of FIG. 2 were used as inputs to the circuits of FIGS. 4-7.

TABLE V
			Max
Outlet		Max input for	input for
number	Function	error = 1%	error = 5%
201	$\frac{1}{1-x}$	0.5	μA	.66	μA
202	$\frac{1}{1+x}$	0.5	μA	.66	μA
203	$\frac{1}{1-x^2}$	0.575	μA	.7	μA
204	$\frac{1}{1+x^2}$	0.575	μA	.7	μA
205	{square root over (1−x)}	0.725	μA	.875	μA
206	{square root over (1+x)}	1.0	μA	1.0	μA
207	$\frac{1}{\sqrt{1-x}}$	0.625	μA	.75	μA
208	$\frac{1}{\sqrt{1+x}}$	0.675	μA	.85	μA
209	{square root over (1−x2)}	0.785	μA	.875	μA
210	{square root over (1+x2)}	0.75	μA	1.0	μA
211	sin(x)	1.0	μA	1.0	μA
212	tan(x)	0.76	μA	0.975	μ A
213	cos(x)	1.0	μA	1.0	μA
214	sinh(x)	1.0	μA	1.0	μA
215	tanh(x)	1.0	μA	1.0	μA
216	cosh(x)	1.01	μA	1.0	μA
217	sin−1(x)	0.73	μA	0.9	μA
218	tan−1(x)	0.655	μA	0.875	μA
219	cos−1(x)	0.73	μA	0.845	μA
220	sec(x)	1.0	μA	1.0	μA
221	sin(x2)	1.0	μA	1.0	μA
222	cos(x2)	0.8	μA	0.96	μA
223	J0(x)	1.0	μA	1.01	μA
224	J1 (x)	1.0	μA	1.0	μA
252	10(x)	1.0	μA	1.0	μA
226	ln(1 − x)	0.625	μA	0.775	μA
227	ln(1 + x)	0.675	μA	0.85	μA
228	$\ln \left[\frac{1+x}{1-x}\right]$	0.625	μA	0.8	μA
229	ex	1.0	μA	1.0	μA
230	e−x	1 .0	μA	1.0	μA
231	ex2	0.925	μA	1.0	μA
232	e−x2	0.8	μA	0.95	μA

The SU plays a key role in the realization of the analog function synthesizer. Therefore, it is essential to investigate its performance in detail. The analysis presented above is based on the assumption that the transistors M₁-M₄, of the class-AB configuration of FIG. 3, are operating in saturation region, characterized by the perfect square-law equation, the process transconductance parameters K_n and K_p, are equal, and the threshold voltages V_thi, i=1-4 of the n- and p-channel MOSFET transistors are equal. It is well known that, for long channel MOS transistors, the perfect square-law equation agrees well with experimental results. This, however, may not be the case for small channel lengths due to the second-order effects, such as channel length modulation, body effect, velocity saturation and mobility reduction.

A minimum gate length, which largely avoids short channel effects due to mobility reduction and velocity saturation, is, therefore, recommended to avoid destroying the square-law equation. Consequently, this minimum gate length ultimately limits the maximum operating frequency of the analog function synthesizer. Mismatch between the process transconductance parameters K_n and K_p, the aspect ratios (W/L), and the threshold voltages V_thi, i=1-4 of the n- and p-channel MOS transistors of the class-AB configuration of FIG. 3 are, therefore, considered as the major contributors to the errors from the ideal performance. In the following subsections, the effects of the mismatch in threshold voltages, process transconductance parameters and the aspect ratios are considered in detail.

Assuming that transistors M₁-M₄ have threshold voltage mismatches, refinement of the translinear loop formed of M₁-M₄ of FIG. 3 yields

$\begin{array}{cc}2\sqrt{I_q}+\sum _{i=1}^4\left(\sqrt{K}\Delta V_{\mathrm{thi}}\right)=\sqrt{I_{D2}}+\sqrt{I_{D4}}& \left(9\right)\end{array}$

where K=K_n=K_p, I_D2 and I_D4 are the currents in the drains of transistors M₂ and M₄, respectively,

$\begin{array}{cc}\sum _{i=1}^4\Delta V_{\mathrm{thi}}=\Delta V_{\mathrm{th}1}+\Delta V_{\mathrm{th}3}-\Delta V_{\mathrm{th}2}-\Delta V_{\mathrm{th}4}\mathrm{and}& \left(10\right)\\ \Delta V_{\mathrm{thi}}=V_{\mathrm{thi}}-V_{\mathrm{th}0}& \left(11\right)\end{array}$

Combining Equations (3), (9)-(11) and using simple mathematical manipulations, the normalized output current of the SU can be expressed as

$\begin{array}{cc}\frac{I_A}{I_q}=\frac{1}{8}{\left(\frac{x}{I_q}\right)}^2+\frac{1}{2I_q}{\left(\sqrt{K}\sum _{i=1}^4\Delta V_{\mathrm{thi}}\right)}^2+\sqrt{\frac{2}{I_q}}\left(\sqrt{K}\sum _{i=1}^4\Delta V_{\mathrm{thi}}\right)-\frac{1}{32}{\left(\frac{x}{I_q}\right)}^2\frac{1}{I_q}{\left(\sqrt{K}\sum _{i=1}^4\Delta V_{\mathrm{thi}}\right)}^2-\frac{1}{4}{\left(\frac{x}{I_q}\right)}^2\sqrt{\frac{1}{2I_q}}\left(\sqrt{K}\sum _{i=1}^4\Delta V_{\mathrm{thi}}\right)& \left(12\right)\end{array}$

On the right-hand side of Equation (12), the first term represents the output current of the ideal SU; the second and third terms represent a DC offset output current and the fourth and fifth terms are proportional to the square of the input current and represent an error in the output of the SU. Using Equation (12), the amount of error can be estimated for different values of x. For example, for V_th0=0.2578V, ΔV_th1=1 mV, ΔV_th2=0.7 mV, ΔV_th3=0.3 mV, ΔV_th4=1.2 mV and K=4.0 μA/V², then for I_in=1 μA, I_q, =1 μA that is x=1.0, the output current of the SU is 0.2 μA, whereas the expected output current assuming identical transistors, with no threshold-voltage mismatch is 0.125 μA. This represents 60% error and shows that a total threshold-voltage mismatch ΣΔV_thi=0.3% results in 60% error in the output current of the SU. This clearly indicates the importance of minimizing the threshold-voltages mismatches of the MOSFET transistors.

Assuming that transistors M₁-M₄ have transconductance parameters mismatch with K_n≠K_p where K_n and K_p of the n- and p-channel MOS transistors, re-analysis of the translinear loop formed of transistors M₁-M₄ of FIG. 3 yields:

$\begin{array}{cc}\sqrt{\frac{I_q}{K_p}}+\sqrt{\frac{I_q}{K_n}}=\sqrt{\frac{I_{D2}}{K_n}}+\sqrt{\frac{I_{D4}}{K_p}}& \left(13\right)\end{array}$

With K_n=K, K_p=K(1+δ), using the approximation (1+x)^−1/2≈1−½x, if x<<1, ignoring terms containing δ², combining Equations (3) and (13), the output current of the SU can be expressed as:

I _D2(16(1−2Δ)I_q+4Δx)=16(1−2Δ)I_q²+(1−4Δ)x²−8(1−3Δ)I_qx  (14)

where Δ=0.5δ. Assuming that 4(1−2Δ)I_q>>Δx, equation (14) yields the following expression for the second-order output current:

$\begin{array}{cc}\frac{I_A}{I_q}\approx \left(\frac{1}{8}-\frac{\Delta }{4}\right){\left(\frac{x}{I_q}\right)}^2+\Delta \frac{x}{I_q}& \left(15\right)\end{array}$

Successive use of Equation (15), ignoring second-order effects, yields the following expressions for the third-, fourth-, fifth- and sixth-order normalized output currents:

$\begin{array}{cc}\frac{I_B}{I_q}\approx 4\left(\frac{1}{8}-\frac{\Delta }{4}\right){\left(\frac{x}{I_q}\right)}^3+2\Delta \frac{x}{I_q}& \left(16\right)\\ \frac{I_c}{I_q}\approx \left(\frac{1}{8}-\frac{\Delta }{4}\right){\left(\frac{x}{I_q}\right)}^4+{\Delta \left(\frac{x}{I_q}\right)}^2& \left(17\right)\\ \frac{I_D}{I_q}\approx 4\left(\frac{1}{8}-\frac{\Delta }{4}\right){\left(\frac{x}{I_q}\right)}^5+2\Delta \frac{x}{I_q}\mathrm{and}& \left(18\right)\\ \frac{I_E}{I_q}\approx \left(\frac{1}{8}-\frac{\Delta }{4}\right){\left(\frac{x}{I_q}\right)}^6+{\Delta \left(\frac{x}{I_q}\right)}^3& \left(19\right)\end{array}$

Equations (15)-(19) clearly show that the normalized output currents I_A, I_B, I_C, I_D, and I_E will be affected by the mismatch of the process transconductance parameters K_n and K_p. In fact each of the currents I_A, I_B, and I_D, are comprised of the required second-, third- and fifth-order components plus a linear one. And the currents I_C and I_E are comprise of the required fourth- and sixth-order components plus a second-order and a third-order component, respectively. Obviously, this will lead to error in the synthesized analog function.

To illustrate the amount of error, consider the following example. To synthesize the function:

$\begin{array}{cc}\begin{array}{c}y={}^x\\ \approx 1+x+\frac{1}{2}x^2+\frac{1}{6}x^3+\frac{1}{24}x^4+\frac{1}{120}x^5+\frac{1}{720}x^6\end{array}& \left(20\right)\end{array}$

the current I_A must be inverted and multiplied by 4, and the current I_B must be divided by 3, the current I_C must be divided by 3, the current I_D must be divided by 60 and the current I_E must be divided by 90. Thus, using Equations (15)-(19), and assuming that the current I_q=1 uA, the output current will be given by:

$\begin{array}{cc}y\approx 1+x\left(1+4.7\Delta \right)+\left(\frac{1}{2}-\frac{2}{3}\Delta \right)x^2+\left(\frac{1}{6}-\frac{13}{90}\Delta \right)x^3+\left(\frac{1}{24}-\frac{\Delta }{12}\right)x^4+\left(\frac{1}{120}-\frac{\Delta }{240}\right)x^5+\left(\frac{1}{720}-\frac{\Delta }{360}\right)x^6& \left(21\right)\end{array}$

For any value of Δ, using (21), the amount of error can be estimated for different values of x. For example, for Δ=0.05, that is 10% transconductance mismatch, x=0.2, the output will be 1.261139, that is e^0.2≈1.261139. The exact value is 1.2214. Thus, the error due to the transconductance mismatch is 3.25%.

Assuming that transistors M₁-M₄ have aspect ratios mismatch with (W/L)₁—(W/L)₂≠(W/L)₃≠(W/L)₄, re-analysis of the translinear loop formed of M₁-M₄ in FIG. 3 yields:

$\begin{array}{cc}\sqrt{\frac{I_q}{{\left(W/L\right)}_1}}+\sqrt{\frac{I_q}{{\left(W/L\right)}_3}}=\sqrt{\frac{I_{D2}}{{\left(W/L\right)}_2}}+\sqrt{\frac{I_{D4}}{{\left(W/L\right)}_4}}& \left(22\right)\end{array}$

With (W/L)_i=(W/L)(1+δ_i), i=1, 2, 3, 4, using the approximation (1+x)^−1/2≈1−½x, if x<<1, ignoring terms containing δ_i², Equation (22) reduces to

$\begin{array}{cc}\sqrt{I_q}\left(1-\frac{1}{2}{\delta }_1\right)+\sqrt{I_q}\left(1-\frac{1}{2}{\delta }_3\right)=\sqrt{I_{D2}}\left(1-\frac{1}{2}{\delta }_2\right)+\sqrt{I_{D4}}\left(1-\frac{1}{2}{\delta }_4\right)& \left(23\right)\end{array}$

Combining Equations (3) and (23), simple mathematical manipulations lead to the following approximate expression for the output current of the SU, after subtracting the current 2I_q.

$\begin{array}{cc}\frac{I_A}{I_q}=\frac{1}{{\left(\frac{{\alpha }_1}{{\alpha }_3}\right)}^2+{\left(\frac{{\alpha }_2}{{\alpha }_3}\right)}^2}-2+\frac{{\left(\frac{{\alpha }_2}{{\alpha }_3}\right)}^4}{\left[{\left(\frac{{\alpha }_1}{{\alpha }_3}\right)}^2+{\left(\frac{{\alpha }_2}{{\alpha }_3}\right)}^2\right]}{\left(\frac{x}{I_q}\right)}^2+\frac{{\left(\frac{{\alpha }_1}{{\alpha }_3}\right)}^2-{\left(\frac{{\alpha }_2}{{\alpha }_3}\right)}^2}{{\left(\frac{{\alpha }_1}{{\alpha }_3}\right)}^2+{\left(\frac{{\alpha }_2}{{\alpha }_3}\right)}^2}\left(\frac{x}{I_q}\right)& \left(24\right)\end{array}$

where =α1=(1−½δ2), α2=(1−½δ4) and α3=2−½(δ1+δ3). On the right hand side of Equation (24), the first two terms represent a DC offset in the normalized output current of the SU, the third term represents the required output of the SU; now modified, and the fourth term is proportional to the normalized input current and represents an error in the output of the SU. For any values of δ1, δ2, 67 3 and δ4 the error can be estimated for different values of x. For example, for δ1=0.1, δ2=0.05, δ3=0.07 and δ4=0.06, then the normalized output current of the SU will have an undesired DC offset component equal to 0.1 instead of zero in the ideal case, the coefficient of the square of the normalized output current will be 1/7.9997 instead of the exact value=⅛ and an undesired output component proportional to the normalized input with coefficient=0.0208. This clearly indicates that the output of the SU will be slightly affected by the mismatches in the aspect ratios W/L.

A simple technique for synthesizing nonlinear functions has been presented. This technique is based on approximating the nonlinear function of interest using the first seven terms in its Taylor-series expansion. The desired function is the summation of the weighted output currents of a number of power-factor raising circuits built around a basic current squarer circuit (SU), a weighted current amplifier (or attenuator) and a DC current component. The technique is very flexible and can simultaneously realize many functions by proper selection of the aspect ratios of MOSFET transistors. Adding higher-order terms to the Taylor series requires the use of additional power-factor raising circuits, which can be easily realized using the current squarer circuit (SU). The simulation results, obtained from the thirty-two functions, verified the operation of the circuit.

The accuracy of the synthesized functions will be primarily decided by the number of Taylor-series terms used in the approximation and the effects of mismatch between transistors used in practical implementation of the required current power-factor raising circuits built around the SU of FIG. 3 and the current-mirrors. The effect of threshold-voltage mismatches, process transconductance parameters mismatches, and aspect ratios mismatches have been studied in detail. The results clearly show that the threshold-voltage mismatches are the major contributors to the errors in the realization of the nonlinear functions reported in this article. Monte Carlo simulations performed on all the thirty-two functions confirmed this conclusion. Moreover, Monte Carlo simulations performed on the thirty-two functions clearly show that for most of the functions, the change of temperature from −35° C. to 85° C. will have a slight effect on the performance of the nonlinear function synthesizer. Also, the change of DC power supply within ±10% has a negligible effect on the performance of the circuit.

The high frequency limitations of the analog function synthesizer will be decided by the high frequency performance of the class-AB configuration of FIGS. 1 and 3 and the current mirrors. Approximate analysis shows that the dominant high-frequency pole of the class-AB configuration is around 175.36 MHz and the dominant high-frequency pole of the current mirror is around 28.43 MHz. It appears, therefore, that the high frequency performance of the analog function synthesizer (400a, 400b, 400c, 400d) is expected to be around 25 MHz, which is appropriate for many applications.

It is to be understood that the present invention is not limited to the embodiment described above, but encompasses any and all embodiments within the scope of the following claims.

Claims

1. A universal CMOS current-mode analog function synthesizer, comprising: a current squarer circuit;power factor raising circuits connected to the current squarer circuit;a DC current source; anda weighted current amplifier connected to the DC current source, the power factor raising circuits, and the current squarer circuit, the weighted current amplifier summing outputs of the DC current source, the current squarer circuit and the power factor raising circuits, the weighted current amplifier having weights representing successive terms of a Taylor series expansion of a nonlinear function of interest;wherein a current output of the weighted current amplifier approximates the nonlinear function of interest using the weights representing the plurality of terms in the Taylor-series expansion of the function.

2. The universal CMOS current-mode analog function synthesizer according to claim 1, further comprising: a plurality of CMOS current squaring circuits arranged in rows, each row having a user selectable width-to-length ratio (W/L); anda plurality of nodes, each node selectively terminating a unique one of said rows, said nodes and said rows forming a function matrix wherein a plurality of functions are independently and concurrently realizable in said circuitry.

3. The universal CMOS current-mode analog function synthesizer according to claim 1, wherein the current squarer circuit has a first output current proportional to a normalized input current, and a second output current proportional to the square of the normalized input current.

4. The universal CMOS current-mode analog function synthesizer according to claim 1, wherein the current squarer circuit comprises a plurality of MOSFETS having a plurality of aspect ratios (w/L) selected to obtain any current of value a1x or a2x2, a1 and a2 being constants.

5. The universal CMOS current-mode analog function synthesizer according to claim 4, further comprising square-difference identity circuitry using the square-difference identity └(A+B)2−(A−B)2=4AB┘, having an input which accepts a sum and difference of inputs A=x and B=x2, and having an output which provides a normalized current proportional to x3.

6. The universal CMOS current-mode analog function synthesizer according to claim 5, further comprising pre-selected aspect ratios (W/L) in the associated square-difference identity circuitry, wherein a weighting factor a3 of the normalized output current proportional to x3 is adjustable.

7. The universal CMOS current-mode analog function synthesizer according to claim 4, further comprising square-difference identity circuitry using the square-difference identity └(A+B)2−(A−B)2=4AB┘, having an input which accepts a sum and difference of inputs A=x2 and B=x3, and having an output which provides a normalized current proportional to x5.

8. The universal CMOS current-mode analog function synthesizer according to claim 4, wherein an input to said current squarer circuit comprises a current proportional to x2, thereby producing a normalized output current proportional to x4.

9. The universal CMOS current-mode analog function synthesizer according to claim 4, wherein an input to said current squarer circuit comprises a current proportional to x3, thereby producing a normalized output current proportional to x6.

10. The universal CMOS current-mode analog function synthesizer according to claim 4, further comprising: successive stages of said current squarer circuit; andsquare-difference identity circuitry having successive stages, said successive stages of said square-difference identity circuitry being connected in combination with said successive stages of said squaring circuit, said combination having an output providing a normalized output current proportional to an arbitrary power of a normalized input current, each said normalized output current having a user selectable weighting factor without recourse to the use of a dedicated current multiplier.