BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of components included in the range of a wafer bias RF power supply to an electrode;
FIG. 2 is an equivalent circuit diagram for the block diagram shown in FIG. 1;
FIG. 3 is a diagram showing frequency characteristics in the configuration shown in FIG. 1;
FIG. 4 shows a result of simulation conducted by using the equivalent circuit shown in FIG. 2;
FIG. 5 is an equivalent circuit diagram for a range from the wafer bias RF power supply to plasma;
FIG. 6 shows a result of simulation conducted by using the equivalent circuit shown in FIG. 5;
FIG. 7 is a frequency characteristic diagram obtained when Vpp on the electrode is set equal to a constant voltage of 20 V;
FIG. 8 is an equivalent circuit diagram obtained when a Vpp detector is incorporated into the electrode;
FIG. 9 is a schematic diagram showing a first embodiment of a plasma etching apparatus;
FIG. 10 is a schematic diagram showing a second embodiment of a plasma etching apparatus;
FIG. 11 is an equivalent circuit diagram for the configuration shown in FIG. 10;
FIGS. 12A-12C show results of simulation conducted by using the equivalent circuit shown in FIG. 11;
FIG. 13 is an equivalent circuit diagram obtained when a phase detector is incorporated into the electrode;
FIG. 14 is an equivalent circuit diagram obtained when the phase detector is disposed outside the electrode;
FIGS. 15A and 15B show results of phase difference simulation conducted using the equivalent circuit shown in FIG. 14; and
FIG. 16 is a schematic diagram showing a plasma etching apparatus according to a third embodiment.
DETAILED DESCRIPTION OF THE INVENTION
As described above, the resonances do not disappear, and correction using a calculation or calibration cannot be conducted. Therefore, it is appreciated that it is important in attaining the object to configure the apparatus so as to cause the voltage and phase information at the measurement point to be equivalent to or have the same quality as the voltage and phase information at an electrode of the measurement subject (an electrode capacitance-coupled to plasma on the wafer or the like). Specifically, it is important to form a configuration having a detection circuit that is not susceptible to the influence of resonance even if the resonances are present.
Such a configuration can be achieved by incorporating the Vpp detector incorporated in the matcher shown in FIGS. 1 and 2 into the electrode. This configuration is shown in FIG. 8. According to this configuration, the Vpp detector becomes unsusceptible to the influence of L1 to L4 causing the resonance and it becomes possible to convert a voltage generated directly at the electrode to a DC voltage and output the DC voltage.
Hereafter, a first embodiment obtained by making the structure shown in FIG. 8 concrete will be described.
FIG. 9 is a longitudinal section diagram of an etching chamber used in the present invention. In the present embodiment, an example of a VHF plasma etching apparatus for forming plasma by utilizing a VHF (Very High Frequency) and a magnetic field is shown. An upper opening part including a cylindrical processing vessel 104, a platelike antenna electrode 103 formed of a conductor such as silicon, and a dielectric window 102 formed of quartz and sapphire capable of transmitting electromagnetic waves is placed on a vacuum vessel 101 via a vacuum seal material 127, such as an O-ring, so as to be hermetically sealed. A processing chamber 105 is formed inside. A magnetic field generating coil 114 is provided on an outer periphery part of the processing chamber 104 so as to surround the processing chamber. The antenna electrode 103 has a perforated structure for letting an etching gas flow. A flon gas such as CF4, C4F6, C4F8, C5F8, CHF3 or CH2F2, an inert gas such as Ar or N2, or O2 or a gas containing an oxide such as CO is controlled by a flow rate adjuster (not illustrated) including an MFC (mass flow controller) provided in a gas supplier 107, and led into the processing chamber 105 via the gas supplier 107. Furthermore, a vacuum exhauster 106 is connected to the vacuum vessel 101. The inside of the processing chamber 105 is kept at a predetermined pressure by a vacuum exhauster (not illustrated) including an MP (turbo-molecular pump) provided in the vacuum exhauster 106 and a pressure governor (not illustrated) including an APC.
A coaxial line 111 is provided over the antenna electrode 103. A high frequency power supply for plasma generation (first high frequency power supply) 108 (having, for example, a frequency of 200 MHz) is connected to the antenna electrode 103 via the coaxial line 111, a coaxial waveguide 125 and a matcher 109. A substrate electrode 115 on which a wafer 116 can be disposed is provided in a lower part in the vacuum vessel 101. In the same way as the antenna electrode 103, a coaxial line 151 is provided under the substrate electrode 115. A wafer bias power supply (second high frequency power supply) 119 (having, for example, a frequency of 4 MHz) is connected to the substrate electrode 115 via the coaxial line 151, a coaxial waveguide 152, a power feeding cable 153, and a matcher 118. The coaxial line 151 and the coaxial waveguide 152 are, for example, the high frequency transmission part in the electrode shown in FIG. 2, and they are in the vacuum. The power feeding cable 153 is on the atmospheric pressure side. An electrostatic chuck electrode 124 having an electrostatic chuck function for adsorbing the wafer 116 electrostatically is buried in the substrate electrode 115. An electrostatic chuck power supply 123 is connected to the electrostatic chuck electrode 124 via a filter 122. The filter 122 passes through DC power from the electrostatic chuck power supply 123, and effectively cuts off power from the plasma generation high frequency power supply 108 and the wafer bias power supply 119.
In the present configuration, a wafer voltage measurement circuit 154 is incorporated right under the electrostatic chuck electrode 124 in the vacuum. The influence of the resonance is eliminated by thus attaching the measurement circuit directly to a place where the voltage to be measured is generated, converting the measured voltage to a DC voltage on the spot, and taking out a resultant signal to the outside of the vacuum. A composite impedance of C6 and C7 in the voltage measurement circuit 154 shown in FIG. 8 must be sufficiently high. To which degree the composite impedances must be high will be described with reference to a second embodiment. However, this method has several problems. The problems are: (1) electric parts (such as resistors, capacitors, coils and diodes) in use are premised on use in the atmosphere, and the performance is not assured for use in the vacuum; (2) since heat generation from the electric parts is inevitable and little heat is radiated in the vacuum, continuous use is impossible; the possibility that part degradation will be caused by a corrosive gas is high; (4) when film deposition occurs, the possibility that circuit operation will be affected is high; (5) the possibility that the circuit will be damaged by turnaround of the high frequency for plasma generation is high; and (6) the possibility that the circuit will be damaged or the circuit operation will be affected by plasma generated around the circuit because of turnaround of the high frequency for plasma generation is high. Each of these problems is not insoluble. For example, the problems can be solved by burying the whole of the voltage measurement circuit 154 into resin, housing the whole of the voltage measurement circuit 154 into a hermetically sealed structure to protect the voltage measurement circuit 154 from the corrosive gas, and housing the whole of the voltage measurement circuit 154 into a hermetically sealed vessel that can be shielded electromagnetically.
A second embodiment in which the problems of the first embodiment are solved more thoroughly is shown in FIG. 10.
In the present configuration, the voltage measurement point is the electrostatic chuck electrode 124 in the same way as FIG. 9. This voltage is taken out to the outside of the vacuum by using a coaxial cable 157. The voltage taken out to the outside of the vacuum is converted to a DC voltage signal by using the voltage measurement circuit 154. This configuration has a merit that the demerit of the configuration shown in FIG. 9 is eliminated because the voltage measurement circuit 154 can be disposed on the atmosphere side. As regards the voltage measurement, the above-described resonance phenomenon loses no relation because it suffices that the voltage at the electrostatic chuck electrode 124 is equal to the voltage at the voltage measurement circuit 154.
A special contrivance becomes necessary in the coaxial cable 157 and the voltage measurement circuit 154 in order to make the voltage at the electrostatic chuck electrode 124 equal to the voltage at the voltage measurement circuit 154.
An equivalent circuit for the apparatus shown in FIG. 10 is shown in FIG. 11. The equivalent circuit shown in FIG. 11 differs from the equivalent circuit shown in FIG. 8 in that a coaxial cable is inserted between the electrode and the voltage measurement circuit. The above-described special contrivance is to make a composite impedance Zs of the coaxial cable and the voltage measurement circuit sufficiently higher than a load impedance Zp inclusive of the plasma. If Zs is small, then a voltage drop is caused by Zs and large reactive current flows, resulting in a heavy burden on the transmission system. If an RF power supply shown in FIG. 11 is controlled to output constant power, then such a demerit is not eliminated completely, but it can be suppressed to a negligible level in an allowable range.
The relation between Zp and Zs will now be described in detail. When seen from the RF power supply shown in FIG. 11, Zp and Zs are connected in parallel as a load circuit. Therefore, a load impedance Z at the time when Zs is not coupled is Z=Zp, whereas a load impedance Z′ at the time when Zs is coupled becomes Z′=Zp·Zs/(Zp+Zs). On the other hand, when the RF power supply is used in power control, V1 which is the voltage to be measured is determined by V1=(WZ)̂0.5 where W is RF power. As a result, the ratio between a voltage V1′ at the time when Zs is coupled and a voltage V1 at the time when Zs is not coupled is represented by V1′/V1=(Zs/(Zp+Zs))̂0.5. Letting V1′/V1=α, α represents precision of the measured voltage value in the state in which the voltage measurement circuit is coupled. Thereafter, a is a value in the range of 0 to 1. From the foregoing description, the relation between Zp and Zs is represented by the following expression (5) using α.
If, for example, the voltage detection precision is made at least 95%, then it is appreciated from this expression that Zs must have an impedance that is at least 9.3 times as large as Zp. It is also possible to replace C6 and C7 in the voltage measurement circuit by resistors. Unless resistances of the resistors are sufficiently high (for example, at least 10 MΩ), however, power loss is caused in the resistors. Accordingly, care should be taken.
This will now be described by using concrete numerical values. A composite impedance Zp obtained by seeing the plasma side from the place of V1 is calculated by using C5=2000 pF, R3=160Ω and other constants. As a result, |Zp|=approximately 15Ω is obtained.
A composite impedance Zs obtained by seeing the voltage measurement circuit side from the place of V1 will now be found. Supposing a coaxial cable corresponding to 3D2V from the viewpoint of the withstand voltage, inductance and capacitance per unit length become 0.27 μH/m and 103 pF/m, respectively. These correspond to L5, L6 and C9 shown in FIG. 11. Supposing that composite capacitance of C6 and C7 is 8 pF and the coaxial cable has a length of 1 m, it follows that Zs=−355i Ω, where i is an imaginary number. It follows that |Zs/Zp|=24, and the measurement precision can be made sufficiently high.
Equivalent circuit simulation results obtained by using circuit constants heretofore described are shown in FIGS. 12A-12C. FIG. 12A shows a voltage ratio between V1 and V2 (indicated in FIG. 11) obtained when the voltage measurement circuit is not connected, and shows the same result as that of FIG. 6. V1/V2 ratio and V1/V3 ratio obtained when the voltage measurement circuit having the above-described circuit constants is connected are shown in FIGS. 12B and 12C, respectively. Comparing the V1/V2 ratio shown in FIG. 12A with that shown in FIG. 12B, it is appreciated that an influence of the voltage measurement circuit is noticeable at 40 MHz or above, but the influence of the voltage measurement circuit is hardly noticeable at 10 MHz or below.
In the V1/V3 ratio shown in FIG. 12C, a drop in voltage ratio caused by resonance in the vicinity of 40 MHz is noticeable. Calculation of the resonant frequency in the voltage measurement circuit is conducted as described below.
First, a composite impedance of L6, C6 and C7 shown in FIG. 11 will now be found. L6 corresponds to a coaxial cable having a length of 50 cm. Therefore, it follows that L6=0.135 μH. Since a composite capacitance of C6 and C7 is 8 pF, the composite impedance of L6, C6 and C7 becomes −5i kΩ. Since this is a capacitive impedance, 8.005 pF is obtained by converting the capacitive impedance into a capacitance.
A composite capacitance of this and C9 becomes 103 pF+8.005 pF=111.005 pF. Since this composite capacitance and L5 (=0.135 μH) cause serial resonance, the resonant frequency (hereafter referred to as Reso_Measure) becomes 41.113 MHz according to the Expression 1.
Because of a voltage variation caused by the resonance in the voltage measurement circuit, there must be a definite relation between the frequency of the voltage to be measured and the resonant frequency.
In order to hold down the voltage measurement precision to ±5%, a frequency satisfying the relation V1/V3>0.95 is checked particularly in the graph representing the V1/V3 ratio shown in FIG. 12C. As a result, the frequency is 8.9 MHz or below. Denoting the frequency of the voltage to be measured by fB, therefore, it follows that Reso_Measure/fB>41.113/8.9=4.6.
Denoting an inductance and a capacitance that determine the resonant frequency of the voltage measurement circuit respectively by L and C, therefore, it is necessary to satisfy the following expression (6) on the basis of the Expression (1).
LC<(9.2 τfB)−2 (6)
The coefficient 9.2 need not be this value necessarily. Since this coefficient depends on the voltage measurement precision, this coefficient should be determined with respect to a required measurement precision by using simulation or actual measurement. For example, if the measurement precision is set to +10% under the same condition as that in FIG. 12C, the frequency satisfying the relation V1/V3>0.90 becomes 12.6 MHz or below, and the coefficient becomes 6.5 (=41.113/12.6*2).
Phase measurement will now be described. If a rectifier circuit using a diode D1 included in the voltage measurement circuit shown in FIGS. 8 and 11 is replaced by a phase detection circuit, the voltage phase can be measured. Block diagrams corresponding to FIGS. 8 and 11 are shown in FIGS. 13 and 14, respectively. Results obtained by simulating phase differences of V1/V2 and V1/V3 in FIG. 14 under the same conditions as FIGS. 12A-12C are shown in FIGS. 15A and 15B, respectively. The phase difference between V1 and V2 exhibits complicated behavior. The phase difference between V1 and V3 suddenly changes from 0° to 180° at a resonant frequency of 41 MHz. This is because the phase detection circuit is formed of only an inductance and a capacitance without using resistances. If resistances are used, then unadvantageously the phase difference exhibits a comparatively gently-sloping change. It is appreciated from this result that there are no problems as regards the phase measurement as long as the restriction represented by the Expression 6 is observed.
The circuit concerning the voltage measurement and phase measurement heretofore described can be applied to not only the electrode having a wafer mounted thereon, but also all electrodes capacitance-coupled to plasma. This embodiment will now be described.
FIG. 16 is a longitudinal section diagram of an etching chamber used in the present invention. FIG. 16 differs from FIG. 10 in that not only the high frequency power supply for plasma generation (the first high frequency power supply) 108 (having a frequency of, for example, 200 MHz) is connected to the antenna electrode 103 via the matcher 109 but also an antenna bias power supply 113 which is a third high frequency power supply is connected to the antenna electrode 103 via a matcher 112. The antenna bias power supply 113 and the wafer bias power supply 119 are connected to a phase controller 120. As a result, phases of the high frequencies output from the antenna bias power supply 113 and the wafer bias power supply 119 can be controlled. In this case, the antenna bias power supply 113 and the wafer bias power supply 119 are made to have the same frequency (for example, 4 MHz). In this system, a difference in phase (for example, 180°) between the antenna biasing high frequency appearing on the antenna electrode 103 and the wafer biasing high frequency appearing on the wafer 116 is controlled, and a bias can be applied to each of the antenna electrode 103 and the wafer 116 effectively. For implementing such a system, the voltage and phase at the electrostatic chuck electrode 124 are detected by pulling out the voltage to the atmospheric pressure side by the use of the coaxial cable 157 and providing a phase measurement circuit 155. In order to detect the voltage and phase at the upper antenna electrode 103, the voltage at the antenna electrode 103 is taken out to the atmospheric pressure side by using a coaxial cable 159 and a phase measurement circuit 156 is provided, in the same way as the lower electrode. A phase controller 120 compares phases obtained from the two phase measurement circuits 155 and 156, and determines a phase difference in high frequencies to be sent to the antenna bias power supply 113 and the wafer bias power supply 119 so as to generate a predetermined phase difference.
In order to raise the reliability of the control, the matcher 109 incorporates a filter 110 for cutting off the frequency of the antenna bias power supply 113. In the same way, the matcher 112 incorporates a filter 121 for cutting off the frequency of the high frequency power supply 108 for plasma generation. Outputs of the two matchers 109 and 112 are combined by using a coaxial cable 158, and a resultant signal is coupled to the coaxial line 111 which is the high frequency transmission system for the antenna electrode.
It should be further understood by those skilled in the art that although the foregoing description has been made on embodiments of the invention, the invention is not limited thereto and various changes and modifications may be made without departing from the spirit of the invention and the scope of the appended claims.
Patent Application
20070235135
